AD8317ACPZ-WP [ADI]
1 MHz to 10 GHz, 50 dB Log Detector/Controller; 1 MHz至10 GHz的50 dB的对数检测器/控制器
| 型号： | AD8317ACPZ-WP |
| 厂家： | 
ADI |
| 描述： | 1 MHz to 10 GHz, 50 dB Log Detector/Controller |
| 文件： | 总20页 (文件大小：1085K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
1 MHz to 10 GHz, 50 dB
Log Detector/Controller
AD8317
FUNCTIONAL BLOCK DIAGRAM
FEATURES
VPOS
TADJ
Wide bandwidth: 1 MHz to 10 GHz
High accuracy: 1.0 dB over temperature
50 dB dynamic range up to 8 GHz
Stability over temperature 0.5 dB
Low noise measurement/controller output VOUT
Pulse response time: 8/10 ns (fall/rise)
Small footprint 2 mm x 3 mm CSP package
Supply operation: 3.0 V to 5.5 V @ 22 mA
Fabricated using high speed SiGe process
GAIN
BIAS
SLOPE
VSET
I
I
V
V
VOUT
CLPF
DET
DET
DET
DET
INHI
INLO
COMM
APPLICATIONS
Figure 1.
RF transmitter PA setpoint control and level monitoring
Power monitoring in radiolink transmitters
RSSI measurement in base stations, WLAN, WiMAX, radar
GENERAL DESCRIPTION
The feedback loop through an RF amplifier is closed via VOUT,
the output of which regulates the amplifier’s output to a magnitude
corresponding to VSET. The AD8317 provides 0 V to (VPOS − 0.1 V)
output capability at the VOUT pin, suitable for controller applica-
tions. As a measurement device, VOUT is externally connected to
VSET to produce an output voltage VOUT that is a decreasing
linear-in-dB function of the RF input signal amplitude.
The AD8317 is a demodulating logarithmic amplifier, capable
of accurately converting an RF input signal to a corresponding
decibel-scaled output. It employs the progressive compression
technique over a cascaded amplifier chain, each stage of which
is equipped with a detector cell. The device can be used in
either measurement or controller modes. The AD8317 maintains
accurate log conformance for signals of 1 MHz to 8 GHz and
provides useful operation to 10 GHz. The input dynamic range
is typically 50 dB (re: 50 Ω) with error less than ±1 dB. The AD8317
has 8/10 ns response time (fall time/rise time) that enables RF
burst detection to a pulse rate of beyond 50 MHz. The device
provides unprecedented logarithmic intercept stability vs. ambient
temperature conditions. A supply of 3.0 V to 5.5 V is required to
power the device. Current consumption is typically 22 mA, and it
decreases to 200 μA when the device is disabled.
The logarithmic slope is −22 mV/dB, determined by the VSET
interface. The intercept is +15 dBm (re: 50 Ω, CW input) using
the INHI input. These parameters are very stable against supply
and temperature variations.
The AD8317 is fabricated on a SiGe bipolar IC process and is
available in a 2 mm × 3 mm, 8-lead LFCSP_VD package for an
operating temperature range of −40oC to +85oC.
The AD8317 can be configured to provide a control voltage to
a power amplifier or a measurement output from the VOUT
pin. Because the output can be used for controller applications,
special attention has been paid to minimize wideband noise. In this
mode, the setpoint control voltage is applied to the VSET pin.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registeredtrademarks arethe property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
Fax: 781.461.3113
www.analog.com
© 2005 Analog Devices, Inc. All rights reserved.
AD8317
TABLE OF CONTENTS
Features .............................................................................................. 1
Input Signal Coupling................................................................ 12
Output Interface ......................................................................... 12
Setpoint Interface ....................................................................... 12
Temperature Compensation of Output Voltage..................... 13
Measurement Mode ................................................................... 13
Setting the Output Slope in Measurement Mode .................. 14
Controller Mode......................................................................... 14
Output Filtering.......................................................................... 16
Operation Beyond 8 GHz ......................................................... 16
Evaluation Board ............................................................................ 17
Outline Dimensions....................................................................... 19
Ordering Guide .......................................................................... 19
Applications....................................................................................... 1
General Description......................................................................... 1
Functional Block Diagram .............................................................. 1
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Absolute Maximum Ratings............................................................ 6
ESD Caution.................................................................................. 6
Pin Configuration and Function Descriptions............................. 7
Typical Performance Characteristics ............................................. 8
Theory of Operation ...................................................................... 11
Using the AD8317 .......................................................................... 12
Basic Connections...................................................................... 12
REVISION HISTORY
10/05—Revision 0: Initial Version
Rev. 0 | Page 2 of 20
AD8317
SPECIFICATIONS
VPOS = 3 V, CLPF = 1000 pF, TA = 25°C, 52.3 ꢀ termination resistor at INHI, unless otherwise noted.
Table 1.
Parameter
Conditions
Min
Typ
Max
Unit
SIGNAL INPUT INTERFACE
Specified Frequency Range
DC Common-Mode Voltage
MEASUREMENT MODE
INHI (Pin 1)
0.001
10
GHz
V
VPOS – 0.6
VOUT (Pin 5) shorted to VSET (Pin 4), sinusoidal
input signal
f = 900 MHz
RTADJ = 18 kΩ
Input Impedance
±1 dB Dynamic Range
1500||0.33
50
Ω||pF
dB
TA = +25°C
46
dB
−40°C < TA < +85°C
±1 dB error
Maximum Input Level
Minimum Input Level
Slope1
−3
dBm
dBm
mV/dB
dBm
V
−53
−22
15
±1 dB error
−25
12
−19.5
21
Intercept1
Output Voltage: High Power In
Output Voltage: Low Power In
f = 1.9 GHz
PIN = –10 dBm
PIN = –40 dBm
RTADJ = 8 kΩ
0.42
1.00
0.58
1.27
0.78
1.40
V
Input Impedance
950||0.38
50
Ω||pF
dB
±1 dB Dynamic Range
TA = +25°C
48
dB
−40°C < TA < +85°C
±1 dB error
Maximum Input Level
Minimum Input Level
Slope1
−4.00
−54
−22
14
dBm
dBm
mV/dB
dBm
V
±1 dB error
−25
10
−19.5
20
Intercept1
Output Voltage: High Power In
Output Voltage: Low Power In
f = 2.2 GHz
PIN = –10 dBm
PIN = –35 dBm
RTADJ = 8 kΩ
0.35
0.75
0.54
1.21
0.80
1.35
V
Input Impedance
810||0.39
50
Ω||pF
dB
±1 dB Dynamic Range
TA = +25°C
47
dB
−40°C < TA < +85°C
±1 dB error
Maximum Input Level
Minimum Input Level
Slope1
−5
dBm
dBm
mV/dB
dBm
V
−55
−22
14
±1 dB error
Intercept1
Output Voltage: High Power In
Output Voltage: Low Power In
f = 3.6 GHz
PIN = –10 dBm
PIN = –40 dBm
RTADJ = 8 kΩ
0.53
1.20
V
Input Impedance
300||0.33
42
Ω||pF
dB
±1 dB Dynamic Range
TA = +25°C
40
dB
−40°C < TA < +85°C
±1 dB error
Maximum Input Level
Minimum Input Level
Slope1
−6
dBm
dBm
mV/dB
dBm
V
−48
−22
11
±1 dB error
Intercept1
Output Voltage: High Power In
Output Voltage: Low Power In
PIN = –10 dBm
PIN = –40 dBm
Rev. 0 | Page 3 of 20
0.47
1.16
V
AD8317
Parameter
Conditions
Min
Typ
Max
Unit
f = 5.8 GHz
RTADJ = 500 Ω
Input Impedance
110||0.05
50
Ω||pF
dB
±1 dB Dynamic Range
TA = +25°C
48
dB
−40°C < TA < +85°C
±1 dB error
Maximum Input Level
Minimum Input Level
Slope1
−4
dBm
dBm
mV/dB
dBm
V
−54
−22
16
±1 dB error
Intercept1
Output Voltage: High Power In
Output Voltage: Low Power In
f = 8.0 GHz
PIN = –10 dBm
PIN = –40 dBm
RTADJ = open
0.59
1.27
V
Input Impedance
28||0.79
44
Ω||pF
dB
±1 dB Dynamic Range
TA = +25°C
dB
−40°C < TA < +85°C
±1 dB error
Maximum Input Level
Minimum Input Level
Slope2
−2
dBm
dBm
mV/dB
dBm
V
−46
−22
21
±1 dB error
Intercept2
Output Voltage: High Power In
Output Voltage: Low Power In
OUTPUT INTERFACE
Voltage Swing
PIN = –10 dBm
0.70
1.39
PIN = –40 dBm
V
VOUT (Pin 5)
VSET = 0 V, RFIN = open
VSET = 1.7 V, RFIN = open
VSET = 0 V, RFIN = open
RFIN = −10 dBm, from CLPF to VOUT
VPOS – 0.1
10
V
mV
Output Current Drive
Small Signal Bandwidth
Output Noise
10
mA
140
90
MHz
nV/√Hz
RF Input = 2.2 GHz, –10 dBm, fNOISE = 100 kHz,
CLPF = open
Fall Time
Fall Time
Rise Time
Rise Time
Input level = no signal to –10 dBm, 90% to 10%,
CLPF = 8 pF
Input level = no signal to –10 dBm, 90% to 10%,
18
6
ns
ns
C
LPF = open; ROUT = 150 Ω
Input level = −10 dBm to no signal, 10% to 90%,
CLPF = 8 pF
Input level = −10 dBm to no signal, 10% to 90%,
CLPF = open, ROUT = 150 Ω
20
10
50
ns
ns
Video Bandwidth (or Envelope
Bandwidth)
MHz
VSET INTERFACE
VSET (Pin 4)
Nominal Input Range
RFIN = 0 dBm, measurement mode
RFIN = –50 dBm, measurement mode
0.35
1.40
−45
40
V
V
Logarithmic Scale Factor
Input Resistance
dB/V
kΩ
RFIN = −20 dBm, controller mode, VSET = 1 V
TADJ (Pin 6)
TADJ INTERFACE
Input Resistance
TADJ = 0.9 V, sourcing 50 μA
TADJ = open
13
kΩ
V
Disable Threshold Voltage
VPOS – 0.4
Rev. 0 | Page 4 of 20
AD8317
Parameter
Conditions
Min
Typ
Max
Unit
POWER INTERFACE
Supply Voltage
Quiescent Current
vs. Temperature
Disable Current
VPOS (Pin 7)
3.0
18
5.5
30
V
22
mA
μA/°C
μA
60
−40°C ≤ TA ≤ +85°C
TADJ = VPOS
200
1 Slope and intercept are determined by calculating the best-fit line between the power levels of −40 dBm and −10 dBm at the specified input frequency.
2 Slope and intercept are determined by calculating the best-fit line between the power levels of −34 dBm and −16 dBm at 8.0 GHz.
Rev. 0 | Page 5 of 20
AD8317
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Rating
Supply Voltage: VPOS
VSET Voltage
5.7 V
0 to VPOS
12 dBm
0.73
Input Power (Single-Ended, Re: 50 Ω)
Internal Power Dissipation
θJA
Maximum Junction Temperature
Operating Temperature Range
Storage Temperature Range
Lead Temperature (Soldering 60 sec)
55°C/W
125°C
−40°C to +85°C
−65°C to +150°C
260°C
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0 | Page 6 of 20
AD8317
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
INHI
COMM
CLPF
1
2
3
4
8
7
6
5
INLO
VPOS
TADJ
VOUT
AD8317
TOP VIEW
(Not to Scale)
VSET
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin
No.
Mnemonic Description
1
INHI
RF Input. Nominal input range of −50 dBm to 0 dBm, re: 50 Ω; ac-coupled RF input.
2
3
COMM
CLPF
Device Common. Connect to a low impedance ground plane.
Loop Filter Capacitor. In measurement mode, this capacitor sets the pulse response time and video bandwidth. In
controller mode, the capacitance on this node sets the response time of the error amplifier/integrator.
4
5
VSET
VOUT
Setpoint Control Input for Controller Mode or Feedback Input for Measurement Mode.
Measurement and Controller Output. In measurement mode, VOUT provides a decreasing linear-in dB representation
of the RF input signal amplitude. In controller mode, VOUT is used to control the gain of a VGA or VVA with a positive
gain sense (increasing voltage increases gain).
6
TADJ
Temperature Compensation Adjustment. Frequency-dependent temperature compensation is set by connecting a
ground-referenced resistor to this pin.
7
8
VPOS
INLO
Positive Supply Voltage: 3.0 V to 5.5 V.
RF Common for INHI. AC-coupled RF common.
Paddle
Internally connected to COMM; solder to a low impedance ground plane.
Rev. 0 | Page 7 of 20
AD8317
TYPICAL PERFORMANCE CHARACTERISTICS
VPOS = 3 V; T = 25°C, –40°C, +85°C; CLPF = 1000 pF, unless otherwise noted. Colors: 25°C ꢀ Black; −40°C ꢀ Blue; 85°C ꢀ Red.
Error is calculated by using the best-fit line between PIN = −40 dBm and PIN = −10 dBm at the specified input frequency, unless otherwise noted
2.0
2.00
2.0
2.00
1.75
1.5
1.75
1.5
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
0.5
0.5
0
0
–0.5
–1.0
–1.5
–2.0
–0.5
–1.0
–1.5
–2.0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
P
(dBm)
P
(dBm)
IN
IN
Figure 6. VOUT and Log Conformance vs. Input Amplitude at 3.6 GHz,
RTADJ = 8 kΩ
Figure 3. VOUT and Log Conformance vs. Input Amplitude at 900 MHz,
RTADJ = 18 kΩ
2.0
1.5
2.00
1.75
2.0
2.00
1.75
1.5
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
0.5
0.5
0
0
–0.5
–1.0
–1.5
–2.0
–0.5
–1.0
–1.5
–2.0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
P
(dBm)
IN
P
(dBm)
IN
Figure 7. VOUT and Log Conformance vs. Input Amplitude at 5.8 GHz,
RTADJ = 500 Ω
Figure 4. VOUT and Log Conformance vs. Input Amplitude at 1.9 GHz,
RTADJ = 8 kΩ
2.0
1.5
2.00
1.75
2.0
2.00
1.75
1.50
1.25
1.00
0.75
0.50
0.25
0
1.5
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
1.0
0.5
0.5
0
0
–0.5
–1.0
–1.5
–2.0
–0.5
–1.0
–1.5
–2.0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
P
(dBm)
P
(dBm)
IN
IN
Figure 8. VOUT and Log Conformance vs. Input Amplitude at 8.0 GHz,
RTADJ = Open, Error Calculated from PIN = −34 dBm to PIN = −16 dBm
Figure 5. VOUT and Log Conformance vs. Input Amplitude at 2.2 GHz,
TADJ = 8 kΩ
R
Rev. 0 | Page 8 of 20
AD8317
2.0
1.5
2.00
1.75
1.50
1.25
1.00
0.75
0.50
0.25
0
2.0
2.00
1.75
1.5
1.0
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
0.5
0.5
0
0
–0.5
–1.0
–1.5
–2.0
–0.5
–1.0
–1.5
–2.0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
10
10
P
(dBm)
IN
P
(dBm)
IN
Figure 12. VOUT and Log Conformance vs. Input Amplitude at 3.6 GHz,
Multiple Devices, RTADJ = 8 kΩ
Figure 9. VOUT and Log Conformance vs. Input Amplitude at 900 MHz,
Multiple Devices, RTADJ = 18 kΩ
2.0
1.5
2.00
1.75
2.0
1.5
2.00
1.75
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
0.5
0.5
0
0
–0.5
–1.0
–1.5
–2.0
–0.5
–1.0
–1.5
–2.0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
10
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
10
P
(dBm)
P
(dBm)
IN
IN
Figure 13. VOUT and Log Conformance vs. Input Amplitude at 5.8 GHz,
Multiple Devices, RTADJ = 500 Ω
Figure 10. VOUT and Log Conformance vs. Input Amplitude at 1.9 GHz,
Multiple Devices, RTADJ = 8 kΩ
2.00
1.75
2.0
2.0
1.5
2.00
1.75
1.5
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
1.0
1.50
1.25
1.00
0.75
0.50
0.25
0
0.5
0.5
0
0
–0.5
–1.0
–1.5
–2.0
–0.5
–1.0
–1.5
–2.0
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
10
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
P
(dBm)
P
(dBm)
IN
IN
Figure 14. VOUT and Log Conformance vs. Input Amplitude at 8.0 GHz,
Multiple Devices, RTADJ =Open,
Figure 11. VOUT and Log Conformance vs. Input Amplitude at 2.2 GHz,
Multiple Devices, RTADJ = 8 kΩ
Error Calculated from PIN = −34 dBm to PIN = −16 dBm
Rev. 0 | Page 9 of 20
AD8317
j1
j2
j0.5
10000
1000
100
j0.2
–60dBm
RF OFF
0
0.2
0.5
1
2
–40dBm
–20dBm
100MHz
900MHz
–j0.2
–10dBm
1900MHz
2200MHz
8000MHz
–j0.5
–j2
0dBm
3600MHz
–j1
START FREQUENCY = 0.05GHz
STOP FREQUENCY = 10GHz
10
1k
10M
10k
100k
1M
5800MHz
10000MHz
FREQUENCY (Hz)
Figure 15. Input Impedance vs. Frequency; No Termination Resistor on INHI
(Impedance De-Embedded to Input Pins), Z0 = 50 Ω
Figure 18. Noise Spectral Density of Output; CLPF = Open
10000
Δ : 1.86V
@ : 1.69V
1000
100
10
3
4
Ch3 500mV Ch4 200mV
M4.00μs
A Ch3
620mV
10M
1k
10k
100k
1M
T
12.7560μs
FREQUENCY (Hz)
Figure 16. Power On/Off Response Time; VP = 3.0 V;
Input AC-Coupling Caps = 10 pF; CLPF = Open
Figure 19. Noise Spectral Density of Output Buffer (from CLPF to VOUT);
CLPF = 0.1 μF
2.0
2.00
1.75
3.3V
3.0V
1.5
3.6V
1.0
1.50
1.25
1.00
0.75
0.50
0.25
Ch1 RISE
10.44ns
0.5
Ch1 FALL
6.113ns
0
–0.5
–1.0
–1.5
–2.0
0
CH1 200mV
M20.0ns
943.600ns
A CH1
1.40V
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
10
T
P
(dBm)
IN
Figure 17. VOUT Pulse Response Time; Pulsed RF Input 0.1 GHz, −10 dBm;
LPF = Open; RLOAD = 150 Ω
Figure 20. Output Voltage Stability vs. Supply Voltage at 1.9 GHz
When VPOS Varies by 10%
C
Rev. 0 | Page 10 of 20
AD8317
THEORY OF OPERATION
The AD8317 is a 6-stage demodulating logarithmic amplifier,
specifically designed for use in RF measurement and power control
applications at frequencies up to 10 GHz. A block diagram is
shown in Figure 21. Sharing much of its design with the AD8318
logarithmic detector/controller, the AD8317 maintains tight inter-
cept variability vs. temperature over a 50 dB range. Additional
enhancements over the AD8318, such as reduced RF burst
response time of 8 ns to 10 ns, 22 mA supply current, and board
space requirements of only 2 mm x 3 mm, add to the low cost
and high performance benefits found in the AD8317.
The logarithmic function is approximated in a piecewise
fashion by six cascaded gain stages. (For a more comprehensive
explanation of the logarithm approximation, please refer to the
AD8307 data sheet, available at www.analog.com.) The cells
have a nominal voltage gain of 9 dB each and a 3 dB bandwidth
of 10.5 GHz. Using precision biasing, the gain is stabilized over
temperature and supply variations. The overall dc gain is high,
due to the cascaded nature of the gain stages. An offset
compensation loop is included to correct for offsets within the
cascaded cells. At the output of each of the gain stages, a square-
law detector cell is used to rectify the signal.
VPOS
TADJ
The RF signal voltages are converted to a fluctuating differential
current having an average value that increases with signal level.
Along with the six gain stages and detector cells, an additional
detector is included at the input of the AD8317, providing a 50 dB
dynamic range in total. After the detector currents are summed
and filtered, the following function is formed at the summing node:
GAIN
BIAS
SLOPE
VSET
I
V
VOUT
CLPF
I
V
DET
DET
DET
DET
INHI
INLO
ID × log10(VIN/VINTERCEPT
)
where:
COMM
Figure 21. Block Diagram
ID is the internally set detector current.
VIN is the input signal voltage.
VINTERCEPT is the intercept voltage (that is, when VIN = VINTERCEPT
the output voltage would be 0 V, if it were capable of going to 0 V).
A fully differential design, using a proprietary, high speed SiGe
process, extends high frequency performance. Input INHI receives
the signal with a low frequency impedance of nominally 500 Ω
in parallel with 0.7 pF. The maximum input with 1 dB log-
conformance error is typically 0 dBm (re: 50 Ω). The noise
spectral density referred to the input is 1.15 nV/√Hz, which is
equivalent to a voltage of 118 ꢁV rms in a 10.5 GHz bandwidth
or a noise power of −66 dBm (re: 50 Ω). This noise spectral
density sets the lower limit of the dynamic range. However, the
low end accuracy of the AD8317 is enhanced by specially shaping
the demodulating transfer characteristic to partially compensate
for errors due to internal noise. The common pin, COMM,
provides a quality low impedance connection to the printed circuit
board (PCB) ground. The package paddle, which is internally
connected to the COMM pin, should also be grounded to the
PCB to reduce thermal impedance from the die to the PCB.
,
Rev. 0 | Page 11 of 20
AD8317
USING THE AD8317
While the input can be reactively matched, in general this is not
necessary. An external 52.3 Ω shunt resistor (connected on the
signal side of the input coupling capacitors, as shown in
Figure 22) combines with the relatively high input impedance
to give an adequate broadband 50 Ω match.
BASIC CONNECTIONS
The AD8317 is specified for operation up to 10 GHz; as a result,
low impedance supply pins with adequate isolation between
functions are essential. A power supply voltage of between 3.0 V
and 5.5 V should be applied to VPOS. Power supply decoupling
capacitors of 100 pF and 0.1 ꢁF should be connected close to
this power supply pin.
The coupling time constant, 50 × CC/2, forms a high-pass
corner with a 3 dB attenuation at fHP = 1/(2π × 50 × CC ), where
C1 = C2 = CC. Using the typical value of 47 nF, this high pass
corner will be ~68 kHz. In high frequency applications, fHP
should be as large as possible to minimize the coupling of
unwanted low frequency signals. In low frequency applications,
a simple RC network forming a low-pass filter should be added
at the input for similar reasons. This should generally be placed
at the generator side of the coupling capacitors, thereby
lowering the required capacitance value for a given high-pass
corner frequency.
V (2.7V–5.5V)
S
C5
0.1μF
R2
0Ω
C4
100pF
SEE TEXT
C2
V
OUT
47nF
8
7
6
5
INLO VPOS
TADJ
VOUT
R4
0Ω
R1
52.3Ω
AD8317
INHI
1
COMM CLPF
VSET
4
OUTPUT INTERFACE
2
3
C1
SIGNAL
INPUT
The VOUT pin is driven by a PNP output stage. An internal 10 ꢀ
resistor is placed in series with the output and the VOUT pin.
The rise time of the output is limited mainly by the slew on
CLPF. The fall time is an RC-limited slew given by the load
capacitance and the pull-down resistance at VOUT. There is an
internal pull-down resistor of 1.6 kꢀ. A resistive load at VOUT
is placed in parallel with the internal pull-down resistor to
provide additional discharge current.
47nF
SEE TEXT
Figure 22. Basic Connections
The paddle of the LFCSP_VD package is internally connected
to COMM. For optimum thermal and electrical performance,
the paddle should be soldered to a low impedance ground plane.
INPUT SIGNAL COUPLING
VPOS
The RF input (INHI) is single-ended and must be ac-coupled.
INLO (input common) should be ac-coupled to ground.
Suggested coupling capacitors are 47 nF ceramic 0402-style
capacitors for input frequencies of 1 MHz to 10 GHz. The
coupling capacitors should be mounted close to the INHI and
INLO pins. The coupling capacitor values can be increased to
lower the input stage’s high-pass cutoff frequency. The high-pass
corner is set by the input coupling capacitors and the internal
10 pF high-pass capacitor. The dc voltage on INHI and INLO is
CLPF
10
Ω
VOUT
+
0.8V
–
1200
Ω
400
Ω
COMM
Figure 24. Output Interface
To reduce the fall time, VOUT should be loaded with a resistive
load of <1.6 kꢀ. For example, with an external load of 150 ꢀ the
AD8317 fall time is <7 ns.
about one diode voltage drop below VPOS
.
VPOS
5pF
CURRENT
5pF
SETPOINT INTERFACE
The VSET input drives the high impedance (20 kΩ) input of an
internal op amp. The VSET voltage appears across the internal
1.5 kΩ resistor to generate ISET. When a portion of VOUT is applied
to VSET, the feedback loop forces
FIRST
GAIN
18.7kΩ
INHI
18.7kΩ
STAGE
2kΩ
A = 9dB
INLO
−ID × log10(VIN/VINTERCEPT) = ISET
.
Gm
STAGE
OFFSET
COMP
If VSET = VOUT/2x, then ISET = VOUT/(2x × 1.5 kΩ).
Figure 23. Input Interface
The result is
V
OUT = (−ID × 1.5 kΩ × 2x) × log10(VIN/VINTERCEPT)
Rev. 0 | Page 12 of 20
AD8317
I
MEASUREMENT MODE
SET
20kΩ
V
SET
VSET
When the VOUT voltage or a portion of the VOUT voltage is fed
back to the VSET pin, the device operates in measurement
mode. As seen in Figure 27, the AD8317 has an offset voltage,
a negative slope, and a VOUT measurement intercept at the high
end of its input signal range.
20kΩ
1.5kΩ
COMM
COMM
2.00
1.75
1.50
1.25
1.00
0.75
0.50
0.25
0
2.0
1.5
1.0
0.5
0
Figure 25. VSET Interface
V
25°C
OUT
ERROR 25°C
The slope is given by –ID × 2x × 1.5 kΩ = −22 mV/dB × x. For
example, if a resistor divider to ground is used to generate a VSET
voltage of VOUT/2, then x = 2. The slope is set to −880 V/decade
or −44 mV/dB.
TEMPERATURE COMPENSATION OF OUTPUT
VOLTAGE
–0.5
–1.0
The primary component of the variation in VOUT vs. temperature,
as the input signal amplitude is held constant, is the drift of the
intercept. This drift is also a weak function of the input signal
frequency, so provision is made for optimization of internal
temperature compensation at a given frequency by providing
Pin TADJ.
RANGE FOR
CALCULATION OF
SLOPE AND INTERCEPT
–1.5
–60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0
5
10 15
INTERCEPT
P
(dBm)
IN
Figure 27. Typical Output Voltage vs. Input Signal
AD8317
V
INTERNAL
The output voltage vs. input signal voltage of the AD8317 is
linear-in-dB over a multidecade range. The equation for this
function is of the form
I
COMP
TADJ
R
TADJ
V
OUT = X × VSLOPE/DEC × log10(VIN/VINTERCEPT) =
(1)
(2)
1.5kΩ
X × VSLOPE/dB × 20 × log10(VIN/VINTERCEPT
)
COMM
COMM
where:
X is the feedback factor in VSET = VOUT/X.
Figure 26. TADJ Interface
V
V
SLOPE/DEC is nominally –440 mV/decade or −22 mV/dB.
INTERCEPT is the x-axis intercept of the linear-in-dB portion of
The Resistor RTADJ is connected between this pin and ground.
The value of this resistor partially determines the magnitude
of an analog correction coefficient, which is used to reduce
intercept drift.
the VOUT vs. VIN curve (Figure 27).
INTERCEPT is +2 dBV for a sinusoidal input signal.
V
An offset voltage, VOFFSET, of 0.35 V is internally added to the
detector signal, so that the minimum value for VOUT is
X × VOFFSET. So for X = 1, minimum VOUT is 0.35 V.
The relationship between output temperature drift and
frequency is not linear and cannot be easily modeled. As
a result, experimentation is required to choose the correct
TADJ resistor. Table 4 shows the recommended values for
some commonly used frequencies.
The slope is very stable vs. process and temperature variation.
When base-10 logarithms are used, VSLOPE/DECADE represents the
volts/decade. A decade corresponds to 20 dB; VSLOPE/DECADE/20 =
Table 4: Recommended RTADJ Resistor Values
Frequency
50 MHz
100 MHz
900 MHz
1.8 GHz
1.9 GHz
2.2 GHz
3.6 GHz
5.3 GHZ
5.8 GHz
8 GHz
Recommended RTADJ
V
SLOPE/dB represents the slope in volts/dB.
18 kΩ
18 kΩ
18 kΩ
8 kΩ
8 kΩ
8 kΩ
As noted in Equation 1 and Equation 2, the VOUT voltage has a
negative slope. This is also the correct slope polarity to control
the gain of many power amplifiers in a negative feedback
configuration. Because both the slope and intercept vary slightly
with frequency, it is recommended to refer to the Specifications
section for application-specific values for slope and intercept.
8 kΩ
500 Ω
500 Ω
Open
Rev. 0 | Page 13 of 20
AD8317
Although demodulating log amps respond to input signal voltage,
not input signal power, it is customary to discuss the amplitude
of high frequency signals in terms of power. In this case, the charac-
teristic impedance of the system, Z0, must be known to convert
voltages to their corresponding power levels. The following
equations are used to perform this conversion:
CONTROLLER MODE
The AD8317 provides a controller mode feature at the VOUT
pin. Using VSET for the setpoint voltage, it is possible for the
AD8317 to control subsystems, such as power amplifiers (PAs),
variable gain amplifiers (VGAs), or variable voltage attenuators
(VVAs) that have output power that increases monotonically
with respect to their gain control signal.
P(dBm) = 10 × log10(Vrms2/(Z0 × 1 mW))
P(dBV) = 20 × log10(Vrms/1 Vrms)
(3)
(4)
(5)
To operate in controller mode, the link between VSET and
VOUT is broken. A setpoint voltage is applied to the VSET
input, VOUT is connected to the gain control terminal of the
variable gain amplifier (VGA), and the detector’s RF input is
connected to the output of the VGA (usually using a directional
coupler and some additional attenuation). Based on the defined
relationship between VOUT and the RF input signal when the device
is in measurement mode, the AD8317 adjusts the voltage on
VOUT (VOUT is now an error amplifier output) until the level
at the RF input corresponds to the applied VSET. When the AD8317
operates in controller mode, there is no defined relationship
between VSET and VOUT voltage; VOUT settles to a value that results in
the correct input signal level appearing at INHI/INLO.
2
P(dBm) = P(dBV) − 10 × log10(Z0 × 1 mW/1 Vrms
)
For example, PINTERCEPT for a sinusoidal input signal expressed in
terms of dBm (decibels referred to 1 mW), in a 50 ꢀ system is
P
INTERCEPT(dBm) = PINTERCEPT (dBV) – 10 × log10(Z0 ×
1 mW/1 Vrms2) =
(6)
+2 dBV − 10 × log10(50×10-3) = +15 dBm
For a square wave input signal in a 200 ꢀ system,
2
P
INTERCEPT = −1 dBV − 10 × log10[(200 ꢀ × 1 mW/1Vrms )] =
+6 dBm
For this output power control loop to be stable, a ground-
referenced capacitor must be connected to the CLPF pin. This
capacitor, CFLT, integrates the error signal (in the form of a
current) to set the loop bandwidth and ensure loop stability.
Further details on control loop dynamics can be found in the
AD8315 data sheet.
Further information on the intercept variation dependence upon
waveform can be found in the AD8313 and AD8307 data sheets.
SETTING THE OUTPUT SLOPE
IN MEASUREMENT MODE
To operate in measurement mode, VOUT must be connected to
VSET. Connecting VOUT directly to VSET yields the nominal
logarithmic slope of approximately −22 mV/dB. The output
swing corresponding to the specified input range is then approxi-
mately 0.35 V to 1.7 V. The slope and output swing can be
increased by placing a resistor divider between VOUT and
VSET (that is, one resistor from VOUT to VSET and one
resistor from VSET to ground). The input impedance of VSET
is approximately 40 kꢀ. Slope-setting resistors should be kept
below 20 kꢀ to prevent this input impedance from affecting
the resulting slope. If two equal resistors are used (for example,
10 kꢀ/10 kꢀ), the slope doubles to approximately −44 mV/dB.
VGA/VVA
GAIN
RFIN
DIRECTIONAL
COUPLER
CONTROL
VOLTAGE
ATTENUATOR
47nF
47nF
VOUT
INHI
AD8317
52.3
Ω
DAC
VSET
INLO
CLPF
C
FLT
Figure 29. AD8317 Controller Mode
AD8317
VOUT
–44mV/dB
Decreasing VSET, which corresponds to demanding a higher
signal from the VGA, increases VOUT. The gain control voltage
of the VGA must have a positive sense. A positive control
voltage to the VGA increases the gain of the device.
10kΩ
10kΩ
VSET
The basic connections for operating the AD8317 in an automatic
gain control (AGC) loop with the ADL5330 are shown in
Figure 30. The ADL5330 is a 10 MHz to 3 GHz variable gain
amplifier. It offers a large gain control range of 60 dB with
0.5 dB gain stability. This configuration is similar to Figure 29.
Figure 28. Increasing the Slope
Rev. 0 | Page 14 of 20
AD8317
The gain of the ADL5330 is controlled by the output pin of the
AD8317. This voltage, VOUT, has a range of 0 V to near VPOS. To
avoid overdrive recovery issues, the AD8317 output voltage can
be scaled down using a resistive divider to interface with the 0 V
to 1.4 V gain control range of the ADL5330.
For the AGC loop to remain in equilibrium, the AD8317 must
track the envelope of the ADL5330’s output signal and provide
the necessary voltage levels to the ADL5330 gain control input.
Figure 32 shows an oscilloscope screenshot of the AGC loop
depicted in Figure 30. A 100 MHz sine wave with 50% AM
modulation is applied to the ADL5330. The output signal from
the VGA is a constant envelope sine wave with amplitude corre-
sponding to a setpoint voltage at the AD8317 of 1.5 V. Also shown
is the gain control response of the AD8317 to the changing input
envelope.
A coupler/attenuation of 21 dB is used to match the desired
maximum output power from the VGA to the top end of the
linear operating range of the AD8317 (approximately −5 dBm
at 900 MHz).
+5V
+5V
AM MODULATED INPUT
RF INPUT
SIGNAL
RF OUTPUT
SIGNAL
120nH
100pF
120nH
VPOS
COMM
OPHI
100pF
100pF
INHI
1
ADL5330
100pF
DIRECTIONAL
COUPLER
INLO
OPLO
GAIN
AD8317 OUTPUT
4.12kΩ
+5V
ATTENUATOR
10kΩ
VOUT
SETPOINT
VOLTAGE
VPOS
47nF
VSET
INHI
DAC
AD8317
52.3Ω
LOG AMP
3
CLPF
INLO
1nF
TADJ
COMM
47nF
18kΩ
ADL5330 OUTPUT
2
CH1 200mV
CH3 50.0mVΩ
Ch2 200mV
M2.00ms
640.00μs
A Ch2 820mV
Figure 30. AD8317 Operating in Controller Mode to Provide Automatic
Gain Control Functionality in Combination with the ADL5330
T
Figure 32. Oscilloscope Screenshot Showing an AM Modulated Input Signal
and the Response from the AD8317
Figure 31 shows the transfer function of the output power vs.
the VSET voltage over temperature for a 900 MHz sine wave with
an input power of −1.5 dBm. Note that the power control of the
AD8317 has a negative sense. Decreasing VSET, which corresponds
to demanding a higher signal from the ADL5330, increases gain.
Figure 33 shows the response of the AGC RF output to a pulse
on VSET. As VSET decreases from 1.7 V to 0.4 V, the AGC loop
responds with an RF burst. In this configuration the input signal to
the ADL5330 is a 1 GHz sine wave at a power level of −15 dBm.
T
The AGC loop is capable of controlling signals just under the
full 60 dB gain control range of the ADL5330. The performance
over temperature is most accurate over the highest power range,
where it is generally most critical. Across the top 40 dB range of
output power, the linear conformance error is well within 0.5 dB
over temperature.
AD8317 VSET PULSE
1
4
30
ADL5330 OUTPUT
2
20
3
2
10
0
1
CH1 2.00V
M10.0μs
699.800μs
CH2 50mVΩ
A CH1
2.48V
T
0
–10
–20
–30
–40
–50
Figure 33. Oscilloscope Screenshot Showing
the Response Time of the AGC Loop
–1
–2
–3
–4
Response time and the amount of signal integration are
controlled by CFLT. This functionality is analogous to the
feedback capacitor around an integrating amplifier. While
it is possible to use large capacitors for CFLT, in most applica-
tions values under 1 nF provide sufficient filtering.
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
1.8
2.0
SETPOINT VOLTAGE (V)
Figure 31. ADL5330 Output Power vs. AD8317 Setpoint Voltage,
PIN = −1.5 dBm
Rev. 0 | Page 15 of 20
AD8317
Calibration in controller mode is similar to the method used in
measurement mode. A simple two-point calibration can be
done by applying two known VSET voltages or DAC codes and
measuring the output power from the VGA. Slope and intercept
can then be calculated with the following equations:
In many log amp applications, it may be necessary to lower the
corner frequency of the postdemodulation filtering to achieve
low output ripple while maintaining a rapid response time to
changes in signal level. An example of a 4-pole active filter is
shown in the AD8307 data sheet.
Slope = (VSET1 − VSET2)/(POUT1 − POUT2
Intercept = POUT1 − VSET1/Slope
)
(7)
(8)
(9)
OPERATION BEYOND 8 GHZ
The AD8317 is specified for operation up to 8 GHz, but it
provides useful measurement accuracy over a reduced dynamic
range of up to 10 GHz. Figure 35 shows the performance of the
AD8317 over temperature at 10 GHz when the device is config-
ured as shown in Figure 22. Dynamic range is reduced at this
frequency, but the AD8317 does provide 30 dB of measurement
range within 3 dB of linearity error.
V
SETX = Slope × (POUTX − Intercept)
More information on the use of the ADL5330 in AGC
applications can be found in the ADL5330 data sheet.
OUTPUT FILTERING
2.0
1.8
1.6
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
5
For applications in which maximum video bandwidth and,
consequently, fast rise time are desired, it is essential that the
CLPF pin be left unconnected and free of any stray capacitance.
4
3
2
The nominal output video bandwidth of 50 MHz can be
reduced by connecting a ground-referenced capacitor (CFLT) to
the CLPF pin, as shown in Figure 34. This is generally done to
reduce output ripple (at twice the input frequency for a
symmetric input waveform such as sinusoidal signals).
1
0
–1
–2
–3
–4
–5
AD8317
I
LOG
VOUT
CLPF
+4
–40
–35
–30
–25
–20
–15
–10
–5
0
5
P
(dBm)
1.5kΩ
IN
3.5pF
Figure 35. VOUT and Log Conformance vs. Input Amplitude at 10.0 GHz, Multiple
Devices, RTADJ = Open, CLPF = 1000 pF
C
FLT
Implementing an impedance match for frequencies beyond 8 GHz
can improve the sensitivity of the AD8317 and measurement
range.
Figure 34. Lowering the Postdemodulation Bandwidth
CFLT is selected using the following equation:
Operation beyond 10 GHz is possible, but part-to-part
variation, most notably in the intercept, becomes significant.
1
(10)
− 3.5 pF
CFLT
=
(
π ×1.5 kΩ × Video Bandwidth
)
The video bandwidth should typically be set to a frequency
equal to about one-tenth the minimum input frequency. This
ensures that the output ripple of the demodulated log output,
which is at twice the input frequency, is well filtered.
Rev. 0 | Page 16 of 20
AD8317
EVALUATION BOARD
Table 5. Evaluation Board (Rev. A) Configuration Options
Component
VPOS, GND
R1, C1, C2
Function
Default Conditions
Not applicable
Supply and Ground Connections.
Input Interface.
R1 = 52.3 Ω (Size 0402)
The 52.3 Ω resistor in position R1 combines with the AD8317's internal input C1 = 47 nF (Size 0402)
impedance to give a broadband input impedance of about 50 Ω. Capacitor
C1 and Capacitor C2 are dc-blocking capacitors. A reactive impedance
match can be implemented by replacing R1 with an inductor and C1 and C2
with appropriately valued capacitors.
C2 = 47 nF (Size 0402)
R5, R7
Temperature Compensation Interface.
R5 = 200 Ω (Size 0402)
The internal temperature compensation network is optimized for input signals
up to 3.6 GHz when R7 is 10 kΩ. This circuit can be adjusted to optimize
performance for other input frequencies by changing the value of the
resistor in position R7. See Table 4 for specific TADJ resistor values.
R7 = open (Size 0402)
R2, R3, R4, R6, RL,
CL
Output Interface—Measurement Mode.
R2 = 0 Ω (Size 0402)
R3 = open (Size 0402)
R4 = open (Size 0402)
R6 = 1 kΩ (Size 0402)
In measurement mode, a portion of the output voltage is fed back to Pin VSET
via R2. The magnitude of the slope of the VOUT output voltage response can
be increased by reducing the portion of VOUT that is fed back to VSET. R6 can
be used as a back-terminating resistor or as part of a single-pole, low-pass
filter.
RL = CL = open (Size 0402)
R2, R3
Output Interface—Controller Mode.
R2 = open (Size 0402)
In this mode, R2 must be open. In controller mode, the AD8317 can control the R3 = open (Size 0402)
gain of an external component. A setpoint voltage is applied to Pin VSET, the
value of which corresponds to the desired RF input signal level applied to the
AD8317 RF input. A sample of the RF output signal from this variable-gain
component is selected, typically via a directional coupler, and applied to
AD8317 RF input. The voltage at Pin VOUT is applied to the gain control of
the variable gain element. A control voltage is applied to Pin VSET. The
magnitude of the control voltage can optionally be attenuated via the
voltage divider comprising R2 and R3, or a capacitor can be installed in
position R3 to form a low-pass filter along with R2.
C4, C5,
C3
Power Supply Decoupling.
C5 = 100 pF (Size 0402)
C4 = 0.1 ꢀF (Size 0603)
The nominal supply decoupling consists of a 100 pF filter capacitor placed
physically close to the AD8317 and a 0.1 ꢀF capacitor placed nearer to the
power supply input pin.
Filter Capacitor.
C3 = 8.2 pF (Size 0402)
The low-pass corner frequency of the circuit that drives Pin VOUT can be
lowered by placing a capacitor between CLPF and ground. Increasing this
capacitor increases the overall rise/fall time of the AD8317 for pulsed input
signals. See the Output Filtering section for more details.
Rev. 0 | Page 17 of 20
AD8317
VPOS
C4
TADJ
GND
R5
200Ω
0.1μF
VOUT_ALT
C5
R4
R7
OPEN
OPEN
100pF
C1
R6
V
OUT
1kΩ
47nF
CL
RL
8
7
6
5
OPEN OPEN
INLO VPOS
TADJ
VOUT
R2
R1
52.3Ω
AD8317
0Ω
INHI
1
COMM CLPF
VSET
4
2
3
RFIN
C2
C3
8.2pF
V
SET
47nF
R3
OPEN
Figure 36. Evaluation Board Schematic
Figure 37. Component Side Layout
Figure 38. Component Side Silkscreen
Rev. 0 | Page 18 of 20
AD8317
OUTLINE DIMENSIONS
1.89
1.74
1.59
3.25
3.00
2.75
0.55
0.40
0.30
0.60
0.45
0.30
5
4
8
*
2.25
2.00
1.75
BOTTOM VIEW
1.95
1.75
1.55
TOP VIEW
EXPOSED PAD
0.15
0.10
0.05
1
2.95
2.75
2.55
PIN 1
INDICATOR
0.25
0.20
0.15
0.50 BSC
12° MAX
0.80 MAX
0.65 TYP
1.00
0.85
0.80
0.05 MAX
0.02 NOM
0.30
0.23
0.18
SEATING
PLANE
0.20 REF
Figure 39. 8-Lead Lead Frame Chip Scale Package [LFCSP_VD]
2 mm x 3 mm Body, Very Thin, Dual Lead
(CP-8-1)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD8317ACPZ-R71
AD8317ACPZ-R21
AD8317ACPZ-WP1, 2
AD8317-EVAL
Temperature Range
Package Description
8-Lead LFCSP_VD
8-Lead LFCSP_VD
8-Lead LFCSP_VD
Evaluation Board
Package Option
CP-8-1
CP-8-1
Branding
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Q1
Q1
Q1
CP-8-1
1 Z = Pb-free part.
2 WP = waffle pack.
Rev. 0 | Page 19 of 20
AD8317
NOTES
©
2005 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D05541-0-10/05(0)
Rev. 0 | Page 20 of 20
相关型号：
©2020 ICPDF网 联系我们和版权申明