CN-0227 [ADI]
High Performance, 16-Bit, 250 MSPS Wideband Receiver with Antialiasing Filter; 高性能， 16位， 250 MSPS宽带接收机具有抗混叠滤波器
| 型号： | CN-0227 |
| 厂家： | 
ADI |
| 描述： | High Performance, 16-Bit, 250 MSPS Wideband Receiver with Antialiasing Filter |
| 文件： | 总5页 (文件大小：269K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
Circuit Note
CN-0227
Devices Connected/Referenced
Circuits from the Lab™ reference circuits are engineered and
tested for quick and easy system integration to help solve today’s
analog, mixed-signal, and RF design challenges. For more
information and/or support, visit www.analog.com/CN0227.
16-Bit, 250 MSPS Analog-to-Digital
Converter
AD9467
3.3 GHz Ultralow Distortion RF/IF
Differential Amplifier
ADL5562
High Performance, 16-Bit, 250 MSPS Wideband Receiver with Antialiasing Filter
EVALUATION AND DESIGN SUPPORT
CIRCUIT DESCRIPTION
Design and Integration Files
Schematics, Layout Files, Bill of Materials
The circuit accepts a single-ended input and converts it to
differential using a wide bandwidth (3 GHz) M/A-COM
ECT1-1-13M 1:1 transformer. The ADL5562 3.3 GHz differential
amplifier has a differential input impedance of 400 Ω when
operating at a gain of 6 dB and 200 Ω when operating at a gain
of 12 dB. A gain option of 15.5 dB is also available.
CIRCUIT FUNCTION AND BENEFITS
The circuit, shown in Figure 1, is a wideband receiver front end
based on the ADL5562 ultralow noise differential amplifier driver
and the AD9467 16-bit, 250 MSPS analog-to-digital converter.
The ADL5562 is an ideal driver for the AD9467, and the fully
differential architecture through the low-pass filter and into the
ADC provides good high frequency common-mode rejection,
as well as minimizes second-order distortion products. The
ADL5562 provides a gain of 6 dB or 12 dB depending on the input
connection. In the circuit, a gain of 6 dB was used to compensate
for the insertion loss of the filter network and transformer
(approximately 1.8 dB), providing an overall signal gain of 3.9 dB.
The third-order Butterworth antialiasing filter is optimized based
on the performance and interface requirements of the amplifier
and ADC. The total insertion loss due to the filter network and
other components is only 1.8 dB.
The overall circuit has a bandwidth of 152 MHz with a pass band
flatness of 1 dB. The SNR and SFDR measured with a 120 MHz
analog input are 72.6 dBFS and 82.2 dBc, respectively.
1.7dB LOSS
5.7dB GAIN
1.2dB LOSS
0.5dB LOSS
0.1dB LOSS
+5V
+3.3V
+1.8V
62pF
ANALOG INPUT
+6.0dBm FS AT 10MHz
XFMR
1:1 Z
ECT1-1-13M
0.1µF
0.1µF
0.1µF
22nH
20Ω
15Ω
AD9467
16-BIT
250MSPS
ADC
VIP2
VIP1
6Ω
243Ω
33Ω
FILTER
12pF
22nH
INPUT
Z = 50Ω
530Ω
3.5pF
ADL5562
G = 6dB
511Ω
Z = 400Ω
I
INTERNAL
INPUT Z
VIN1
VIN2
6Ω
243Ω
0.1µF
0.1µF
15Ω
20Ω
0.1µF
33Ω
62pF
OVERALL GAIN = 3.9dB
FS = 2V p-p DIFF
203Ω
38.6Ω
269Ω
Figure 1. 16-Bit, 250 MSPS Wideband Receiver Front End (Simplified Schematic: All Connections and Decoupling Not Shown)
Gains, Losses, and Signal Levels Measured Values at 10 MHz
Rev. A
Circuits from the Lab™ circuits from Analog Devices have been designed and built by Analog Devices
engineers. Standard engineering practices have been employed in the design and construction of
each circuit, and their function and performance have been tested and verified in a lab environment at
room temperature. However, you are solely responsible for testing the circuit and determining its
suitability and applicability for your use and application. Accordingly, in no event shall Analog Devices
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Fax: 781.461.3113 ©2011–2012 Analog Devices, Inc. All rights reserved.
CN-0227
Circuit Note
An input signal of 6.0 dBm produces a full-scale 2 V p-p
differential signal at the ADC input.
Table 1. Measured Performance of the Circuit
Performance Specs at 2 V p-p FS
Cutoff Frequency (−3 dB)
Pass-Band Flatness (6 MHz to 125 MHz)
SNRFS at 120 MHz
Final Results
152 MHz
1 dB
72.6 dBFS
82.2 dBc
The antialiasing filter is a third-order Butterworth filter designed
with a standard filter design program. A Butterworth filter was
chosen because of its flat response within the pass band. A third
order filter yields an ac noise bandwidth ratio of 1.05 and can be
designed with the aid of several free filter programs such as
Nuhertz Technologies Filter Free, or the Quite Universal Circuit
Simulator (Qucs) Free Simulation.
SFDR at 120 MHz
H2/H3 at 120 MHz
Overall Gain at 10 MHz
Input Drive at 10 MHz
86.6 dBc/82.2 dBc
3.9 dB
6.0 dBm
To achieve best performance, load the ADL5562 with a net
differential load of 200 Ω. The 15 Ω series resistors isolate the
filter capacitance from the amplifier output, and the 243 Ω resistors
in parallel with the downstream impedance yield a net load
impedance of 203 Ω when added to the 30 Ω series resistance.
0
–2
−3dB AT 152MHz
–4
–6
The 20 Ω resistors in series with the ADC inputs isolate internal
switching transients from the filter and the amplifier. The 511 Ω
resistor in parallel with the ADC serves to reduce the input
impedance of the ADC for more predictable performance.
–8
–10
–12
–14
–16
–18
–20
The third-order Butterworth filter was designed with a source
impedance of 38.6 Ω, a load impedance of 269 Ω, and a 3 dB
bandwidth of 180 MHz. The calculated values from the program
are shown in Figure 1. The values chosen for the filter passive
components were the closest standard values to those generated
by the program.
1
10
100
1000
ANALOG INPUT FREQUENCY (MHz)
23.83nH
19.3ꢀ
Figure 3. Pass Band Flatness Performance vs. Frequency
100
95
90
85
80
75
70
65
60
269ꢀ
12pF
35.79pF
19.3ꢀ
23.83nH
Figure 2. Design for Third-Order Differential Butterworth Filter with
ZS = 38.6 Ω, ZL = 269 Ω, FC = 180 MHz
SFDR (dBc)
The internal 3.5 pF capacitance of the ADC was subtracted
from the value of the second shunt capacitor to yield a value of
32.29 pF. In the circuit, this capacitor was realized using two
62 pF capacitors connected to ground as shown in Figure 1.
This provides the same filtering effect and offers some ac
common-mode rejection.
SNRFS (dBFS)
The measured performance of the system is summarized in
Table 1, where the 3 dB bandwidth is 152 MHz. The total insertion
loss of the network is approximately 2 dB. The bandwidth response
is shown in Figure 3; the SNR and SFDR performance are
shown in Figure 4.
0
20
40
60
80
100
120
140
160
180
ANALOG INPUT FREQUENCY (MHz)
Figure 4. SNR/SFDR Performance vs. Frequency
Rev. A | Page 2 of 5
Circuit Note
CN-0227
C
AAF2
0.1µF
R
L
0.1µF
XFMR
1:1 Z
ECT1-1-13M
R
AAF
R
A
KB
0.1µF
ANALOG INPUT
ADC
INTERNAL
INPUT Z
Z
Z
/2
TAMP
O
FILTER
AAF1
R
T
R
R
C
R
I
ADC
ADC
GAIN
/2
TADC
INPUT
Z = 50ꢀ
C
R
O
TAMP
0.1µF
L
AAF
0.1µF
Z = 2R ║ R
T
I
R
R
KB
A
0.1µF
C
R
AAF2
T
Z
Z
Z
AAFL
AL
AAFS
Z
Z
Z
= R
║ (R
+ 2R
)
AAFL
TADC
ADC
KB
= 2R + (Z
║ 2R
)
AL
A
AAFL
TAMP
= 2R
║ (Z + 2R )
O A
AAFS
TAMP
Figure 5. Generalized Differential Amplifier/ADC Interface with Low-Pass Filter
Filter and Interface Design Procedure
5. Select RTAMP so that the total load seen by the amplifier, ZAL,
is optimum for the particular differential amplifier chosen
using the equation:
In this section, a general approach to the design of the amplifier/
ADC interface with filter is presented. To achieve optimum
performance (bandwidth, SNR, and SFDR), there are certain
design constraints placed on the general circuit by the amplifier
and the ADC:
Z
AL = 2RA + (ZAAFL || 2RTAMP).
6. Calculate the filter source resistance:
AAFS = 2RTAMP || (ZO + 2RA).
7. Using a filter design program or tables design the filter
Z
1. The amplifier should see the correct dc load recommended
by the data sheet for optimum performance.
2. The correct amount of series resistance must be used between
the amplifier and the load presented by the filter. This is to
prevent undesired peaking in the pass band.
3. The input to the ADC should be reduced by an external
parallel resistor, and the correct series resistance should be
used to isolate the ADC from the filter. This series resistor
also reduces peaking.
using the source and load impedances, ZAAFS and ZAAFL
type of filter, bandwidth, and order. Use a bandwidth that
is about 40% higher than one-half the sampling rate to
ensure flatness in the frequency span between dc and fs/2.
8. The internal ADC capacitance, CADC, should be subtracted
from the final shunt capacitor value generated by the
program. The program will give the value CSHUNT2 for the
differential shunt capacitor. The final common-mode
shunt capacitance is
,
The generalized circuit shown in Figure 5 applies to most high
speed differential amplifier/ADC interfaces and will be used as
a basis for the discussion. This design approach will tend to
minimize the insertion loss of the filter by taking advantage of
the relatively high input impedance of most high speed ADCs
and the relatively low impedance of the driving source (amplifier).
CAAF2 = 2(CSHUNT2 − CADC).
After running these preliminary calculations, the circuit should
be given a quick review for the following items.
1. The value of CAAF2 should be at least 10 pF so that it is several
times larger than CADC. This minimizes the sensitivity of
The basic design process is as follows:
1. Select the external ADC termination resistor RTADC so that
the parallel combination of RTADC and RADC is between 200 Ω
and 400 Ω.
2. Select RKB based on experience and/or the ADC data sheet
recommendations, typically between 5 Ω and 36 Ω.
3. Calculate the filter load impedance using:
the filter to variations in CADC
.
2. The ratio of ZAAFL to ZAAFS should not be more than about 7 so
that the filter is within the limits of most filter tables and
design programs.
3. The value of CAAF1 should be at least 5 pF to minimize
sensitivity to parasitic capacitance and component
variations.
Z
AAFL = RTADC || (RADC + 2RKB)
4. The inductor, LAAF, should be a reasonable value of at least
several nH.
4. Select the amplifier external series resistor RA. Make RA less
than 10 Ω if the amplifier differential output impedance is
100 Ω to 200 Ω. Make RA between 5 Ω and 36 Ω if the output
impedance of the amplifier is 12 Ω or less.
Rev. A | Page 3 of 5
CN-0227
Circuit Note
In some cases, the filter design program may provide more than
one unique solution, especially with higher order filters. The
solution that uses the most reasonable set of component values
should always be chosen. Also, choose a configuration that ends
in a shunt capacitor so that it can be combined with the ADC
input capacitance.
Balancing these trade-offs can be somewhat difficult. In this
design, each parameter was given equal weight; therefore, the
values chosen are representative of the interface performance
for all the design characteristics. In some designs, different values
may be chosen to optimize SFDR, SNR, or input drive level,
depending on system requirements.
Circuit Optimization Techniques and Trade-Offs
The SFDR performance in this design is determined by two
factors: the amplifier and ADC interface component values as
shown in Figure 1, and the setting of the internal front-end
buffer bias current in the AD9467 via an internal register. The
final SFDR performance numbers shown in Table 1 and Figure 4
were obtained after following the SFDR optimization described
in the AD9467 data sheet.
The parameters in this interface circuit are very interactive,
therefore, it is almost impossible to optimize the circuit for all
key specifications (bandwidth, bandwidth flatness, SNR, SFDR,
and gain). However, the peaking, which often occurs in the
bandwidth response, can be minimized by varying RA and RKB.
Notice in Figure 6 how the pass-band peaking is reduced as the
value of the output series resistance, RA, is increased. However,
as the value of this resistance increases, there is more signal
attenuation, and the amplifier must drive a larger signal to fill
the ADCs full-scale input range.
Another trade-off that can be made in this particular design is
the ADC full-scale setting. The full-scale ADC differential input
voltage was set for 2 V p-p for the data obtained with this design,
which optimizes SFDR. Changing the full-scale input range to
2.5 V p-p yields about 1.5 dB improvement in SNR but slightly
degrades the SFDR performance. The input range is set by the
value loaded into an internal register in the AD9467 as described in
the data sheet.
The value of RA also affects SNR performance. Larger values,
while reducing the bandwidth peaking, tend to slightly increase
the SNR because of the higher signal level required to drive the
ADC full scale.
Note that the signal in this design is ac coupled with the 0.1 μF
capacitors to block the common-mode voltages between the
amplifier, its termination resistors, and the ADC inputs. Refer to
the AD9467 data sheet for further details regarding common-
mode voltages.
Select the RKB series resistor on the ADC inputs to minimize
distortion caused by any residual charge injection from the
internal sampling capacitor within the ADC. Increasing this
resistor also tends to reduce bandwidth peaking.
However, increasing RKB increases signal attenuation, and the
amplifier must drive a larger signal to fill the ADC input range.
Passive Component and PC Board Parasitic
Considerations
Another method for optimizing the pass-band flatness is to vary
the filter shunt capacitor, CAAF2, by a small amount.
The performance of this or any high speed circuit is highly
dependent on proper PCB layout. This includes, but is not
limited to, power supply bypassing, controlled impedance lines
(where required), component placement, signal routing, and
power and ground planes. See Tutorials MT-031 and MT-101 for
more detailed information regarding PCB layout for high speed
ADCs and amplifiers.
Normally, the ADC input termination resistor, RTADC, is selected
to make the net ADC input impedance between 200 Ω and 400 Ω.
Making it lower reduces the effect of the ADC input capacitance
and may stabilize the filter design; however, increases the insertion
loss of the circuit. Increasing the value also reduces peaking.
5
Use low parasitic surface-mount capacitors, inductors, and
resistors for the passive components in the filter. The inductors
chosen are from the Coilcraft 0603CS series. The surface mount
capacitors used in the filter are 5%, C0G, 0402-type for stability
and accuracy.
10ꢀ
0
15ꢀ
30ꢀ
–5
See the CN-0227 Design Support Package for complete
documentation on the system.
–10
–15
–20
1
10
100
1000
ANALOG INPUT FREQUENCY (MHz)
Figure 6. Pass-Band Flatness Performance vs. Amplifier Output Series
Resistance, RA
Rev. A | Page 4 of 5
Circuit Note
CN-0227
COMMON VARIATIONS
LEARN MORE
For applications that require less bandwidth and lower power,
the ADL5561 differential amplifier can be used. The ADL5561
has a bandwidth of 2.9 GHz and only uses 40 mA of current.
For even lower power and bandwidth, the ADA4950-1 can also
be used. This device has a 1 GHz bandwidth and only uses 10 mA
of current. For higher bandwidth, the 6 GHz ADL5565 differential
amplifier is pin compatible with the others previously listed.
CN-0227 Design Support Package:
http://www.analog.com/CN0227-DesignSupport
UG-200: Evaluating the AD9467 16-Bit, 200 MSPS/250 MSPS ADC
Arrants, Alex, Brad Brannon and Rob Reeder, AN-835
Application Note, Understanding High Speed ADC Testing
and Evaluation, Analog Devices.
Ardizzoni, John. A Practical Guide to High-Speed Printed-
Circuit-Board Layout, Analog Dialogue 39-09, September 2005.
CIRCUIT EVALUATION AND TEST
This circuit uses a modified AD9467-250EBZ circuit board and
the HSC-ADC-EVALCZ FPGA-based data capture board. The
two boards have mating high speed connectors, allowing for the
quick setup and evaluation of the circuit’s performance. The
modified AD9467-250EBZ board contains the circuit evaluated
as described in this note, and the HSC-ADC-EVALCZ data
capture board is used in conjunction with Visual Analog evaluation
software, as well as the SPI Controller software to properly control
the ADC and capture the data. See User Guide UG-200 for the
schematics, BOM, and layout for the AD9467-250EBZ board.
The readme.txt file in the CN-0227 Design Support Package
describes the modifications made to the standard AD9467-250EBZ
board. Application Note AN-835 contains complete details on
how to set up the hardware and software to run the tests described
in this circuit note.
MT-031 Tutorial, Grounding Data Converters and Solving the
Mystery of “AGND” and “DGND”, Analog Devices.
MT-101 Tutorial, Decoupling Techniques, Analog Devices.
Quite Universal Circuit Simulator
Nuhertz Technologies, Filter Free Filter Design Program
Reeder, Rob, Achieve CM Convergence between Amps and ADCs,
Electronic Design, July 2010.
Reeder, Rob, Mine These High-Speed ADC Layout Nuggets For
Design Gold, Electronic Design, September 15, 2011.
Rarely Asked Questions: Considerations of High-Speed
Converter PCB Design, Part 1: Power and Ground Planes,
November 2010.
Rarely Asked Questions: Considerations of High-Speed
Converter PCB Design, Part 2: Using Power and Ground
Planes to Your Advantage, February 2011.
Rarely Asked Questions: Considerations of High-Speed Converter
PCB Design, Part 3: The E-Pad Low Down, June 2011.
Data Sheets and Evaluation Boards
AD9467 Data Sheet
ADL5562 Data Sheet
Circuit Evaluation Board (AD9467-250EBZ)
Standard Data Capture Platform (HSC-ADC-EVALCZ)
REVISION HISTORY
2/12—Rev. 0 to Rev. A
Changes to Figure 1 ..........................................................................1
11/11—Revision 0: Initial Version
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©2011–2012 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
CN10071-0-2/12(A)
Rev. A | Page 5 of 5
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