LT1319CS_技术文档

LT1319

Multiple Modulation Standard

Infrared Receiver

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DESCRIPTIO

EATURE

S

F

The LT^®1319 is a general purpose building block that

contains all the circuitry necessary to transform modu-

lated photodiode signals back to digital signals. The

circuit’sflexibilitypermitsittoreceivemultiplemodulation

methods. A low noise, high frequency preamplifier per-

forms a current-to-voltage conversion while rejecting low

frequency ambient interference with an AC coupling loop.

Two separate high impedance filter buffer inputs are

provided so that off-chip filtering can be tailored for

specificmodulationschemes.Thefilterbuffersdrivesepa-

rate differential gain stages that end in comparators with

internalhysteresis. Thecomparatorthresholdsareadjust-

able externally by the current into Pin 11. One channel has

ahighspeed25nscomparatorrequiredforhighdatarates.

The second channel’s comparator has a 60ns response

time and is well suited to more modest data rates. A power

savingshutdownfeatureisusefulinportableapplications.

■

Receives Multiple IR Modulation Methods

Low Noise, High Speed Preamp: 2pA/√Hz, 7MHz

Low Frequency Ambient Rejection Loops

Dual Gain Channels: 8MHz, 400V/V

25ns and 60ns Comparators

16-Lead SO Package

5V Single Supply Operation

Supply Current: 14mA

Shutdown Supply Current: 500µA

External Comparator Threshold Setting

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ODULATIO STA DARDS

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IRDA: SIR, FIR

Sharp/Newton

TV Remote

■

High Data Rate Modulation Methods

For IRDA 4PPM contact the LTC Marketing Department.

, LTC and LT are registered trademarks of Linear Technology Corporation.

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TYPICAL APPLICATION

IRDA and Sharp/Newton Data Receiver

SHUTDOWN INPUT

IRDA DATA

SHARP/NEWTON DATA

V

CC

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

AN_GND

BYPASS

IN

LT1319

V

CC

FILT1

PREOUT

C

B3

10µF

SHDN

DATAL

C

R

B1

0.1µF

B2

T1

10µF

30k

V

DIG_GND

BIAS

L

F1

R

F1

1k

F2

FILTINL

FILT2L

FILTIN

V

TH

D1*

100µH

2k

C

T1

1µF

DATA

FILT2

C

B4

1µF

C

F5

1µF

C

F3

F2

F4

F1

100pF

1nF

2.2nF

10nF

*BPW34FA OR BPV22NF

DGND

AGND

LT1319 • TA01

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LT1319

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/O

ABSOLUTE AXI U RATI GS

PACKAGE RDER I FOR ATIO

Total Supply Voltage (V_CCto GND) ........................... 6V

Differential Voltage (Any Two Pins) .......................... 6V

Maximum Junction Temperature ......................... 150°C

Operating Temperature Range .................... 0°C to 70°C

Specified Temperature Range..................... 0°C to 70°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

TOP VIEW

ORDER PART

AN_GND

IN

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

BYPASS

NUMBER

V

CC

FILT1

SHDN

LT1319CS

PREOUT

DATAL

DIG_GND

V

BIAS

FILTINL

FILT2L

FILTIN

V

TH

DATA

FILT2

S PACKAGE

16-LEAD PLASTIC SO

T_JMAX= 150°C, θ_JA= 100°C/W

Consult factory for Industrial or Military grade parts.

ELECTRICAL CHARACTERISTICS

T_A= 25°C, V₁₅= 5V, V₁= V₁₂= 0V, V₆= V₈= V₁₄= 2V, unless otherwise specified.

SYMBOL PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

V

OS

Preamp Input Offset Voltage

Preamp Output Offset Voltage

Preamp Loop Offset Voltage

High Gain Loop Offset Voltage

Low Gain Loop Offset Voltage

V (Pin 2) – V (Pin 5)

V (Pin 4) – V (Pin 5)

V (Pin 3) – V (Pin 5)

V (Pin 9) – V (Pin 5)

V (Pin 7) – V (Pin 5)

4

15

25

250

950

mV

10

50

600

150

800

A

VP

Preamp Transimpedance

±10µA Into Pin 2, Measure ∆V (Pin 4), Fix Pin 3

10

15

17

kΩ

Preamp Output Swing, Positive

Preamp Output Swing, Negative

100µA Out of Pin 2, Measure ∆V (Pin 4), Fix Pin 3

100µA Into Pin 2, Measure ∆V (Pin 4), Fix Pin 3

0.25

–0.55

0.4

–0.4

0.55

–0.25

V

BW

Preamp Bandwidth

C (Pin 3) = 1µF, Measure f

C (Pin 3) = 1µF, f = 10kHz

7

2

MHz

P

–3dB

i

n

Preamp Input Noise Current

pA/√Hz

Preamp Loop Rejection, Positive

Preamp Loop Rejection, Negative

50µA Into Pin 2, Measure ∆V (Pin 4)

50µA Out of Pin 2, Measure ∆V (Pin 4)

–3

–1

1

3

mV

Preamp Loop Output Current, Positive

Preamp Loop Output Current, Negative

100µA Out of Pin 2, Measure I (Pin 3), (Note 1)

100µA Into Pin 2, Measure I (Pin 3), (Note 1)

–150

50

–100

100

–50

150

µA

V

Bias Voltage

V (Pin 5)

1.7

4.75

0.1

1.9

4.9

0.5

40

2.1

4.95

1.4

V

µA

BIAS

Bypass Voltage

Filter Buffer Input Bias Current

Filter Buffer Input Resistance

V (Pin 16)

I (Pin 6), I (Pin 8)

BYPASS

I

B

R

∆V = 0.1V, Measure ∆I Pin 6, Pin 8

MΩ

IN

B

Gain Stage Loop Rejection, Positive

Gain Stage Loop Rejection, Negative

∆V = 50mV (Pin 6, Pin 8), Measure ∆V (Pin 7, Pin 9)

0.33

0.45

–0.45

0.57

–0.33

V

∆V = –50mV (Pin 6, Pin 8), Measure ∆V (Pin 7, Pin 9) –0.57

A

Gain Stages Voltage Gain

Gain Stages Bandwidth

(Note 2)

400

8

V/V

VG

BW

C (Pin 7) = C (Pin 9) = 1µF

MHz

G

t

Fast Comparator Response Time

Slow Comparator Response Time

10mV Overdrive

25

60

ns

r

V

Fast Comparator Hysteresis Voltage

Slow Comparator Hysteresis Voltage

(Note 3)

35

40

mV

HYS

Fast Comparator Output High Voltage

Slow Comparator Output High Voltage

∆V (Pin 9) = –200mV, 1mA Out of Pin 10 (Note 4)

∆V (Pin 7) = –200mV, 0.1mA Out of Pin 13 (Note 4)

2.4

3.5

3.9

V

OH

Fast Comparator Output Low Voltage

Slow Comparator Output Low Voltage

∆V (Pin 9) = 200mV, 800µA Into Pin 10

∆V (Pin 7) = 200mV, 800µA Into Pin 13

0.35

0.39

0.5

V

OL

2

LT1319

ELECTRICAL CHARACTERISTICS

T_A= 25°C, V₁₅= 5V, V₁= V₁₂= 0V, V₆= V₈= V₁₄= 2V, unless otherwise specified.

SYMBOL PARAMETER

Threshold Transimpedance

CONDITIONS

100µA Into Pin 11 (Note 5)

100µA Into Pin 11, V (Pin 11)

MIN

TYP

2

MAX

UNITS

kΩ

V

Threshold External Voltage

Shutdown Input High Voltage

Shutdown Input Low Voltage

Shutdown Input High Current

Shutdown Input Low Current

Supply Current

0.8

2

0.9

1.2

TH

IH

IL

V

0.8

–10

–130

18

I

V (Pin 14) = 2.4V

V (Pin 14) = 0.4V

–140

–400

10

–60

–260

14

µA

mA

µA

IH

IL

V (Pin 14) = 2V

S

Supply Current in Shutdown

V (Pin 14) = 0.8V, V (Pin 6) = V (Pin 8) = 0V

300

500

800

SHDN

0°C ≤ T_A≤ 70°C, V₁₅= 5V, V₁= V₁₂= 0V, V₆= V₈= V₁₄= 2V, unless otherwise specified.

SYMBOL PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

V

OS

Preamp Input Offset Voltage

Preamp Output Offset Voltage

Preamp Loop Offset Voltage

High Gain Loop Offset Voltage

Low Gain Loop Offset Voltage

V (Pin 2) – V (Pin 5)

V (Pin 4) – V (Pin 5)

V (Pin 3) – V (Pin 5)

V (Pin 9) – V (Pin 5)

V (Pin 7) – V (Pin 5)

4

17

27

350

1200

mV

10

30

400

150

800

A

VP

Preamp Transimpedance

±10µA Into Pin 2, Measure ∆V (Pin 4)

8.5

15

18.5

kΩ

Preamp Output Swing, Positive

Preamp Output Swing, Negative

100µA Out of Pin 2, Measure ∆V (Pin 4)

100µA Into Pin 2, Measure ∆V (Pin 4)

0.2

–0.6

0.4

–0.4

0.6

–0.2

V

Preamp Loop Rejection, Positive

Preamp Loop Rejection, Negative

50µA Into Pin 2, Measure ∆V (Pin 4)

50µA Out of Pin 2, Measure ∆V (Pin 4)

–3.5

–1

1

3.5

mV

Preamp Loop Output Current, Positive

Preamp Loop Output Current, Negative

100µA Out of Pin 2, Measure I (Pin 3), (Note 1)

100µA Into Pin 2, Measure I (Pin 3), (Note 1)

–160

40

–100

100

–40

160

µA

V

Bias Voltage

Bypass Voltage

V (Pin 5)

V (Pin 16)

1.5

4.7

1.9

4.9

0.5

0.45

–0.45

2.3

4.97

1.6

0.6

–0.3

V

BIAS

BYPASS

I

Filter Buffer Input Bias Current

Gain Stage Loop Rejection, Positive

Gain Stage Loop Rejection, Negative

I (Pin 6), I (Pin 8)

∆V = 50mV (Pin 6, Pin 8), Measure ∆V (Pin 7, Pin 9)

∆V = –50mV (Pin 6, Pin 8), Measure ∆V (Pin 7, Pin 9) –0.6

0.05

0.3

µA

V

B

V

Fast Comparator Output High Voltage

Slow Comparator Output High Voltage

∆V (Pin 9) = –200mV, 1mA Out of Pin 10 (Note 4)

∆V (Pin 7) = –200mV, 0.1mA Out of Pin 13 (Note 4)

2.4

3.5

3.9

V

OH

Fast Comparator Output Low Voltage

Slow Comparator Output Low Voltage

∆V (Pin 9) = 200mV, 800µA Into Pin 10

∆V (Pin 7) = 200mV, 800µA Into Pin 13

0.35

0.39

0.5

V

OL

V

Threshold External Voltage

Shutdown Input High Voltage

Shutdown Input Low Voltage

Shutdown Input High Current

Shutdown Input Low Current

Supply Current

100µA Into Pin 11, V (Pin 11)

0.7

2

0.9

1.3

V

TH

IH

IL

0.8

0

V

I

V (Pin 14) = 2.4V

–160

–450

9

–60

–260

14

µA

mA

µA

IH

V (Pin 14) = 0.4V

V (Pin 14) = 2V

–80

20

IL

S

Supply Current in Shutdown

V (Pin 14) = 0.8V, V (Pin 6) = V (Pin 8) = 0V

200

500

900

SHDN

Note 1: Measure V (Pin 3) without input current for Pin 2. Force Pin 3 to

this measured voltage (which disables the preamp loop). Measure the

current into and out of Pin 3 when Pin 2 is driven.

Note 4: Measure V (Pin 7) and V (Pin 9). Force these voltages to 200mV

below their nominal value to switch the comparators high.

Note 5: The current into Pin 11 is multiplied by 4 and then applied to a

500Ω resistor on the positive comparator inputs. The threshold is

I (Pin 11) • 4 • 500Ω.

Note 2: The gain is the differential voltage at the comparator inputs divided

by the differential voltage between the filter buffer output and V

parameter is not tested.

. This

BIAS

Note 3: Hysteresis is the difference in comparator trip point measured

when the output is high and when the output is low. This parameter is

not tested.

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LT1319

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TYPICAL PERFORMANCE CHARACTERISTICS

Preamp Highpass vs

Gain Stage Highpass vs

Capacitance on FILT2 or FILT2L

Preamp Frequency Response vs

Capacitance on FILT1

Input Capacitance

5

4

1000

100

10

1000

100

10

T

= 25°C

T = 25°C

A

T

= 25°C

A

3

2

50pF

10pF

1

0

–1

–2

–3

–4

–5

30pF

10M

1

0.1

100

1

100k

1M

100M

1k

10k

100k

1M

1k

10k

100k

1M

10M

FREQUENCY (Hz)

HIGHPASS CORNER FREQUENCY (Hz)

1319 G01

1319 G02

1319 G03

Input-Referred Noise vs

Lowpass Filter on PREOUT

FILTIN- or FILTINL- Referred

Threshold Voltage vs R_T1

Preamp Output Noise vs

Input Capacitance

20

15

10

5

1.0

0.9

0.8

0.7

0.6

0.5

250

200

150

100

50

T

= 25°C

T

= 25°C

A

T = 25°C

A

FILTER

A

R

= 1k

50pF

30pF

10pF

0

0.1

1

10

20

30

(kΩ)

40

10k

100k

1M

FREQUENCY (Hz)

10M

100M

FILTER CUTOFF FREQUENCY (MHz)

R

T1

1319 G04

1319 G05

1319 G06

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CIRCUIT DESCRIPTIO

The LT1319 is a general purpose low noise, high speed,

high gain, infrared receiver designed to easily provide IR

communications with portable computers, PDAs, desktop

computers and peripherals. The receiver takes the photo-

current from an infrared photodiode (Siemens BPW34FA

or Temic BPV22NF) and performs a current-to-voltage

conversion. After external filtering that is tailored for the

desired communication standard, two filter buffers are

provided. There are dual gain chains with nominal gain of

400V/Vthatfeedinternalcomparatorswithhysteresis. The

comparatorthresholdsaresetexternallywithacurrentinto

theV_THpin.Thehighfrequencycomparatorhasaresponse

time of 25ns and is well-suited to high data rates.

The low frequency comparator responds in 60ns and is

useful for more modest data rates such as Sharp/Newton

andIRDA-SIR.Thecircuitalsocontainsshutdowncircuitry

to reduce power consumption. Rejection of ambient inter-

ferenceisaccomplishedwithACcouplingloopsaroundthe

preamp and the two gain stages. The rejection frequency is

set with an internal resistor and an external capacitor to

ground. This feature allows changing of the break fre-

quencybysimplyswitchinginadditionalcapacitors. Toaid

in rejection of power supply noise there is internal supply

regulation and a fully differential topology after the filter

buffers.

4

LT1319

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BLOCK DIAGRA

DS2

DS1

AN_GND

1

BYPASS

16

+

C

B3

10µF

R

S3

20k

S5

20k

Q3

R

S2

FB

S4

V

5V

CC

15k

20k

R

S6

Q2

15

1k

+

0.1µF

+

IPD

C

IN

2

C

B1

B2

Q4

V

Q1

–

10µF

R

S1

SHDN

14

PHOTO-

DIODE

20k

PREAMP

R

+

L1

REG

R

H1

10k

50k

R

C1

G1

+

V

BIAS

A1

1k

500Ω

FILTER

BUFFER

–

+

–

+

1

DATAL

13

GM1

A2

= 20

A3

= 20

R

C2

500Ω

G2

FILT1

3

COMP 1

+

A

–

A

V

–

g

1k

V

m

4k

+

–

C

F1

LOW FREQUENCY

COMPARATOR

10nF

V

BIAS

R

L2

10k

+

g

m

+

DIG_GND

12

1

PREOUT

4

4k

–

GM2

V

BIAS

5

R

F2

F1

2k

1k

5V

+

C

R

B4

T1

SC

V

TH

L

1µF

30k

F1

2k

V

TH

100µH

11

GEN

+

C

T1

FILTINL

1µF

6

R

H2

50k

+

C

F3

FILT2L

7

100pF

R

C3

G3

+

A4

1k

500Ω

+

C

FILTER

BUFFER

–

F4

+

–

DATA

10

2.2nF

A5

= 20

A6

= 20

R

C4

500Ω

G4

COMP 2

A

–

A

V

–

1k

V

FILTIN

8

HIGH FREQUENCY

COMPARATOR

V

GM3

+

BIAS

C

F2

R

L3

10k

1nF

+

g

m

+

1

4k

–

FILT2

9

+

C

F5

1µF

NOTE: EXTERNAL COMPONENTS ARE SHOWN FOR AN IRDA AND SHARP/NEWTON DATA RECEIVER.

LT1319 • BD

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LT1319

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APPLICATIONS INFORMATION

ing the voltage noise gain. Referring to the Block Diagram,

at frequencies beyond the corner frequency of the AC

coupling loop, the preamp is in a noise gain of 2.5 due to

the ratio of (R_FB+ R_L1)/R_L1. At high frequencies the input

capacitanceapproachesthesameimpedanceasR_L1sothe

noise gain increases. For example, at 500kHz the 30pF

input capacitance looks like 10.6kΩ which increases the

noise gain to almost 4. The preamp is compensated to

provideaflatcurrent-to-voltagefrequencyresponsewitha

–3dB corner at 7MHz. The input current noise peaks up

considerably if full bandwidth is used. To obtain best noise

performance,theoutputofthepreampshouldbefilteredto

the minimum bandwidth required for the desired modula-

tion scheme. The graph of input-referred noise versus

lowpass filtering on the preamp output shows the noise

penalty for higher bandwidths.

Layout and Passive Components

TheLT1319requirescarefullayouttechniquestominimize

parasitic signal coupling to the preamp input. A sample

board layout for the circuit on the first page is shown in the

Typical Application section. The lead lengths on the photo-

diode must be as short as possible to Pin 2. Shielding is

recommended over the entire circuit. A ground plane must

be used and connected to Pin 1. The ground plane should

extendunderthepackageandsurroundPins1to9andPin

16. A single point connection should be made to the

groundplaneatPin12(DIG_GND).TheleadsonPins6and

8 should be short to prevent pickup into the gain stages.

The comparator output leads (Pins 10 and 13) should be

as short as possible to minimize coupling back to the input

via parasitic capacitance.

Capacitance on Pin 10 should be minimized as the com-

parator output is pulled up by an internal 5k resistor. The

associated digital circuitry should be located on the oppo-

site side of the PC board from the LT1319 or separated as

much as possible if on the same side of the board. Filter

components should be located on the analog ground side

of the package. Bypass capacitors should be used on Pins

5, 11, 15 and 16 for best supply rejection.

AC Coupling Loops

There are three AC loops in the circuit that reject low

frequency inputs. The first loop is around the preamp and

provides rejection of ambient light sources. The operation

can be explained by looking at the Block Diagram. For low

frequency signals the transconductance amplifier, GM1,

compares the preamp output to the V_BIASvoltage. This

differential voltage is transformed into a current that is fed

into the high impedance node at Pin 3 and transformed

back to a voltage. There is a voltage gain of approximately

60dB to this point which is then buffered to drive a 10k

resistor that is connected back to the input of the preamp.

This high gain loop attenuates the effect of low frequency

signalsbytheamountoftheloopgaintimestheratioofR_L1

to R_FB(i.e., 1000V/V • 15/10 = 1500). For higher frequen-

ciestheattenuationdecreasesduetotheexternalcapacitor

onPin3. Atfrequenciesbeyondwheretheloopgainequals

10/15, signals are no longer attenuated. This high fre-

quency cutoff is at:

Preamp

The LT1319 preamp is a low noise, high speed current-to-

voltage converter that has been optimized for an input

capacitance of 30pF (which corresponds to the capaci-

tance of the above-mentioned photodiodes with approxi-

mately 2V of back bias). A range of 0pF to 50pF is

acceptable. The amplifier obtains high bandwidth by pro-

vidingalowimpedanceinputsothattheinputcurrentisnot

filtered by the photodiode capacitance.

The dynamic range of the circuit will be limited at the low

end by the input-referred current noise of the preamplifier

and the desired signal-to-noise ratio. At the other extreme

ofthedynamicrangeforverylargeinputsignals,theoutput

of the preamp is clamped by Schottky diodes across the

feedback resistor.

f = (15/10)/(2π • 4kΩ • C_PIN3

)

where 1/(4kΩ) is the transconductance of the loop ampli-

fier. For example, if C_PIN3= 300pF, the highpass frequency

is 200kHz which can aid in rejection of a wide range of

ambient interference.

The noise bandwidth is shaped by filtering at the output of

the preamplifier and by the AC coupling loop. The input

capacitance causes noise peaking for high bandwidth

applications. Noise peaking can be explained by consider-

The other two loops operate similarly around the gain

stages and also provide low frequency rejection. In addi-

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LT1319

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APPLICATIONS INFORMATION

tion, the loops around the gain stages provide an accurate

DC threshold setting for the comparators. At DC, the loops

force the differential voltages at the output of the gain

stages to zero. The comparator threshold is set by the

currents provided by the V_THgenerator through the 500Ω

resistors R_C1and R_C3. These currents are equal to 4 times

thecurrentintoPin11.For100µAintoPin11,thecompara-

tor thresholds are nominally 200mV.

pass filter network. The application on the first page of the

datasheetisrepeatedintheBlockDiagramandcanbeused

to illustrate the filtering for IRDA-SIR and Sharp/Newton.

ThepreamphighpasszeroissetbyGM1andC_F1.Thebreak

frequency is located at:

f = (15kΩ/10kΩ)/(2π • 4kΩ • 10nF) = 6kHz

Onthelowspeedchannelthereisalowpassfilterat800kHz

set by R_F2and C_F3. The gain stage has a highpass filter set

by GM2 and C_F4at approximately 500kHz. The high speed

channel has an LC tank circuit at 500kHz with Q = 3 set by

R_F1. The high speed gain stage has a highpass character-

istic set by GM3 and C_F5with a break frequency of 1.1kHz.

These filters are suitable for the 1.6µs pulses and up to

115kBd data rates of IRDA-SIR on the slow channel. The

fast channel is used for Sharp/Newton ASK Modulation

with 500kHz bursts at data rates up to 38.4kBd.

Power Supply Rejection and Biasing

TheLT1319hasveryhighgainandbandwidthsogreatcare

is taken to reduce false output transitions due to power

supplynoise. AsafirststeptheV_CCinputisregulateddown

to approximately 4V to power all the analog sections of the

circuitwhicharealsotiedtoAnalogGround(Pin1)asisthe

substrate of the die. Additionally, the internal 4V is by-

passedatPin16. Thedigitalcircuitry(thecomparatorsand

shutdown logic) is powered directly off of V_CCand is

returned to Digital Ground (Pin 12). To provide a clean bias

point for the preamp, filter buffers and the gain stages, a

1.9V reference is generated from the 4V rail and is by-

passed at Pin 5. The gain stages are pure differential

designs which inherently reject supply variations.

A second circuit is shown in the Typical Applications

section for IRDA SIR/FIR and Sharp. This circuit is Demo

Board 54. The first filter is a preamp highpass loop set at

600Hz by C_F7for IRDA or 180kHz by C_F1for Sharp. Sharp

modulation is run on the low speed channel and is next

filtered by a tank circuit formed by R_F2, L_F1and C_F3and

centered at 500kHz. L_F1also provides the DC bias for the

filter buffer input. A final highpass for the lower speed

channel is set by C_F4at 130kHz. The high speed channel is

usedbyIRDASIRandFIRwhichuse1.6µsand220nswide

pulses. A lowpass formed by R_F1and C_F2limit the noise

bandwidth. The final highpass is set by C_F5(2.5MHz for

FIR)orC_F6(450kHz).ThesquelchcircuitformedbyQ1,Q2,

Q3 and R_C1to R_C6extends the short range performance

and will be discussed later.

Filtering

Filtering is needed for two main reasons: sensitivity and

ambient rejection. Lowpass filtering is needed to limit the

bandwidth in order to minimize the noise. Low noise

permits reliable detection of smaller input signals over a

larger distance. Highpass filtering is used to reject interfer-

ing ambient signals. Interference includes low frequency

sources of infrared light such as sunlight, incandescent

lights, and ordinary fluorescent lights, as well as high

frequencysourcessuchasTVremotecontrols(40kHz)and

high frequency fluorescent lighting (40kHz to 80kHz).

In designing custom filters for different applications, the

following guidelines should be used.

1. Limit the noise bandwidth with a lowpass filter that has

a rise time equal to half the pulse width. For example, for

1µs pulses a 700kHz lowpass filter has a 10% to 90%

rise time of 0.35/700kHz = 500ns.

Thecircuittopologyallowsforfilteringbetweenthepream-

plifier and the filter buffers as well as filtering with the three

internalhighpassloops.Withtwochannelsthefilteringcan

be optimized for different modulation schemes. The high

speed channel (with a 25ns comparator) is ideal for modu-

lation schemes using frequencies above 1MHz. Carrier-

based methods as well as narrow pulse schemes can have

superior ambient rejection by adding in a dedicated high-

2. Limit the maximum highpass to 1/(4 • pulse width). For

1µs pulses, 1/4µs = 250kHz.

3. In setting the highpass filters, space the filters apart by

a factor of 5 to 10 to reduce overshoot due to filter

7

LT1319

U

W U U

APPLICATIONS INFORMATION

interaction. Overshoot becomes especially important

for high input levels because it can cause false pulses

which may not be tolerated in certain modulation

schemes. It is also more of a problem in modulation

schemes such as IRDA-SIR and FIR where the duty

cycle can get very low (i.e., transmitting data with lots of

ones which are signaled with the absence of pulses). AC

coupled receivers when faced with low duty cycle data

set their thresholds close to the baseline DC level of the

data stream which converts small overshoots into erro-

neously received pulses.

which translates to a photodiode current as follows (using

the BPW34FA data sheet specs):

2

7mm

I

= 40mW/sr

(

•

)

PD MIN

(

)

2

)

1000mm

(

0.65A / W 0.95 0.95 = 164nA

)( )(

(

)

The7mm²termisthephotodiodearea. The1000mmisthe

distance from the light source. The 0.65A/W is the spectral

sensitivity at 880nm wavelength. The first 0.95 term is the

relative sensitivity at 850nm wavelength and the second

term is the sensitivity at 15° off axis. Similar calculations

are detailed in the Infrared Data Association Serial Infrared

(SIR) Physical Layer Link Specification, version 1.0. This

minimum photocurrent implies that the input-referred

noise current of the receiver be less than 13.7nA rms for a

bit error rate of 1E-9. With an 800kHz lowpass filter on the

preampoutputtheLT1319hasapproximately3.6nArmsof

input-referred current noise. The maximum photodiode

current at 20mm, on-axis with 500mW/sr intensity:

4. As a general rule, place the lowest frequency highpass

aroundthepreampandthehighesthighpassaroundthe

gain stage or between the preamp and gain stage. The

reason for this is again due to high signal levels where

there can be slow photocurrent tails. The tail response

can be filtered out by high enough frequency filters.

5. Inallcaseswithcustomfiltering,orwhenmodifyingone

of the applications presented in this data sheet, try the

system over the full distance range with a full range of

dutycycledatastreams. Modulationmethodswithfixed

orlimiteddutycyclearesuperiorbecausetheyhavelittle

or no data dependent problems.

2

7mm

Dynamic Range

I

= 500mW/sr

•

(

)

PD MAX

(

)

2

20mm

The calculation of dynamic range can only be made in the

context of a specific modulation scheme and with the

system variations taken into account. The required infor-

mation includes: minimum signal-to-noise ratio (or BER,

Bit Error Rate requirement), photodiode capacitance at

1.9V back bias, preamp noise spectrum, preamp output

filtering, AC loop cutoff frequencies, modulation method,

demodulation method including allowable pulse widths

and the effect of missing or extra pulses, photodiode rise

and fall times, and ambient interference. The best solution

istoexperimentallydeterminethemaximumandminimum

distancesatwhichadesiredBERisobtained.Thismeasure

ofdynamicrangeismoremeaningfulintermsoftheoverall

system than any analytic solution.

(

)

0.65A/ W 0.95 = 5.4mA

)(

(

)

so we see that the dynamic range requirement is 90.4dB.

What is not obvious, however, is that the photodiode

output current is not simply a pulse of current, there is a

significanttailathighcurrentlevelsthathasatimeconstant

of more than 1µs which can cause distortion in the output

pulse width of the LT1319. This tail can be shown in the

following photograph which shows the voltage across a 5k

resistor that is connected between the anode of a photo-

diode and ground. The cathode of the photodiode is

connected to 2V. There is a 2pF Schottky diode across the

resistor to clamp the voltage swing to less than 0.5V. With

about 30pF photodiode capacitance and 10pF for an oscil-

loscope probe, any tail observed with a time constant

greater than 210ns is due to decaying photocurrent. The

Using the IRDA-SIR modulation scheme as an example,

however, we can illustrate how some limits on the required

receiver/photodiode combination can be obtained. The

minimum light intensity in the angular range is 40mW/sr

8

LT1319

U

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APPLICATIONS INFORMATION

or nominally 0.68mV for R_T1= 30k. The largest practical

value of R_T1is 39k. The limitation tends to be switching

transients at the comparator outputs parasitically coupling

to the FILTIN or FILTINL inputs and is layout dependent.

first trace in the photograph shows the current with the

photodiode 10cm from a source with 100mW/sr intensity.

At 200mV/div, there is about 40µA of peak current and the

decay is consistent with the 210ns time constant. The

lower trace shows the current with the photodiode 2cm

from the LEDs where the photodiode current is theoreti-

cally25timesgreaterthanat10cm. Thevoltageisclamped

by the photodiode to nearly 0.4V, but what is now notice-

able is that there is a tail with a time constant a bit greater

than 1µs. If the signal is AC coupled and has a low duty

cycle, the waveform will be centered at the very bottom

which can result in very wide output pulses. This issue will

be discussed later in more detail and a method to circum-

vent it will be shown.

Extending Short Range Performance

The short range performance of the LT1319 is normally

limited by the photocurrent tail, but in some instances the

peak current level cannot be supported by the output of the

preamplifier and the input will sag at Pin 2. Typically the

maximum input current is 6mA. To increase this current to

20mAormore, placeanNPNtransistorwithitsemittertied

to Pin 2, the base to Pin 4 and collector to the 5V supply.

The choice of transistor is dependent on the bandwidth

required for the preamp. The base-emitter capacitance of

the transistor (C_JE), is in parallel with the 15k feedback

resistor of the preamplifier and performs a lowpass filter-

ing function. For modest data rates such as IRDA-SIR and

Sharp/Newton a 2N3904 limits the bandwidth to 2MHz

which is ample. For the highest data rates, a transistor with

f_Tgreater than 1GHz is needed such as MMBR941LT1.

Photocurrent Waveforms

10cm

Another issue with large input signals is the photocurrent

tail. WhenthistailisACcoupledandthedatahasalowduty

cycle, the output pulse width can become so wide that it

extendsintothenextbitinterval.Ahighpassfiltercanreject

thistail,butforthecaseofIRDA-SIR,rejectingthe1µstime

constantcancauserejectionofthe1.6µspulsewhichleads

toalossofsensitivityandreducedmaximumlinkdistance.

Thecircuitonthefrontpageofthedatasheetusesa500kHz

highpass that trades off some sensitivity for rejection of

this tail. Unfortunately both maximum and minimum dis-

tance are compromised. An alternative is shown in the

IRDA-SIR/FIRapplication.Inthisinstancethefinalhighpass

filter for SIR is moved into 450kHz, but a clamp/squelch

circuitconsistingofQ1, Q2, D3andR_C1toR_C6isadded. Q1

isusedasdescribedabovetoclamptheinput, buttheinput

currentlevelatwhichtheclampengageshasbeenmodified

by R_C1and R_C2.

2cm

1319 AI01

Threshold Adjustment

The comparator thresholds are set by the current into Pin

11. The simplest method of setting this current is by a

resistor, R_T1tied between Pin 11 and Pin 15 (V_CC). Pin 11

should be bypassed. The current is given by:

V − 0.9V

(

)

CC

I_TH=

R + 2kΩ

(

)

T1

The threshold referred to the input of the filter buffer is:

Without the resistors, Q1 would turn on when the voltage

across the 15k resistor in the preamp reaches about 0.7V

(a current of 0.7V/15kΩ = 47µA). The drop across R_C1

reducesthisvoltagebyabout365mV.Thedropissetbythe

I

• 4 • 500Ω

400V/ V

TH

V

=

TH

9

LT1319

U

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APPLICATIONS INFORMATION

current through R_C2which is [V_CC– (V_BIAS+ 0.365V)]/

15kΩ = 182µA where V_BIAS= 1.9V. At this new level

(0.335V/15kΩ = 22.3µA), Q1 turns on which clamps the

preamp output. The collector current of Q1 provides base

drive for Q2 which saturates and pulls its collector close to

5V. The FILT1, FILT2L and FILT2 inputs are now pulled

positive by R_C3, R_C4and R_C6which forces an offset at the

inputs to the gain stages and preamp. Referring to the

Block Diagram, pulling FILT2L or FILT2 positive a voltage

∆V provides a voltage of ∆V/11 at the inverting input of the

first gain stage. This offset effectively cuts off a portion of

the tail at high input levels. The magnitude of ∆V is set by

the value of R_C3, the current sinking capability of the

transconductancestages(100µA),thevalueofC_F4,C_F5and

the duty cycle of the data pulses. Likewise an offset of

∆V/10kΩ is created at the preamp input to reduce tail

current contributions.

RF quality capacitor to reduce the high frequency spikes.

The current must be selected to achieve the minimum

output light intensity at a given angle and must be lower

than the manufacturer’s maximum current rating at the

maximum duty cycle of the modulation method. The opti-

mum current is a function of the LED output, the LED

forward voltage, the drop across the transistor and the

minimum supply voltage.

V − V_LED− V_SW

(

)

CC

I_LED

=

R_SERIES

The minimum light output then can be obtained from the

LEDdatasheet.ForIRDA-SIRtheminimumintensityat15°

off axis is 40mW/sr. For IRDA-FIR the spec rises to

100mW/sr. To increase light output and distance of the

link, a second LED can be inserted in series with the first to

obtain twice the light output without consuming additional

supply current. The current variation will now be greater

becausetwoLEDforwarddropsmustbeaccountedforand

the drop across the series resistor is greatly reduced.

LED Drive Circuits

There are several simple circuits for driving LEDs. For low

speed modulation methods such as IRDA-SIR and Sharp/

Newton with pulses over 1µs, a 2N3904 in a SOT-23

packagecanbeusedasaswitchwithaseriesresistorinthe

collector to limit the current drive. This circuit is shown

below with a suggested limiting resistor of 16Ω which

typically sets the current at 200mA. The supply voltage

mustbewellbypassedattheconnectiontotheLEDinorder

for the supply not to sag when hit with a fast current pulse.

A10µFlowESRcapacitorshouldbeusedaswellasa0.1µF

For pulse widths less than 500ns the NPN should be

replaced by an N-channel MOSFET with on-resistance of

less than 1Ω with 5V on the gate. The FET can turn off

much more quickly than the saturated NPN and provides

a lower effective on-resistance. A suggested circuit is

shown below and includes three devices available in the

SOT-23 package.

U

TYPICAL APPLICATIONS

LED Drive Circuit

for IRDA-SIR and Sharp/Newton

2 LED Drive Circuit

for IRDA-FIR

Optional Clamp Circuit

V

CC

V

CC

HSDL4220

TSH5400

DN304

HSDL4220

TSH5400

DN304

2N3904 FOR <1MHz

MMBR941LT1 FOR >1MHz

R

F3

16Ω

R

3.9Ω

D2

R

470Ω

PIN 2

PIN 4

PREOUT

F4

IN

1319 TA03

Q1

2N3904

R

D1

100Ω

V

IN

Q1

NDS351N

TN0201T

2N7002

V

IN

1319 TA04

1319 TA05

10

LT1319

U

TYPICAL APPLICATIONS

IRDA-SIR/FIR and Sharp or TV Remote Data Receiver

E1

SHDN

E2

CC

E3

GND

V

CC

V

CC

D2

HSDL-4220

R

C5

1M

R

D2

R

C2

15k

Q2

6.8Ω

R

C1

2k

MMBT3906LT1

1/2W

R

D1

100Ω

Q1

R

C4

10k

E6

TX

MMBT941LT1

Q3

2N7002

Q4

2N7002

R

D3

10k

R

D3

BAS16

C3

10k

R

C6

1k

DRIVER

1

16

BYPASS

AN_GND

C

B3

2

15

14

13

12

11

10

9

V

IN

CC

10µF

3

U1

LT1319

R

C

B2

T1

30k

B1

0.1µF

SHDN

FILT1

10µF

E4

4

5

6

7

8

DATAL

PREOUT

SHARP OR

TV DATA

D1

TEMIC

BPV22NF

DIG_GND

V

BIAS

R

F2

1k

R

F1

1k

L

F1

100µH

V

FILTINL

FILT2L

FILTIN

TH

DATA

FILT2

C

T1

1µF

E5

IRDA-SIR/FIR

DATA

C

F4

10nF

DGND

B4

C

F3

1nF

C

F2

33pF

F1

330pF

C

F5

470pF

1µF

V

CC

C

CC

F7

F6

AGND

0.1µF

2.7nF

LT1319 • TA02

1

2

1

2

R

S1

5.1k

R

S2

5.1k

IRDA

SIR

NOTES:

1. FOR IRDA-SIR/FIR OR TV REMOTE,

Q5 SHOULD BE TURNED ON WITH

A HIGH LOGIC INPUT

Q5

Q6

JMP1

JMP2

FIR

MMBT3904LT1

SHARP

3

2. FOR IRDA-SIR, Q6 SHOULD BE TURNED

ON WITH A HIGH LOGIC INPUT

3. FOR SHARP ASK, C = 1nF, L = 100µH

4. FOR TV REMOTE, C = 33nF, L = 470µH

DGND

F3 F1

F3

F1

PC Board Layout for IRDA-SIR/FIR and Sharp or TV Remote Data Receiver

COMPONENT

TOP

BOTTOM

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

11

LT1319

U

Dimensions in inches (millimeters) unless otherwise noted.

PACKAGE DESCRIPTION

S Package

16-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.386 – 0.394*

(9.804 – 10.008)

16

15

14

13

12

11

10

9

0.150 – 0.157**

(3.810 – 3.988)

0.228 – 0.244

(5.791 – 6.197)

5

7

8

1

2

3

4

6

0.010 – 0.020

(0.254 – 0.508)

× 45°

0.053 – 0.069

(1.346 – 1.752)

0.004 – 0.010

(0.101 – 0.254)

0.008 – 0.010

(0.203 – 0.254)

0° – 8° TYP

0.050

(1.270)

TYP

0.014 – 0.019

(0.355 – 0.483)

0.016 – 0.050

0.406 – 1.270

S16 0695

*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

RELATED PARTS

PART NUMBER

DESCRIPTION

COMMENTS

LT1113

Dual, Low Noise Precision JFET Input Op Amp

4.5nV/√Hz Input Voltage Noise

LT1169

Dual, Low Noise Picoampere Bias Current Op Amp

Low Noise, High Speed Op Amp (A ≥ 10)

JFET Input, 10pA Max

LT1222

500MHz Gain Bandwidth, External Comp Pin

V

LT/GP 1095 2K REV A • PRINTED IN USA

LINEAR TECHNOLOGY CORPORATION 1995

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

12

●

(408) 432-1900 FAX: (408) 434-0507 TELEX: 499-3977


型号：	LT1319CS
厂家：	Linear
描述：	Multiple Modulation Standard Infrared Receiver 多种调制标准的红外接收器
文件：	总12页 (文件大小：283K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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