LT1961IMS8E_技术文档

LT1961

1.5A, 1.25MHz Step-Up

Switching Regulator

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FEATURES

DESCRIPTIO

The LT^®1961 is a 1.25MHz monolithic boost switching

regulator. A high efficiency 1.5A, 0.2Ω switch is included

on the die together with all the control circuitry required to

complete a high frequency, current-mode switching regu-

lator. Current-mode control provides fast transient re-

sponse and excellent loop stability.

■

1.5A Switch in a Small MSOP Package

■

Constant 1.25MHz Switching Frequency

■

Wide Operating Voltage Range: 3V to 25V

■

High Efficiency 0.2Ω Switch

1.2V Feedback Reference Voltage

■

±2% Overall Output Voltage Tolerance

■

Uses Low Profile Surface Mount External

New design techniques achieve high efficiency at high

switchingfrequenciesoverawideoperatingvoltagerange.

A low dropout internal regulator maintains consistent

performance over a wide range of inputs from 24V sys-

tems to Li-Ion batteries. An operating supply current of

1mA maintains high efficiency, especially at lower output

currents. Shutdown reduces quiescent current to 6μA.

Maximum switch current remains constant at all duty

cycles. Synchronization allows an external logic level

signal to increase the internal oscillator from 1.5MHz to

2MHz.

Components

■

Low Shutdown Current: 6μA

■

Synchronizable from 1.5MHz to 2MHz

■

Current-Mode Loop Control

■

Constant Maximum Switch Current Rating at All Duty

Cycles*

■

Thermally Enhanced Exposed Pad 8-Lead Plastic

^{MSOP Package}U

APPLICATIO S

■

DSL Modems

The LT1961 is available in an exposed pad, 8-pin MSOP

package. Full cycle-by-cycle switch current limit protec-

tion and thermal shutdown are provided. High frequency

operationallowsthereductionofinputandoutputfiltering

components and permits the use of chip inductors.

■

Portable Computers

■

Battery-Powered Systems

Distributed Power

■

, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.

All other trademarks are the property of their respective owners. *Patent Pending

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TYPICAL APPLICATIO

Efficiency vs Load Current

5V to 12V Boost Converter

90

6.8μH

85

UPS120

V

OUT

80

75

70

65

60

1

5

2

6

V

IN

12V

V

IN

SW

FB

5V

0.5A*

LT1961

2.2μF

90.9k

OPEN

OR

HIGH

= ON

CERAMIC

SHDN

SYNC GND

3,4

V

C

8

7

10μF

CERAMIC

10k

1%

6800pF

6.8k

100pF

V

= 5V

OUT

IN

= 12V

0

100

200

300

400

500

*MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING.

1961 TA01

LOAD CURRENT (mA)

1961 TA01a

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LT1961

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ABSOLUTE MAXIMUM RATINGS

PI CO FIGURATIO

(Note 1)

TOP VIEW

Input Voltage .......................................................... 25V

Switch Voltage......................................................... 35V

SHDN Pin ............................................................... 25V

FB Pin Current ....................................................... 1mA

SYNC Pin Current .................................................. 1mA

Operating Junction Temperature Range (Note 2)

LT1961E, LT1961I ........................... –40°C to 125°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

V

1

2

3

4

8 SYNC

IN

SW

GND

7 V

C

6

5

FB

SHDN

MS8E PACKAGE

8-LEAD PLASTIC MSOP

GROUND PAD CONNECTED

TO LARGE COPPER AREA

T_JMAX= 125°C, θ_JA= 50°C/W

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W

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ORDER I FOR ATIO

LEAD FREE FINISH

LT1961EMS8E#PBF

LT1961IMS8E#PBF

LEAD BASED FINISH

LT1961EMS8E

TAPE AND REEL

PART MARKING*

LTQY

PACKAGE DESCRIPTION

TEMPERATURE RANGE

–40°C to 125°C

LT1961EMS8E#TRPBF

LT1961IMS8E#TRPBF

TAPE AND REEL

8-Lead Plastic MSOP

PACKAGE DESCRIPTION

8-Lead Plastic MSOP

LTQY

–40°C to 125°C

PART MARKING*

LTQY

TEMPERATURE RANGE

–40°C to 125°C

LT1961EMS8E#TR

LT1961IMS8E#TR

LT1961IMS8E

LTQY

–40°C to 125°C

Consult LTC Marketing for parts specified with wider operating temperature ranges. *Temperature grades are identified by a label on the shipping container.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/

ELECTRICAL CHARACTERISTICS ^The●denotesthespecificationswhichapplyoverthefulloperatingtemperature

range, otherwise specifications are at T_A= 25°C. V_IN= 15V, V_C= 0.8V, SHDN, SYNC and switch open unless otherwise noted.

PARAMETER

CONDITION

MIN

3

TYP

MAX

25

UNITS

V

Recommended Operating Voltage

Maximum Switch Current Limit

Oscillator Frequency

●

1.5

1

2

3

A

3.3V < V < 25V

1.5

500

2.73

1.3

MHz

mV

V

IN

Switch On Voltage Drop

I

= 1.5A

310

2.6

0.9

27

SW

V

Undervoltage Lockout

Supply Current

(Note 3)

2.47

IN

I

= 0A

mA

mA/A

SW

Supply Current/I

= 1.5A

SW

Shutdown Supply Current

V

= 0V, V = 25V, V = 25V

6

20

45

μA

SHDN

IN

SW

●

Feedback Voltage

3V < V < 25V, 0.4V < V < 0.9V

1.182

1.176

1.2

1.218

1.224

V

IN

C

1961fa

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LT1961

ELECTRICAL CHARACTERISTICS ^The●denotesthespecificationswhichapplyoverthefulloperatingtemperature

range, otherwise specifications are at T_A= 25°C. V_IN= 15V, V_C= 0.8V, SHDN, SYNC and switch open unless otherwise noted.

PARAMETER

CONDITION

MIN

0

TYP

–0.2

350

850

–120

110

2.4

MAX

UNITS

FB Input Current

●

–0.4

μA

FB to V Voltage Gain

0.4V < V < 0.9V

150

500

–85

70

C

FB to V Transconductance

ΔI = ±10μA

VC

●

1300

–165

165

μMho

μA

A/V

V

C

V Pin Source Current

C

V

= 1V

FB

V Pin Sink Current

C

= 1.4V

V Pin to Switch Current Transconductance

C

V Pin Minimum Switching Threshold

C

Duty Cycle = 0%

0.3

V Pin 1.5A I Threshold

0.9

V

C

SW

Maximum Switch Duty Cycle

V = 1.2V, I = 100mA

●

80

75

70

90

80

75

%

C

SW

V = 1.2V, I = 1A, 25°C ≤ T ≤ 125°C

C

SW

A

V = 1.2V, I = 1A, T ≤ 25°C

C

SW

A

SHDN Threshold Voltage

●

1.28

–7

4

1.35

–10

7

1.42

–13

10

V

μA

SHDN Input Current (Shutting Down)

SHDN Threshold Current Hysteresis

SYNC Threshold Voltage

SHDN = 60mV Above Threshold

SHDN = 100mV Below Threshold

μA

1.5

2.2

2

V

SYNC Input Frequency

1.5

MHz

kΩ

SYNC Pin Resistance

I

= 1mA

20

SYNC

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 3: Minimum input voltage is defined as the voltage where the

internal regulator enters lockout. Actual minimum input voltage to

maintain a regulated output will depend on output voltage and load

current. See Applications Information.

Note 2: The LT1961E is guaranteed to meet performance specifications

from 0°C to 125°C junction temperature. Specifications over the –40°C to

125°C operating junction temperature range are assured by design,

characterization and correlation with statistical process controls. The

LT1961I is guaranteed over the – 40ºC to 125ºC operating junction

temperature range.

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LT1961

TYPICAL PERFORMANCE CHARACTERISTICS

U W

FB vs Temperature

Switch On Voltage Drop

Oscillator Frequency

1.22

1.21

1.20

1.19

1.18

400

350

300

250

200

150

100

50

1.5

1.4

1.3

1.2

1.1

T

= 25°C

A

125°C

25°C

–40°C

0

–50 –25

0

25

50

75 100 125

0

0.5

1

1.5

–50 –25

0

25

50

75 100 125

TEMPERATURE (°C)

SWITCH CURRENT (A)

TEMPERATURE (°C)

1961 G01

1961 G02

1961 G03

SHDN Threshold vs Temperature

SHDN Supply Current vs V_IN

SHDN I_PCurrent vs Temperature

1.40

1.38

1.36

1.34

1.32

1.30

7

6

5

4

3

2

1

0

–12

–10

–8

–6

–4

–2

0

T

= 25°C

A

SHDN = 0V

SHUTTING DOWN

STARTING UP

–50 –25

0

25

50

75 100 125

0

5

10

15

(V)

20

25

30

–50 –25

0

25

50

75 100 125

TEMPERATURE (°C)

V

TEMPERATURE (°C)

IN

1961 G04

1961 G05

1961G06

SHDN Supply Current

Input Supply Current

Current Limit Foldback

1200

1000

800

600

400

200

0

300

250

200

150

100

50

2.0

1.5

1.0

0.5

0

40

30

20

10

0

T

= 25°C

IN

T

= 25°C

T

= 25°C

A

V

= 15V

SWITCH CURRENT

MINIMUM

INPUT

VOLTAGE

FB CURRENT

0

0.2 0.4 0.6 0.8

1

1.2 1.4

0

0.2

0.4

0.6

0.8

1

1.2

0

5

10

15

20

25

30

SHUTDOWN VOLTAGE (V)

FEEDBACK VOLTAGE (V)

INPUT VOLTAGE (V)

1961 G07

1961 G09

1961 G08

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LT1961

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PIN FUNCTIONS

FB:Thefeedbackpinisusedtosetoutputvoltageusingan

external voltage divider that generates 1.2V at the pin with

the desired output voltage. If required, the current limit

can be reduced during start up when the FB pin is below

0.5V (see the Current Limit Foldback graph in the Typical

Performance Characteristics section). An impedance of

less than 5kΩ at the FB pin is needed for this feature to

operate.

V_SW: The switch pin is the collector of the on-chip power

NPN switch and has large currents flowing through it.

Keep the traces to the switching components as short as

possible to minimize radiation and voltage spikes.

SYNC: The sync pin is used to synchronize the internal

oscillator to an external signal. It is directly logic compat-

ible and can be driven with any signal between 20% and

80% duty cycle. The synchronizing range is equal to initial

operating frequency, up to 2MHz. See Synchronization

section in Applications Information for details. When not

in use, this pin should be grounded.

V_IN: This pin powers the internal circuitry and internal

regulator. Keep the external bypass capacitor close to this

pin.

GND: Short GND pins 3 and 4 and the exposed pad on the

PCB. The GND is the reference for the regulated output, so

load regulation will suffer if the “ground” end of the load

is not at the same voltage as the GND of the IC. This

condition occurs when the load current flows through the

metal path between the GND pins and the load ground

point. Keep the ground path short between the GND pins

and the load and use a ground plane when possible. Keep

thepathbetweentheinputbypassandtheGNDpinsshort.

Theexposedpadshouldbeattachedtoalargecopperarea

to improve thermal resistance.

SHDN: The shutdown pin is used to turn off the regulator

and to reduce input drain current to a few microamperes.

The 1.35V threshold can function as an accurate under-

voltage lockout (UVLO), preventing the regulator from

operating until the input voltage has reached a predeter-

mined level. Float or pull high to put the regulator in the

operating mode.

V_C: The V_Cpin is the output of the error amplifier and the

input of the peak switch current comparator. It is normally

used for frequency compensation, but can do double duty

as a current clamp or control loop override. This pin sits

at about 0.3V for very light loads and 0.9V at maximum

load.

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LT1961

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BLOCK DIAGRAM

The LT1961 is a constant frequency, current-mode boost

converter. This means that there is an internal clock and

twofeedbackloopsthatcontrolthedutycycleofthepower

switch. In addition to the normal error amplifier, there is a

current sense amplifier that monitors switch current on a

cycle-by-cycle basis. A switch cycle starts with an oscilla-

tor pulse which sets the R_Sflip-flop to turn the switch on.

When switch current reaches a level set by the inverting

input of the comparator, the flip-flop is reset and the

switch turns off. Output voltage control is obtained by

using the output of the error amplifier to set the switch

current trip point. This technique means that the error

amplifier commands current to be delivered to the output

rather than voltage. A voltage fed system will have low

phase shift up to the resonant frequency of the inductor

and output capacitor, then an abrupt 180° shift will occur.

The current fed system will have 90° phase shift at a much

lower frequency, but will not have the additional 90° shift

until well beyond the LC resonant frequency. This makes

itmucheasiertofrequencycompensatethefeedbackloop

and also gives much quicker transient response.

A comparator connected to the shutdown pin disables the

internal regulator, reducing supply current.

V

IN

1

INTERNAL

CC

2.5V BIAS

REGULATOR

V

SLOPE COMP

Σ

0.3V

SW

2

1.25MHz

S

R

8

SYNC

Q1

POWER

SWITCH

OSCILLATOR

R

DRIVER

CURRENT

COMPARATOR

S

FLIP-FLOP

CIRCUITRY

+

–

CURRENT SENSE

SHUTDOWN

COMPARATOR

AMPLIFIER VOLTAGE

GAIN = 40

7μA

–

+

–

0.01Ω

1.35V

–

SHDN

5

6

FB

3μA

+

ERROR

3

4

GND

V

C

7

AMPLIFIER

1.2V

g

m

= 850μMho

GND

1767 F01

Figure 1. Block Diagram

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LT1961

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APPLICATIONS INFORMATION

FB RESISTOR NETWORK

defines the pole frequency of the output stage, an X7R or

X5R type ceramic, which have good temperature stability,

is recommended.

The suggested resistance (R2) from FB to ground is 10k

1%. This reduces the contribution of FB input bias current

to output voltage to less than 0.2%. The formula for the

resistor (R1) from V_OUTto FB is:

Tantalum capacitors are usually chosen for their bulk

capacitance properties, useful in high transient load appli-

cations. ESR rather than absolute value defines output

ripple at 1.25MHz. Values in the 22μF to 100μF range are

generally needed to minimize ESR and meet ripple current

ratings. Care should be taken to ensure the ripple ratings

are not exceeded.

R2 V_OUT−1.2

(

)

R1=

1.2 − R2(0.2μA)

Table 1. Surface Mount Solid Tantalum Capacitor ESR and

Ripple Current

V

LT1961

SW

OUTPUT

E Case Size

ESR (Max,

Ω

)

Ripple Current (A)

0.7 to 1.1

0.4

ERROR

AMPLIFIER

AVX TPS, Sprague 593D

AVX TAJ

0.1 to 0.3

0.7 to 0.9

1.2V

+

–

R1

+

FB

D Case Size

AVX TPS, Sprague 593D

C Case Size

0.1 to 0.3

0.2 (typ)

0.7 to 1.1

0.5 (typ)

R2

10k

1961 F02

AVX TPS

V

C

GND

Figure 2. Feedback Network

INPUT CAPACITOR

Unlike the output capacitor, RMS ripple current in the

input capacitor is normally low enough that ripple current

rating is not an issue. The current waveform is triangular,

with an RMS value given by:

OUTPUT CAPACITOR

Step-up regulators supply current to the output in pulses.

The rise and fall times of these pulses are very fast. The

output capacitor is required to reduce the voltage ripple

this causes. The RMS ripple current can be calculated

from:

0.29 V V_OUT− V

(

^IN)(

IN

)

I_{RIPPLE RMS}

=

(

)

L f V_OUT

( )( )(

)

I_{RIPPLE RMS}= I_OUTV_OUT− V

/

V

IN

(

IN

)

(

)

Athigherswitchingfrequency,theenergystoragerequire-

ment of the input capacitor is reduced so values in the

range of 1μF to 4.7μF are suitable for most applications.

Y5V or similar type ceramics can be used since the

absolute value of capacitance is less important and has no

significant effect on loop stability. If operation is required

close to the minimum input voltage required by either the

output or the LT1961, a larger value may be necessary.

This is to prevent excessive ripple causing dips below the

minimum operating voltage resulting in erratic operation.

The LT1961 will operate with both ceramic and tantalum

output capacitors. Ceramic capacitors are generally cho-

sen for their small size, very low ESR (effective series

resistance), and good high frequency operation, reducing

outputripplevoltage. TheirlowESRremovesausefulzero

in the loop frequency response, common to tantalum

capacitors. To compensate for this, the V_Cloop compen-

sationpolefrequencymusttypicallybereducedbyafactor

of 10. Typical ceramic output capacitors are in the 1μF to

10μF range. Since the absolute value of capacitance

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LT1961

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APPLICATIONS INFORMATION

INDUCTOR CHOICE AND MAXIMUM OUTPUT

CURRENT

(V )²(V_OUT– V )

0.4(V_OUT)²(I_OUT)(f)

IN

L_MIN

=

When choosing an inductor, there are 2 conditions that

limit the minimum inductance; required output current,

and avoidance of subharmonic oscillation. The maximum

output current for the LT1961 in a standard boost con-

verter configuration with an infinitely large inductor is:

The inductor value may need further adjustment for other

factors such as output voltage ripple and filtering require-

ments. Remember also, inductance can drop significantly

with DC current and manufacturing tolerance.

The inductor must have a rating greater than its peak

operating current to prevent saturation resulting in effi-

ciency loss. Peak inductor current is given by:

V • η

V_OUT

IN

I_OUT(MAX)= 1.5A

Where η = converter efficiency (typically 0.87 at high

current).

(V_OUT)(I_OUT) V (V_OUT− V )

IN

I_LPEAK

=

+

V • η

2V_OUT(L)(f)

As the value of inductance is reduced, ripple current

increases and I_OUT(MAX)is reduced. The minimum induc-

tance for a required output current is given by:

IN

Also, consideration should be given to the DC resistance

of the inductor. Inductor resistance contributes directly to

the efficiency losses in the overall converter.

V (V_OUT– V )

IN

L_MIN

=

SuitableinductorsareavailablefromCoilcraft,Coiltronics,

Dale, Sumida, Toko, Murata, Panasonic and other manu-

factures.

⎛

⎞

(V_OUT)(I_OUT

)

2V_OUT(f) 1.5 –

⎜

⎟

V • η

IN

⎝

⎠

Table 2

PART NUMBER VALUE (uH) I

Coiltronics

The second condition, avoidance of subharmonic oscilla-

tion, mustbemetiftheoperatingdutycycleisgreaterthan

50%. The slope compensation circuit within the LT1961

prevents subharmonic oscillation for inductor ripple cur-

rents of up to 0.7A_P-P, defining the minimum inductor

value to be:

(Amps) DCR (Ω) HEIGHT (mm)

SAT(DC)

TP1-2R2

2.2

4.7

10

1.3

0.188

0.111

0.181

0.146

1.8

2.2

3.0

TP2-2R2

1.5

TP3-4R7

TP4- 100

Murata

V (V_OUT– V )

IN

L_MIN

=

LQH1C1R0M04

LQH3C1R0M24

LQH3C2R2M24

LQH4C1R5M04

Sumida

1.0

2.2

1.5

0.51

1.0

0.79

1

0.28

0.06

0.1

1.8

2.0

2.6

0.7V_OUT(f)

These conditions define the absolute minimum induc-

tance. However, it is generally recommended that to

prevent excessive output noise, and difficulty in obtaining

stability, the ripple current is no more than 40% of the

average inductor current. Since inductor ripple is:

0.09

CD73- 100

10

2.2

6.2

10

1.44

1.32

1.4

0.080

0.058

0.071

0.048

3.5

1.8

2.8

CDRH4D18-2R2

CDRH5D18-6R2

CDRH5D28-100

Coilcraft

V (V_OUT– V )

IN

1.3

I_{P−P RIPPLE}

=

V_OUT(L)(f)

1008PS-272M

LPO1704-222M

LPO1704-332M

2.7

2.2

3.3

1.3

1.6

1.3

0.14

0.12

0.16

2.7

1.0

The recommended minimum inductance is:

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LT1961

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APPLICATIONS INFORMATION

CATCH DIODE

shutdown pin can be used. The threshold voltage of the

shutdown pin comparator is 1.35V. A 3μA internal current

sourcedefaultstheopenpinconditiontobeoperating(see

TypicalPerformanceGraphs). Currenthysteresisisadded

above the SHDN threshold. This can be used to set voltage

hysteresis of the UVLO using the following:

The suggested catch diode (D1) is a UPS120 or 1N5818

Schottky. It is rated at 1A average forward current and

20V/30V reverse voltage. Typical forward voltage is 0.5V

at 1A. The diode conducts current only during switch off

time. Peak reverse voltage is equal to regulator output

voltage. Average forward current in normal operation is

equal to output current.

V_H− V_L

R1=

7μA

SHUTDOWN AND UNDERVOLTAGE LOCKOUT

1.35V

R2 =

Figure 4 shows how to add undervoltage lockout (UVLO)

to the LT1961. Typically, UVLO is used in situations where

the input supply is current limited, or has a relatively high

source resistance. A switching regulator draws constant

power from the source, so source current increases as

source voltage drops. This looks like a negative resistance

loadtothesourceandcancausethesourcetocurrentlimit

or latch low under low source voltage conditions. UVLO

prevents the regulator from operating at source voltages

where these problems might occur.

V_H− 1.35V

(

)

+ 3μA

R1

V_H– Turn-on threshold

V_L– Turn-off threshold

Example: switching should not start until the input is

above 4.75V and is to stop if the input falls below 3.75V.

V_H= 4.75V

V_L= 3.75V

LT1961

4.75V − 3.75V

7μA

IN

INPUT

R1=

= 143k

1.35V

7μA

1.35V

R1

R2

3μA

V

CC

SHDN

R2 =

= 50.4k

4.75V − 1.35V

(

)

C1

+ 3μA

GND

143k

1961 F04

Keep the connections from the resistors to the SHDN pin

short and make sure that the interplane or surface capaci-

tance to the switching nodes are minimized. If high resis-

torvaluesareused,theSHDNpinshouldbebypassedwith

a 1nF capacitor to prevent coupling problems from the

switch node.

Figure 4. Undervoltage Lockout

An internal comparator will force the part into shutdown

below the minimum V_INof 2.6V. This feature can be used

to prevent excessive discharge of battery-operated sys-

tems. If an adjustable UVLO threshold is required, the

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LT1961

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APPLICATIONS INFORMATION

SYNCHRONIZATION

high speed switching current path, shown in Figure 5,

must be kept as short as possible. This is implemented in

the suggested layout of Figure 6. Shortening this path will

also reduce the parasitic trace inductance of approxi-

mately 25nH/inch. At switch off, this parasitic inductance

produces a flyback spike across the LT1961 switch. When

operating at higher currents and output voltages, with

poor layout, this spike can generate voltages across the

LT1961 that may exceed its absolute maximum rating. A

ground plane should always be used under the switcher

circuitry to prevent interplane coupling and overall noise.

The SYNC pin, is used to synchronize the internal oscilla-

tor to an external signal. The SYNC input must pass from

a logic level low, through the maximum synchronization

threshold with a duty cycle between 20% and 80%. The

input can be driven directly from a logic level output. The

synchronizing range is equal to initial operating frequency

up to 2MHz. This means that minimum practical sync

frequency is equal to the worst-case high self-oscillating

frequency (1.5MHz), not the typical operating frequency

of 1.25MHz. Caution should be used when synchronizing

above 1.7MHz because at higher sync frequencies the

amplitude of the internal slope compensation used to

prevent subharmonic switching is reduced. Higher induc-

tor values will tend to eliminate this problem. See Fre-

quency Compensation section for a discussion of an

entirely different cause of subharmonic switching before

assuming that the cause is insufficient slope compensa-

tion. Application Note 19 has more details on the theory

of slope compensation.

The V_Cand FB components should be kept as far away as

possible from the switch node. The LT1961 pinout has

been designed to aid in this. The ground for these compo-

nents should be separated from the switch current path.

Failure to do so will result in poor stability or subharmonic

like oscillation.

Board layout also has a significant effect on thermal

resistance. The exposed pad is the copper plate that runs

undertheLT1961die.Thisisthebestthermalpathforheat

out of the package. Soldering the pad onto the board will

reduce die temperature and increase the power capability

of the LT1961. Provide as much copper area as possible

aroundthispad.Addingmultiplesolderfilledfeedthroughs

under and around the pad to the ground plane will also

help. Similar treatment to the catch diode and inductor

terminations will reduce any additional heating effects.

LAYOUT CONSIDERATIONS

As with all high frequency switchers, when considering

layout, care must be taken to achieve optimal electrical,

thermal and noise performance. For maximum efficiency,

switch rise and fall times are typically in the nanosecond

range. To prevent noise both radiated and conducted, the

L1

D1

C3

V

OUT

SW

LT1961

HIGH

FREQUENCY

SWITCHING

PATH

V

IN

C1 LOAD

GND

1961 F05

Figure 5. High Speed Switching Path

1961fa

10

LT1961

U

W U U

APPLICATIONS INFORMATION

L1

6.8μH

D1

UPS120

OUTPUT

12V

0.5A*

INPUT

5V

V

IN

SW

FB

C3

OPEN

OR

HIGH

= ON

LT1961

R1

90.9k

2.2μF

CERAMIC

SHDN

SYNC GND

V

C

C1

R2

10k

1%

C2

6800pF

R3

6.8k

10μF

C4

100pF

CERAMIC

*MAXIMUM OUTPUT CURRENT IS SUBJECT TO THERMAL DERATING.

INPUT

GND

R3

L1

C4

C3

LT1961EMS8E

C2

KEEP FB AND V

COMPONENTS

AWAY FROM

C

D1

HIGH FREQUENCY,

HIGH INPUT

COMPONENTS

MINIMIZE

LT1961,

C1, D1 LOOP

U1

C1

R2

R1

GND

V

OUT

SOLDER EXPOSED

GROUND PAD

TO BOARD

KELVIN SENSE

PLACE FEEDTHROUGHS

V

OUT

AROUND GROUND PIN FOR

GOOD THERMAL CONDUCTIVITY

Figure 6. Typical Application and Suggested Layout (Topside Only Shown)

1961fa

11

LT1961

U

W U U

APPLICATIONS INFORMATION

THERMAL CALCULATIONS

with no device power, in an oven. The same measurement

can then be used in operation to indicate the die tempera-

ture.

Power dissipation in the LT1961 chip comes from four

sources:switchDCloss,switchACloss,drivecurrent,and

inputquiescentcurrent.Thefollowingformulasshowhow

to calculate each of these losses. These formulas assume

continuous mode operation, so they should not be used

for calculating efficiency at light load currents.

FREQUENCY COMPENSATION

Loop frequency compensation is performed on the output

of the error amplifier (V_Cpin) with a series RC network.

The main pole is formed by the series capacitor and the

output impedance (≈500kΩ) of the error amplifier. The

pole falls in the range of 2Hz to 20Hz. The series resistor

creates a “zero” at 1kHz to 5kHz, which improves loop

stability and transient response. A second capacitor, typi-

cally one-tenth the size of the main compensation capaci-

tor, is sometimes used to reduce the switching frequency

ripple on the V_Cpin. V_Cpin ripple is caused by output

voltage ripple attenuated by the output divider and multi-

plied by the error amplifier. Without the second capacitor,

V_Cpin ripple is:

(V_OUT− V )

IN

DC, duty cycle =

V_OUT

(V_OUT)(I_OUT

)

I_SW

=

V

IN

Switch loss:

P

SW

= (DC)(I_SW)²(R_SW)+ 17n I

V_OUT

f

(

^SW)(

)( )

V_INloss:

(V )(I_SW)(DC)

IN

P_VIN

=

+ 1mA(V )

IN

50

1.2(V

)(g )(R )

m C

RIPPLE

(V

V Pin Ripple =

C

R_SW= Switch resistance (≈0.27Ω hot)

)

OUT

V

m

= Output ripple (V

)

RIPPLE

P–P

Example: V_IN= 5V, V_OUT= 12V and I_OUT= 0.5A

g = Error amplifier transconductance

(≈850μmho)

Total power dissipation = 0.23 + 0.31 + 0.07 + 0.005 =

0.62W

R = Series resistor on V pin

C

V

OUT

= DC output voltage

Thermal resistance for LT1961 package is influenced by

the presence of internal or backside planes. With a full

plane under the package, thermal resistance will be about

50°C/W. Tocalculatedietemperature, usetheappropriate

thermalresistancenumberandaddinworst-caseambient

temperature:

To prevent irregular switching, V_Cpin ripple should be

kept below 50mV_P–P. Worst-case V_Cpin ripple occurs at

maximum output load current and will also be increased if

poor quality (high ESR) output capacitors are used. The

addition of a 47pF capacitor on the V_Cpin reduces switch-

ingfrequencyrippletoonlyafewmillivolts. Alowvaluefor

R_Cwill also reduce V_Cpin ripple, but loop phase margin

may be inadequate.

T_J= T_A+ θ_JA(P_TOT

)

If a true die temperature is required, a measurement of the

SYNC to GND pin resistance can be used. The SYNC pin

resistance across temperature must first be calibrated,

1961fa

12

LT1961

U

TYPICAL APPLICATIO S

Dual Output Flyback Converter

R2

10k

1%

R1

115k

1%

UPS140

T1*

V

IN

5V TO 10V

OUT

15V

+

2, 3

7

C1

+

•

P6KE-20A

C4

4.7μF

47μF

4

10

1N4148

•

FB

S/S

V

IN

8, 9

ON

V

SW

OFF

C5

47μF

LT1961

1

–V

OUT

–15V

UPS140

V

C

GND

C2

2.2nF

R3

10k

C3

100pF

*DALE LPE-4841-100MB

LT1961 • TA02

4V-9V_INto 5V_OUTSEPIC Converter**

**

4V TO 9V

V

IN

L1A*

10μH

D1

UPS120

V

IN

†

•

V

OUT

ON

V

S/S

SW

OFF

5V

R2

31.6k

1%

C2

LT1961

4.7μF

C1

+

FB

C

•

4.7μF

GND

+

C3

20V

L1B*

10μH

47μF

10V

R3

10k

1%

R1

10k

C4

2.2nF

C5

100pF

LT1961 • TA03

†

MAX I

OUT

*

**

BH ELECTRONICS 511-1012

INPUT VOLTAGE MAY BE GREATER OR

LESS THAN OUTPUT VOLTAGE

I

V

IN

OUT

0.59A 4V

0.65A 5V

0.70A 6V

0.74A 7V

0.80A 9V

1961fa

13

LT1961

U

TYPICAL APPLICATIO S

Single Li-Ion Cell to 5V

D1

UPS120

L1

4.7μH

V

OUT

5V

R1

31.6k

1%

V

IN

ON

V

S/S

SW

OFF

LT1961

SINGLE

Li-Ion

CELL

+

C4

FB

C

C1

10μF

47μF

GND

V

10V

R2

10k

1%

C2

2.2nF

C3

100pF

R3

10k

LT1961 • TA04

I

V

IN

OUT

0.75A 2.7V

0.93A 3.3V

1.0A 3.6V

1961fa

14

LT1961

U

PACKAGE DESCRIPTION

MS8E Package

8-Lead Plastic MSOP, Exposed Die Pad

(Reference LTC DWG # 05-08-1662 Rev D)

BOTTOM VIEW OF

EXPOSED PAD OPTION

2.06 ± 0.102

(.081 ± .004)

1

1.83 ± 0.102

(.072 ± .004)

0.889 ± 0.127

(.035 ± .005)

2.794 ± 0.102

(.110 ± .004)

5.23

(.206)

MIN

3.20 – 3.45

(.126 – .136)

2.083 ± 0.102

(.082 ± .004)

8

3.00 ± 0.102

(.118 ± .004)

(NOTE 3)

0.52

(.0205)

REF

0.65

(.0256)

BSC

0.42 ± 0.038

(.0165 ± .0015)

TYP

8

7 6 5

RECOMMENDED SOLDER PAD LAYOUT

3.00 ± 0.102

(.118 ± .004)

(NOTE 4)

4.90 ± 0.152

(.193 ± .006)

DETAIL “A”

0° – 6° TYP

0.254

(.010)

GAUGE PLANE

1

2

3

4

0.53 ± 0.152

(.021 ± .006)

1.10

(.043)

MAX

0.86

(.034)

REF

DETAIL “A”

0.18

(.007)

SEATING

PLANE

0.22 – 0.38

(.009 – .015)

TYP

0.1016 ± 0.0508

(.004 ± .002)

0.65

(.0256)

BSC

MSOP (MS8E) 0307 REV D

NOTE:

1. DIMENSIONS IN MILLIMETER/(INCH)

2. DRAWING NOT TO SCALE

3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.

MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.

INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE

5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX

1961fa

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

15

LT1961

U

TYPICAL APPLICATIO

High Voltage Laser Power Supply

0.01μF

5kV

1800pF

10kV

47k

5W

1800pF

10kV

8

11

HV DIODES

L1

3

1

5

4

LASER

2

+

2.2μF

0.47μF

Q1

Q2

150Ω

L2

10μH

MUR405

V

10k

SW

LT1961

GND

10k

V

IN

12V TO 25V

V

FB

IN

V

1N4002

(ALL)

+

V

L1 =

Q1, Q2 =

COILTRONICS CTX02-11128

ZETEX ZTX849

WIMA 3X 0.15μF TYPE MKP-20

SEMTECH-FM-50

HUGHES 3121H-P

0.1μF

IN

190Ω

1%

2.2μF

0.47μF =

C

HV DIODES =

LASER =

+

10μF

LT1961 • TA05

COILTRONICS (407) 241-7876

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COMMENTS

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LTC3402

Single Cell, High Current (2A), Micropower, Synchronous 3MHz

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V

IN

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OUT

Synchronizable, Oscillator from 100kHz to 3MHz, MS10 Package

LTC3405/

LTC3405A

1.5MHz High Efficiency, I

Step-Down Regulator

= 300mA, Monolithic Synchronous

V

IN

= 2.5V to 5.5V, V to 0.8V, Up to 95% Efficiency, 100%

OUT

Duty Cycle, I = 20μA, ThinSOT Package

Q

ThinSOT is a trademark of Linear Technology Corporation.

1961fa

LT 0707 REV A • PRINTED IN USA

LinearTechnology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

16

●

(408) 432-1900 FAX: (408) 434-0507 www.linear.com


型号：	LT1961IMS8E
厂家：	Linear
描述：	1.5A, 1.25MHz Step-Up Switching Regulator 1.5A ， 1.25MHz的升压型开关稳压器稳压器开关式稳压器或控制器电源电路开关式控制器光电二极管
文件：	总16页 (文件大小：149K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

LT1961IMS8E [Linear]

相关型号：

LT1961IMS8E#PBF

LT1961IMS8E#TRPBF

LT1961IMS8EPBF

LT1961IMS8ETR

LT1961IMS8ETRPBF

LT1962

LT1962EMS8

LT1962EMS8#PBF

LT1962EMS8#TR

LT1962EMS8#TRPBF

LT1962EMS8-1.5

LT1962EMS8-1.5#PBF