LTC1625IS#TR [Linear]
LTC1625 - No RSENSE Current Mode Synchronous Step-Down Switching Regulator; Package: SO; Pins: 16; Temperature Range: -40°C to 85°C;
| 型号： | LTC1625IS#TR |
| 厂家： | 
Linear |
| 描述： | LTC1625 - No RSENSE Current Mode Synchronous Step-Down Switching Regulator; Package: SO; Pins: 16; Temperature Range: -40°C to 85°C 稳压器 开关 |
| 文件： | 总24页 (文件大小：457K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC1625
No R
TM Current Mode
SENSE
Synchronous Step-Down
Switching Regulator
U
DESCRIPTION
FEATURES
The LTC®1625 is a synchronous step-down switching
regulator controller that drives external N-Channel power
MOSFETs using few external components. Current mode
control with MOSFET VDS sensing eliminates the need for
a sense resistor and improves efficiency. The frequency of
a nominal 150kHz internal oscillator can be synchronized
to an external clock over a 1.5:1 frequency range.
Burst ModeTM operation at low load currents reduces
switchinglossesandlowdropoutoperationextendsoper-
ating time in battery-powered systems. A forced continu-
ous mode control pin can assist secondary winding
regulation by disabling Burst Mode operation when the
main output is lightly loaded.
■
Highest Efficiency Current Mode Controller
■
No Sense Resistor Required
■
Stable High Current Operation
■
Dual N-Channel MOSFET Synchronous Drive
■
Wide VIN Range: 3.7V to 36V
■
Wide VOUT Range: 1.19V to VIN
■
±1% 1.19V Reference
■
Programmable Fixed Frequency with Injection Lock
■
Very Low Drop Out Operation: 99% Duty Cycle
■
Forced Continuous Mode Control Pin
■
Optional Programmable Soft Start
■
Pin Selectable Output Voltage
■
Foldback Current Limit
■
Output Overvoltage Protection
Fault protection is provided by foldback current limiting
and an output overvoltage comparator. An external ca-
pacitor attached to the RUN/SS pin provides soft start
capability for supply sequencing. A wide supply range
allows operation from 3.7V (3.9V for LTC1625I) to 36V at
the input and 1.19V to VIN at the output.
■
Logic Controlled Micropower Shutdown: IQ < 30µA
■
Available in 16-Lead Narrow SSOP and SO Packages
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APPLICATIONS
■
Notebook and Palmtop Computers, PDAs
, LTC and LT are registered trademarks of Linear Technology Corporation.
No RSENSE and Burst Mode are trademarks of Linear Technology Corporation.
■
Cellular Telephones and Wireless Modems
■
Battery Chargers
Distributed Power
■
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TYPICAL APPLICATION
Efficiency vs Load Current
100
V
V
= 10V
IN
IN
V
= 5V
5V TO
28V
OUT
SYNC
V
IN
C
IN
+
RUN/SS
TK
10µF
30V
×2
C
SS
M1
90
80
70
60
TG
0.1µF
Si4410DY
L1
10µH
V
= 3.3V
OUT
LTC1625
V
OUT
I
SW
TH
C
0.22µF
3.3V
B
R
BOOST
C
C
+
OUT 4.5A
D1
D
10k
B
100µF
10V
MBRS140T3
C
VCC
4.7µF
CMDSH-3
V
PROG
INTV
CC
C
C
×3
+
2.2nF
M2
Si4410DY
SGND
BG
V
PGND
OSENSE
1625 F01
0.01
0.1
1
10
LOAD CURRENT (A)
Figure 1. High Efficiency Step-Down Converter
1625 TA01
1
LTC1625
W W U W
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER I FOR ATIO
(Note 1)
ORDER PART
NUMBER
Input Supply Voltage (VIN, TK) ................. 36V to –0.3V
Boosted Supply Voltage (BOOST)............. 42V to –0.3V
Boosted Driver Voltage (BOOST – SW) ...... 7V to –0.3V
Switch Voltage (SW).....................................36V to –5V
EXTVCC Voltage ...........................................7V to –0.3V
ITH Voltage................................................2.7V to –0.3V
FCB, RUN/SS, SYNC Voltages .....................7V to –0.3V
TOP VIEW
EXTV
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
V
CC
IN
SYNC
RUN/SS
FCB
TK
LTC1625CGN
LTC1625CS
LTC1625IGN
LTC1625IS
SW
TG
I
BOOST
TH
SGND
INTV
BG
CC
V
OSENSE
V
OSENSE, VPROG Voltages ........(INTVCC + 0.3V) to –0.3V
V
PGND
PROG
Peak Driver Output Current < 10µs (TG, BG) ............ 2A
INTVCC Output Current ........................................ 50mA
Operating Ambient Temperature Range
GN PACKAGE
S PACKAGE
16-LEAD PLASTIC SSOP 16-LEAD PLASTIC SO
TJMAX = 125°C, θJA = 130°C/W (GN)
TJMAX = 125°C, θJA = 110°C/W (S)
LTC1625C............................................... 0°C to 70°C
LTC1625I (Note 5).............................. –40°C to 85°C
Junction Temperature (Note 2)............................. 125°C
Storage Temperature Range ................ –65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loop
I V
IN OSENSE
Feedback Current
V
PROG
Pin Open, I = 1.19V (Note 3)
10
50
nA
TH
V
Regulated Output Voltage
1.19V (Adjustable) Selected
3.3V Selected
I
= 1.19V (Note 3)
TH
OUT
V
V
V
Pin Open
= 0V
●
●
●
1.178
3.220
4.900
1.190
3.300
5.000
1.202
3.380
5.100
V
V
V
PROG
PROG
PROG
5V Selected
= INTV
CC
V
V
V
Reference Voltage Line Regulation
V
V
= 3.6V to 20V, I = 1.19V (Note 3),
0.001
0.01
%/V
LINEREG
IN
TH
Pin Open
PROG
Output Voltage Load Regulation
I
I
= 2V (Note 3)
= 0.5V (Note 3)
●
●
– 0.020
0.035
–0.2
0.2
%
%
LOADREG
TH
TH
Forced Continuous Threshold
Forced Continuous Current
Output Overvoltage Lockout
V
V
V
Ramping Negative
= 1.19V
●
1.16
1.24
1.19
–1
1.22
–2
V
µA
V
FCB
FCB
FCB
I
FCB
V
Pin Open
1.28
1.32
OVL
PROG
I
V
Input Current
PROG
PROG
3.3V V
V
PROG
V
PROG
= 0V
= 5V
–3.5
3.5
–7
7
µA
µA
OUT
5V V
OUT
I
Input DC Supply Current
Normal Mode
EXTV = 5V (Note 4)
CC
Q
500
15
µA
µA
Shutdown
V
= 0V, 3.7V < V < 15V
30
2
RUN/SS
IN
V
RUN/SS Pin Threshold
●
0.8
1.2
120
1.4
2.5
150
V
µA
RUN/SS
I
Soft Start Current Source
Maximum Current Sense Threshold
V
V
= 0V
4
RUN/SS
RUN/SS
∆V
= 1V, V
Pin Open
PROG
170
mV
SENSE(MAX)
OSENSE
TG Transition Time
Rise Time
TG t
TG t
C
LOAD
C
LOAD
= 3300pF
= 3300pF
50
50
150
150
ns
ns
R
F
Fall Time
2
LTC1625
ELECTRICAL CHARACTERISTICS TA = 25°C, VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
BG Transition Time
Rise Time
BG t
BG t
C
C
= 3300pF
= 3300pF
50
50
150
150
ns
ns
R
F
LOAD
LOAD
Fall Time
Internal V Regulator
CC
V
V
V
V
Internal V Voltage
6V < V < 30V, V = 4V
EXTVCC
●
●
5.0
5.2
–1
5.4
–2
V
%
INTVCC
LDOINT
LDOEXT
EXTVCC
CC
IN
INTV Load Regulation
I
I
I
= 20mA, V
= 20mA, V
= 20mA, V
= 4V
CC
CC
CC
CC
EXTVCC
EXTVCC
EXTVCC
EXTV Voltage Drop
= 5V
180
4.7
300
mV
V
CC
EXTV Switchover Voltage
Ramping Positive
4.5
CC
Oscillator
f
Oscillator Freqency
135
150
1.5
0.9
50
165
1.2
kHz
OSC
f /f
Maximum Synchronized Frequency Ratio
SYNC Pin Threshold (Figure 4)
SYNC Pin Input Resistance
H
OSC
V
Ramping Positive
V
SYNC
R
kΩ
SYNC
The
●
denotes specifications which apply over the full operating
Note 3: The LTC1625 is tested in a feedback loop that adjusts V
to
OSENSE
temperature range.
achieve a specified error amplifier output voltage (I ).
TH
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 4: Typical in application circuit with EXTV tied to V
= 5V,
CC
OUT
I
= 0A and FCB = INTV . Dynamic supply current is higher due
OUT CC
to the gate charge being delivered at the switching frequency. See
Applications Information.
Note 2: T is calculated from the ambient temperature T and power
J
A
dissipation P according to the following formula:
D
Note 5: Minimum input supply voltage is 3.9V at –40°C for industrial
grade parts.
LTC1625CGN/LTC1625IGN: T = T + (P • 130°C/W)
J
A
D
LTC1625CS/LTC1625IS: T = T + (P • 110°C/W)
J
A
D
3
LTC1625
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TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Input Voltage,
VOUT = 3.3V
Efficiency vs Input Voltage,
Efficiency vs Load Current
VOUT = 5V
100
95
90
85
80
75
70
100
95
90
85
80
75
70
100
90
80
70
60
50
FIGURE 1 CIRCUIT
= 2A
FIGURE 1 CIRCUIT
BURST
MODE
I
LOAD
I
= 2A
OPERATION
LOAD
CONTINUOUS
MODE
I
= 200mA
LOAD
I
= 200mA
LOAD
V
V
= 10V
OUT
EXTV = V
CC
IN
= 5V
OUT
0
10
15
20
25
30
0
10
15
20
25
30
0.001
0.01
0.1
LOAD CURRENT (A)
1
5
5
10
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
1625 G02
1625 G02
1625 G01
ITH Pin Voltage vs Load Current
Load Regulation
3.0
2.5
2.0
1.5
400
300
200
100
0
0
–0.05
–0.10
–0.15
–0.20
–0.25
FIGURE 1 CIRCUIT
FIGURE 1 CIRCUIT
= 5V – 5% DROP
FIGURE 1 CIRCUIT
V
V
= 20V
= 5V
V
IN
OUT
OUT
CONTINUOUS
MODE
1.0
0.5
0
Burst Mode
OPERATION
4
6
7
0
1
2
3
4
5
0
1
2
3
5
0
1
3
4
5
2
LOAD CURRENT (A)
LOAD CURRENT (A)
LOAD CURRENT (A)
1625 G05
1625 G04
1625 G06
Input and Shutdown Current
vs Input Voltage
INTVCC Load Regulation
1000
800
600
400
200
0
50
40
30
20
10
0
0
500
400
300
200
100
0
EXTV OPEN
CC
–0.5
–1.0
–1.5
–2.0
–2.5
SHUTDOWN
EXTV = 5V
CC
0
5
10
15
20
25
30
35
0
10
30
40
50
20
0
10
20
30
40
50
INPUT VOLTAGE (V)
INTV LOAD CURRENT (mA)
INTV LOAD CURRENT (mA)
CC
CC
1625 G07
1625 G08
1625 G09
4
LTC1625
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TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency
vs Temperature
Maximum Current Sense Voltage
vs Temperature
Maximum Current Sense Voltage
vs Duty Cycle
200
150
100
50
300
250
200
150
160
155
150
145
SYNC = 1.5V
SYNC = 0V
100
50
0
140
0
–40 –15 10
35
60
85 110 135
110 135
0
0.2
0.4
0.5
0.8
1.0
–40 –15
10
35
60
TEMPERATURE (°C)
85
TEMPERATURE (°C)
DUTY CYCLE
1625 G11
1625 G10
1625 G12
RUN/SS Pin Current
vs Temperature
Soft Start:
FCB Pin Current vs Temperature
Load Current vs Time
0
–1
–2
–3
–4
–5
0
–0.25
–0.50
–0.75
INDUCTOR
CURRENT
2A/DIV
RUN/SS
2V/DIV
–1.00
–1.25
–1.50
20ms/DIV
VIN = 20V
1625 F06
VOUT = 5V
RLOAD = 1Ω
FIGURE 1 CIRCUIT
60
TEMPERATURE (°C)
110 135
–40 –15 10
35
60
85 110 135
–40 –15
10
35
85
TEMPERATURE (°C)
1625 G14
1625 G13
Transient Response
(Burst Mode Operation)
Burst Mode Operation
Transient Response
VOUT
50mV/DIV
VOUT
50mV/DIV
VOUT
50mV/DIV
ITH
100mV/DIV
200µs/DIV
1625 F07
500µs/DIV
1625 F08
50µs/DIV
1625 F09
VIN = 20V
VOUT = 5V
VIN = 20V
V
IN = 20V
VOUT = 5V
VOUT = 5V
I
LOAD = 1A TO 4A
ILOAD = 50mA
FIGURE 1 CIRCUIT
ILOAD = 50mA TO 1A
FIGURE 1 CIRCUIT
FIGURE 1 CIRCUIT
5
LTC1625
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PIN FUNCTIONS
EXTVCC (Pin 1): INTVCC Switch Input. When the EXTVCC
voltage is above 4.7V, the switch closes and supplies
INTVCC power from EXTVCC. Do not exceed 7V at this pin.
Leaving VPROG open allows the output voltage to be set by
an external resistive divider between the output and
VOSENSE
.
SYNC (Pin 2): Synchronization Input for Internal Oscilla-
tor.Theoscillatorwillnominallyrunat150kHzwhenopen,
225kHz when tied above 1.2V, and will lock over a 1.5:1
clock frequency range.
PGND (Pin 9): Driver Power Ground. Connects to the
source of the bottom N-channel MOSFET, the (–) terminal
of CVCC and the (–) terminal of CIN.
BG (Pin 10): Bottom Gate Drive. Drives the gate of the
bottom N-channel MOSFET between ground and INTVCC.
RUN/SS (Pin 3): Run Control and Soft Start Input. A
capacitor to ground at this pin sets the ramp time to full
current output (approximately 1s/µF). Forcing this pin
below 1.4V shuts down the device.
INTVCC (Pin 11): Internal 5.2V Regulator Output. The
driver and control circuits are powered from this voltage.
Decouple this pin to power ground with a minimum of
4.7µF tantalum capacitance.
FCB (Pin 4): Forced Continuous Input. Tie this pin to
ground to force synchronous operation at low load, to a
resistive divider from the secondary output when using
a secondary winding, or to INTVCC to enable Burst Mode
operation at low load.
BOOST (Pin 12): Topside Floating Driver Supply. The (+)
terminalofthebootstrapcapacitorconnectshere.Thispin
swings from a diode drop below INTVCC to VIN + INTVCC.
TG (Pin 13): Top Gate Drive. Drives the top N-channel
MOSFET with a voltage swing equal to INTVCC minus a
diode drop, superimposed on the switch node voltage.
ITH (Pin 5): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 2.4V.
SW (Pin 14): Switch Node. The (–) terminal of the boot-
strap capacitor connects here. This pin swings from a
diode drop below ground up to VIN.
SGND (Pin 6): Signal Ground. Connect to the (–) terminal
of COUT
.
VOSENSE (Pin 7): Output Voltage Sense. Feedback input
from the remotely sensed output voltage or from an
external resistive divider across the output.
TK (Pin 15): Top MOSFET Kelvin Sense. MOSFET VDS
sensingrequiresthispintoberoutedtothedrainofthetop
MOSFET separately from VIN.
VPROG (Pin 8): Output Voltage Programming. When
VOSENSE is connected to the output, VPROG < 0.8V selects
a 3.3V output and VPROG > 3.5V selects a 5V output.
VIN (Pin 16): Main Supply Input. Decouple this pin to
ground with an RC filter (4.7Ω, 0.1µF) for applications
above 3A.
6
LTC1625
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FUNCTIONAL DIAGRA
TK
15
V
IN
SYNC
2
+
TA
×11
+
C
IN
–
–
BA
×11
0.95V
+
I
TH
5
+
0.6V
–
OSC
+
–
REV
R
C
I
2
+
–
S
Q
R
C
C1
TOP
I
1
I
THB
BOOST
12
+
–
0.5V
–
+
SLEEP
0.6V
C
B
B
TG
13
CL
SWITCH
LOGIC/
DROPOUT
COUNTER
M1
Ω
SW
14
g
= 1m
V
m
–
+
FB
INTV
CC
SHUTDOWN
D
B
EA
11
1.19V
+
OVERVOLTAGE
FCNT
+
0.6V
–
C
VCC
BG
10
–
+
M2
3µA
PGND
9
RUN/SS
3
6V
C
SS
1.19V
REF
–
V
1.28V
IN
16
OV
+
5.2V
LDO REG
SGND
6
1.19V
4.7V
+
–
+
–
F
1µA
L1
V
PROG
V
FCB
EXTV
CC
8
7
4
1
OSENSE
+
C
OUT
1625 BD
7
LTC1625
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OPERATIO
Main Control Loop
will attempt to turn on the top MOSFET continuously
(‘’dropout’’). A dropout counter detects this condition and
forces the top MOSFET to turn off for about 500ns every
tenth cycle to recharge the bootstrap capacitor.
The LTC1625 is a constant frequency, current mode
controller for DC/DC step-down converters. In normal
operation, the top MOSFET is turned on when the RS latch
is set by the on-chip oscillator and is turned off when the
current comparator I1 resets the latch. While the top
MOSFET is turned off, the bottom MOSFET is turned on
until either the inductor current reverses, as determined
by the current reversal comparator I2, or the next cycle
begins. Inductor current is measured by sensing the VDS
potential across the conducting MOSFET. The output of
the appropriate sense amplifier (TA or BA) is selected by
the switch logic and applied to the current comparator.
The voltage on the ITH pin sets the comparator threshold
corresponding to peak inductor current. The error ampli-
fier EA adjusts this voltage by comparing the feedback
signal VFB from the output voltage with the internal 1.19V
reference. The VPROG pin selects whether the feedback
voltage is taken directly from the VOSENSE pin or is derived
from an on-chip resistive divider. When the load current
increases, it causes a drop in the feedback voltage relative
to the reference. The ITH voltage then rises until the
average inductor current again matches the load current.
An overvoltage comparator OV guards against transient
overshoots and other conditions that may overvoltage the
output. In this case, the top MOSFET is turned off and the
bottom MOSFET is turned on until the overvoltage condi-
tion is cleared.
Foldback current limiting for an output shorted to ground
is provided by a transconductance amplifer CL. As VFB
drops below 0.6V, the buffered ITH input to the current
comparator is gradually pulled down to a 0.95V clamp.
This reduces peak inductor current to about one fifth of its
maximum value.
Low Current Operation
The LTC1625 is capable of Burst Mode operation at low
load currents. If the error amplifier drives the ITH voltage
below 0.95V, the buffered ITH input to the current com-
paratorwillremainclampedat0.95V.Theinductorcurrent
peak is then held at approximately 30mV/RDS(ON)(TOP). If
ITH then drops below 0.5V, the Burst Mode comparator B
will turn off both MOSFETs. The load current will be
supplied solely by the output capacitor until ITH rises
above the 50mV hysteresis of the comparator and switch-
ing is resumed. Burst Mode operation is disabled by
comparator F when the FCB pin is brought below 1.19V.
This forces continuous operation and can assist second-
ary winding regulation.
The internal oscillator can be synchronized to an external
clock applied to the SYNC pin and can lock to a frequency
between 100% and 150% of its nominal 150kHz rate.
When the SYNC pin is left open, it is pulled low internally
and the oscillator runs at its normal rate. If this pin is taken
above 1.2V, the oscillator will run at its maximum 225kHz
rate.
Pulling the RUN/SS pin low forces the controller into its
shutdown state and turns off both MOSFETs. Releasing
the RUN/SS pin allows an internal 3µA current source to
charge up an external soft start capacitor CSS. When this
voltage reaches 1.4V, the controller begins switching, but
with the ITH voltage clamped at approximately 0.8V. As
CSS continuestocharge,theclampisraiseduntilfullrange
operation is restored.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
of the internal circuitry of the LTC1625 is derived from the
INTVCC pin. When the EXTVCC pin is left open, an internal
5.2V low dropout regulator supplies the INTVCC power
from VIN. If EXTVCC is raised above 4.7V, the internal
regulator is turned off and an internal switch connects
EXTVCC to INTVCC. This allows a high efficiency source,
suchastheprimaryorasecondaryoutputoftheconverter
itself, to provide the INTVCC power.
The top MOSFET driver is powered from a floating boot-
strap capacitor CB. This capacitor is normally recharged
from INTVCC through a diode DB when the top MOSFET is
turned off. As VIN decreases towards VOUT, the converter
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LTC1625
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APPLICATIONS INFORMATION
ThebasicLTC1625applicationcircuitisshowninFigure1.
External component selection is primarily determined by
themaximumloadcurrentandbeginswiththeselectionof
the sense resistance and power MOSFETs. Because the
LTC1625usesMOSFETVDS sensing, thesenseresistance
is the RDS(ON) of the MOSFETs. The operating frequency
and the inductor are chosen based largely on the desired
amount of ripple current. Finally, CIN is selected for its
ability to handle the large RMS current into the converter
and COUT is chosen with low enough ESR to meet the
output voltage ripple specification.
The ρT is a normalized term accounting for the significant
variation in RDS(ON) with temperature, typically about
0.4%/°C as shown in Figure 2. Junction to case tempera-
ture TJC is around 10°C in most applications. For a
maximumambienttemperatureof70°C, usingρ80°C 1.3
intheaboveequationisareasonablechoice.Thisequation
is plotted in Figure 3 to illustrate the dependence of
maximum output current on RDS(ON). Some popular
MOSFETs from Siliconix are shown as data points.
2.0
1.5
1.0
0.5
0
Power MOSFET Selection
The LTC1625 requires two external N-channel power
MOSFETs, one for the top (main) switch and one for the
bottom (synchronous) switch. Important parameters for
the power MOSFETs are the breakdown voltage V(BR)DSS
threshold voltage VGS(TH), on-resistance RDS(ON), reverse
transfer capacitance CRSS and maximum current ID(MAX)
,
.
The gate drive voltage is set by the 5.2V INTVCC supply.
Consequently, logic level threshold MOSFETs must be
used in LTC1625 applications. If low input voltage opera-
tion is expected (VIN < 5V), then sub-logic level threshold
MOSFETs should be used. Pay close attention to the
V(BR)DSS specification for the MOSFETs as well; many of
the logic level MOSFETs are limited to 30V or less.
50
100
–50
150
0
JUNCTION TEMPERATURE (°C)
1625 F02
Figure 2. RDS(ON) vs Temperature
10
8
The MOSFET on-resistance is chosen based on the
required load current. The maximum average output cur-
rent IO(MAX) is equal to the peak inductor current less half
the peak-to-peak ripple current ∆IL. The peak inductor
current is inherently limited in a current mode controller
by the current threshold ITH range. The corresponding
maximum VDS sense voltage is about 150mV under nor-
mal conditions. The LTC1625 will not allow peak inductor
current to exceed 150mV/RDS(ON)(TOP). The following
equation is a good guide for determining the required
RDS(ON)(MAX) at 25°C (manufacturer’s specification), al-
lowing some margin for ripple current, current limit and
variationsintheLTC1625andexternalcomponentvalues:
Si4420
Si4410
6
4
Si4412
2
Si9936
0
0
0.02
0.04
0.06
0.08
0.10
R
(Ω)
DS(ON)
1625 F03
Figure 3. Maximum Output Current vs RDS(ON) at VGS = 4.5V
The power dissipated by the top and bottom MOSFETs
strongly depends upon their respective duty cycles and
the load current. When the LTC1625 is operating in con-
tinuous mode, the duty cycles for the MOSFETs are:
120mV
R
DS(ON)(MAX)
I
(
ρ
)(
T
)
O(MAX)
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APPLICATIONS INFORMATION
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7V
V
V
OUT
TopDutyCycle =
IN
V – V
IN
OUT
BottomDutyCycle =
1.2V
V
IN
1µs
4µs
The MOSFET power dissipations at maximum output
current are:
±
1625 F04
0
Figure 4. SYNC Clock Waveform
2
V
V
OUT
P
=
(I
)(ρ )(R
T(TOP) DS(ON)
)
TOP
O(MAX)
IN
Inductor Value Selection
2
+ (k)(V )(I
)(C )(f)
IN O(MAX) RSS
Given the desired input and output voltages, the inductor
value and operating frequency directly determine the
ripple current:
2
V – V
IN
OUT
P
=
(I
)(ρ
)(R
)
BOT
O(MAX)
T(BOT)
DS(ON)
V
IN
V
V
OUT
V
IN
OUT
(f)(L)
∆I =
1–
L
Both MOSFETs have I2R losses and the PTOP equation
includesanadditionaltermfortransitionlosses, whichare
largest at high input voltages. The constant k = 1.7 can be
usedtoestimatetheamountoftransitionloss. Thebottom
MOSFETlossesaregreatestathighinputvoltageorduring
a short circuit when the duty cycle is nearly 100%.
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and output voltage
ripple. Thus, highest efficiency operation is obtained at
low frequency with small ripple current. To achieve this,
however, requires a large inductor.
A reasonable starting point is to choose a ripple current
that is about 40% of IO(MAX). Note that the largest ripple
current occurs at the highest VIN. To guarantee that ripple
current does not exceed a specified maximum, the induc-
tor should be chosen according to:
Operating Frequency and Synchronization
The choice of operating frequency and inductor value is a
trade-off between efficiency and component size. Low
frequency operation improves efficiency by reducing
MOSFET switching losses, both gate charge loss and
transition loss. However, lower frequency operation
requires more inductance for a given amount of ripple
current.
V
V
OUT
OUT
L ≥
1–
(f)(∆I
)
V
IN(MAX)
L(MAX)
Theinternaloscillatorrunsatanominal150kHzfrequency
when the SYNC pin is left open or connected to ground.
Pulling the SYNC pin above 1.2V will increase the fre-
quency by 50%. The oscillator will injection lock to a clock
signal applied to the SYNC pin with a frequency between
165kHz and 200kHz. The clock high level must exceed
1.2V for at least 1µs and no longer than 4µs as shown in
Figure 4. The top MOSFET turn-on will synchronize with
the rising edge of the clock.
Burst Mode Operation Considerations
The choice of RDS(ON) and inductor value also determines
the load current at which the LTC1625 enters Burst Mode
operation. When bursting, the controller clamps the peak
inductor current to approximately:
30mV
I
=
BURST(PEAK)
R
DS(ON)
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Thecorrespondingaveragecurrentdependsontheamount
of ripple current. Lower inductor values (higher ∆IL) will
reduce the load current at which Burst Mode operation
begins.
maximum values for RDS(ON), but not a minimum. A
reasonable, but perhaps overly conservative, assumption
is that the minimum RDS(ON) lies the same amount below
the typical value as the maximum RDS(ON) lies above it.
Consult the MOSFET manufacturer for further guidelines.
The output voltage ripple can increase during Burst Mode
operation if ∆IL is substantially less than IBURST. This will
primarily occur when the duty cycle is very close to unity
(VIN is close to VOUT) or if very large value inductors are
chosen. This is generally only a concern in applications
with VOUT ≥ 5V. At high duty cycles, a skipped cycle
causes the inductor current to quickly descend to zero.
However, it takes multiple cycles to ramp the current back
up to IBURST(PEAK). During this interval, the output capaci-
tor must supply the load current and enough charge may
be lost to cause significant droop in the output voltage. It
The LTC1625 includes current foldback to help further
limit load current when the output is shorted to ground. If
the output falls by more than half, then the maximum
sense voltage is progressively lowered from 150mV to
30mV. Under short-circuit conditions with very low duty
cycle, the LTC1625 will begin skipping cycles in order to
limit the short-circuit current. In this situation the bottom
MOSFET RDS(ON) will control the inductor current trough
rather than the top MOSFET controlling the inductor
current peak. The short-circuit ripple current is deter-
mined by the minimum on-time tON(MIN) of the LTC1625
(approximately 0.5µs), the input voltage, and inductor
value:
is a good idea to keep ∆IL comparable to IBURST(PEAK)
.
Otherwise, one might need to increase the output capaci-
tance in order to reduce the voltage ripple or else disable
Burst Mode operation by forcing continuous operation
with the FCB pin.
∆IL(SC) = tON(MIN) VIN/L.
The resulting short-circuit current is:
Fault Conditions: Current Limit and Output Shorts
30mV
1
2
I
=
+ ∆I
SC
L(SC)
The LTC1625 current comparator can accommodate a
maximum sense voltage of 150mV. This voltage and the
sense resistance determine the maximum allowed peak
inductor current. The corresponding output current limit
is:
R
ρ
T
DS(ON)(BOT)
Normally,thetopandbottomMOSFETswillbeofthesame
type. A bottom MOSFET with lower RDS(ON) than the top
may be chosen if the resulting increase in short-circuit
current is tolerable. However, the bottom MOSFET should
neverbechosentohaveahighernominalRDS(ON) thanthe
top MOSFET.
150mV
1
2
ILIMIT
=
– ∆IL
R
ρ
(
DS(ON))( )
T
The current limit value should be checked to ensure that
ILIMIT(MIN) > IO(MAX). The minimum value of current limit
generally occurs with the largest VIN at the highest ambi-
enttemperature,conditionswhichcausethehighestpower
dissipation in the top MOSFET. Note that it is important to
check for self-consistency between the assumed junction
temperature of the top MOSFET and the resulting value of
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot
affordthecorelossfoundinlowcostpowderedironcores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
the inductance selected. As inductance increases, core
losses go down. Unfortunately, increased inductance
requires more turns of wire and therefore copper losses
will increase.
ILIMIT which heats the junction.
Caution should be used when setting the current limit
based upon RDS(ON) of the MOSFETs. The maximum
current limit is determined by the minimum MOSFET on-
resistance. Data sheets typically specify nominal and
Kool Mµ is a registered trademark of Magnetics, Inc.
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Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can con-
centrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that induc-
tance collapses rapidly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
2000hoursoflife.Thismakesitadvisabletofurtherderate
the capacitor or to choose a capacitor rated at a higher
temperaturethanrequired.Severalcapacitorsmayalsobe
placedinparalleltomeetsizeorheightrequirementsinthe
design.
The selection of COUT is primarily determined by the ESR
required to minimize voltage ripple. The output ripple
∆VOUT is approximately bounded by:
Molypermalloy (from Magnetics, Inc.) is a very good, low
losscorematerialfortoroids,butitismoreexpensivethan
ferrite. A reasonable compromise from the same manu-
facturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, designsforsurfacemountareavailable
which do not increase the height significantly.
1
∆V
≤ ∆I ESR +
L
OUT
(8)(f)(C
)
OUT
Since ∆IL increases with input voltage, the output ripple is
highestatmaximuminputvoltage.Typically,oncetheESR
requirement is satisfied the capacitance is adequate for
filtering and has the required RMS current rating.
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest product of
ESR and size of any aluminum electrolytic at a somewhat
higher price.
Schottky Diode Selection
The Schottky diode D1 shown in Figure 1 conducts during
the dead time between the conduction of the power
MOSFETs. This prevents the body diode of the bottom
MOSFET from turning on and storing charge during the
dead time, which could cost as much as 1% in efficiency.
A 1A Schottky diode is generally a good size for 3A to 5A
regulators. The diode may be omitted if the efficiency loss
can be tolerated.
In surface mount applications, multiple capacitors may
have to be placed in parallel to meet the ESR requirement.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount packages. In the case of
tantalum, it is critical that the capacitors have been surge
tested for use in switching power supplies. An excellent
choice is the AVX TPS series of surface mount tantalum,
availableincaseheightsrangingfrom2mmto4mm.Other
capacitor types include Sanyo OS-CON, Nichicon PL se-
ries, and Sprague 593D and 595D series. Consult the
manufacturer for other specific recommendations.
CIN and COUT Selection
In continuous mode, the drain current of the top MOSFET
is approximately a square wave of duty cycle VOUT/VIN. To
prevent large input voltage transients, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS current is given by:
1/2
INTVCC Regulator
V
V
V
IN
V
OUT
OUT
I
I
−1
RMS O(MAX)
An internal P-channel low dropout regulator produces the
5.2V supply which powers the drivers and internal cir-
cuitry within the LTC1625. The INTVCC pin can supply up
to50mAandmustbebypassedtogroundwithaminimum
of 4.7µF tantalum or low ESR electrolytic capacitance.
Good bypassing is necessary to supply the high transient
currents required by the MOSFET gate drivers.
IN
This formula has a maximum at VIN = 2VOUT, where IRMS
= IO(MAX)/2. This simple worst-case condition is com-
monlyusedfordesignbecauseevensignificantdeviations
do not offer much relief. Note that ripple current ratings
from capacitor manufacturers are often based on only
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High input voltage applications in which large MOSFETs
arebeingdrivenathighfrequenciesmaycausetheLTC1625
to exceed its maximum junction temperature rating. Most
of the supply current drives the MOSFET gates unless an
external EXTVCC source is used. The junction temperature
can be estimated from the equations given in Note 2 of the
Electrical Characteristics. For example, the LTC1625CGN
is limited to less than 14mA from a 30V supply:
3. EXTVCC connectedtoanoutput-derivedboostnetwork.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage which has been boosted to
greater than 4.7V. This can be done with either an
inductive boost winding as shown in Figure 5a or a
capacitive charge pump as shown in Figure 5b.
4. EXTVCC connected to an external supply. If an external
supply isavailable in the 5V to 7Vrange (EXTVCC <VIN),
it may be used to power EXTVCC providing it is compat-
ible with the MOSFET gate drive requirements.
TJ = 70°C + (14mA)(30V)(130°C/W) = 125°C
Topreventthemaximumjunctiontemperaturefrombeing
exceeded, the input supply current must be checked when
operating in continuous mode at high VIN.
V
C
IN
+
EXTVCC Connection
V
SEC
IN
V
IN
The LTC1625 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
Whenever the EXTVCC pin is above 4.7V the internal 5.2V
regulator shuts off, the switch closes and INTVCC power is
supplied via EXTVCC until EXTVCC drops below 4.5V. This
allows the MOSFET gate drive and control power to be
derived from the output or other external source during
normal operation. When the output is out of regulation
(start-up,shortcircuit)powerissuppliedfromtheinternal
regulator. Do not apply greater than 7V to the EXTVCC pin
and ensure that EXTVCC ≤ VIN.
TK
1N4148
TG
•
OPTIONAL
+
LTC1625
EXTV
C
SEC
EXTV
CC
SW
1µF
CC
CONNECTION
V
OUT
R4
R3
5V < V
< 7V
•
SEC
T1
1:N
+
C
FCB
OUT
BG
SGND
PGND
1625 F05a
Figure 5a: Secondary Output Loop and EXTVCC Connection
V
≈ 2(V
– V )
OUT D
PUMP
Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current supplying
the driver and control currents will be scaled by a factor of
DutyCycle/Efficiency.For5Vregulatorsthissimplymeans
connecting the EXTVCC pin directly to VOUT. However, for
3.3V and other lower voltage regulators, additional cir-
cuitry is required to derive INTVCC power from the output.
+
1µF
V
C
IN
+
BAT85
IN
V
IN
BAT85
L1
0.22µF
TK
TG
BAT85
LTC1625
VN2222LL
SW
EXTV
CC
V
OUT
+
The following list summarizes the four possible connec-
tions for EXTVCC:
C
OUT
BG
PGND
1. EXTVCC left open (or grounded). This will cause INTVCC
tobepoweredfromtheinternal5.2Vregulatorresulting
in an efficiency penalty of up to 10% at high input
voltages.
1625 F05b
Figure 5b: Capacitive Charge Pump for EXTVCC
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
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VPROG pin is left open and the VOSENSE pin is connected to
feedback resistors as shown in Figure 6b. The output
voltage is set by the divider as:
Note that RDS(ON) also varies with the gate drive level. If
gate drives other than the 5.2V INTVCC are used, this must
be accounted for when selecting the MOSFET RDS(ON)
.
Particular care should be taken with applications where
EXTVCC is connected to the output. When the output
voltage is between 4.7V and 5.2V, INTVCC will be con-
nected to the output and the gate drive is reduced. The
resulting increase in RDS(ON) will also lower the current
limit. Even applications with VOUT > 5.2V will traverse this
region during start-up and must take into account the
reduced current limit.
R2
R1
V
= 1.19V 1+
OUT
LTC1625
V
OUT
= 5V: INTV
OUT
CC
V
V
PROG
V
= 3.3V:
GND
V
OUT
OSENSE
+
Topside MOSFET Driver Supply (CB, DB)
C
OUT
SGND
An external bootstrap capacitor (CB in the functional
diagram) connected to the BOOST pin supplies the gate
drive voltage for the topside MOSFET. This capacitor is
chargedthroughdiodeDB fromINTVCC whentheSWnode
is low. Note that the voltage across CB is about a diode
drop below INTVCC. When the top MOSFET turns on, the
switch node voltage rises to VIN and the BOOST pin rises
to approximately VIN + INTVCC. During dropout operation,
CB suppliesthetopdriverforaslongastencyclesbetween
refreshes. Thus, the boost capacitance needs to store
about 100 times the gate charge required by the top
MOSFET. In many applications 0.22µF is adequate.
1625 F06a
Figure 6a. Fixed 3.3V or 5V VOUT
LTC1625
OPEN
V
V
PROG
R2
R1
+
C
OUT
OSENSE
SGND
1625 F06b
When adjusting the gate drive level , the final arbiter is the
total input current for the regulator. If you make a change
and the input current decreases, then you improved the
efficiency. If there is no change in input current, then there
is no change in efficiency.
Figure 6b. Adjustable VOUT
Run/Soft Start Function
The RUN/SS pin is a dual purpose pin that provides a soft
startfunctionandameanstoshutdowntheLTC1625.Soft
start reduces surge currents from VIN by gradually in-
creasing the controller’s current limit ITH(MAX). This pin
can also be used for power supply sequencing.
Output Voltage Programming
The LTC1625 has a pin selectable output voltage deter-
mined by the VPROG pin as follows:
VPROG = 0V
VPROG = INTVCC
VPROG = Open
VOUT = 3.3V
Pulling the RUN/SS pin below 1.4V puts the LTC1625 into
alowquiescentcurrentshutdown(IQ<30µA).Thispincan
be driven directly from logic as shown in Figure 7. Releas-
ing the RUN/SS pin allows an internal 3µA current source
to charge up the external capacitor CSS. If RUN/SS has
been pulled all the way to ground there is a delay before
starting of approximately:
VOUT = 5V
VOUT = Adjustable
Remote sensing of the output voltage is provided by the
VOSENSE pin. For fixed 3.3V and 5V output applications an
internal resistive divider is used and the VOSENSE pin is
connected directly to the output voltage as shown in
Figure 6a. When using an external resistive divider, the
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then VSEC will droop. An external resistor divider from
1.4V
3µA
VSEC to the FCB pin sets a minimum voltage VSEC(MIN)
:
t
=
C
= 0.5s/µF C
(
)
DELAY
SS
SS
R4
R3
V
1.19V 1+
When the voltage on RUN/SS reaches 1.4V the LTC1625
begins operating with a clamp on ITH at 0.8V. As the
voltage on RUN/SS increases to approximately 3.1V, the
clamp on ITH is raised until its full 2.4V range is restored.
This takes an additional 0.5s/µF. During this time the load
currentwillbefoldedbacktoapproximately30mV/RDS(ON)
until the output reaches half of its final value.
SEC(MIN)
If VSEC drops below this level, the FCB voltage forces
continuous operation until VSEC is again above its
minimum.
Minimum On-Time Considerations
Diode D1 in Figure 7 reduces the start delay while allowing
Minimum on-time tON(MIN) is the smallest amount of time
that the LTC1625 is capable of turning the top MOSFET on
and off again. It is determined by internal timing delays and
the amount of gate charge required to turn on the top
MOSFET. Low duty cycle applications may approach this
minimum on-time limit and care should be taken to ensure
that:
C
SS to charge up slowly for the soft start function. This
diodeandCSS canbedeletedifsoftstartisnotneeded.The
RUN/SS pin has an internal 6V zener clamp (See Func-
tional Diagram).
3.3V
OR 5V
RUN/SS
RUN/SS
D1
V
C
SS
OUT
C
SS
t
<
ON(MIN)
(V )(f)
1625 F07
IN
Ifthedutycyclefallsbelowwhatcanbeaccommodatedby
the minimum on-time, the LTC1625 will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple current and ripple voltage will increase.
Figure 7. RUN/SS Pin Interfacing
FCB Pin Operation
When the FCB pin drops below its 1.19V threshold,
continuous synchronous operation is forced. In this case,
the top and bottom MOSFETs continue to be driven
regardless of the load on the main output. Burst Mode
operation is disabled and current reversal is allowed in the
inductor.
Theminimumon-timefortheLTC1625isgenerallyabout
0.5µs. However, as the peak sense voltage (IL(PEAK) •
RDS(ON)) decreases, the minimum on-time gradually
increases up to about 0.7µs. This is of particular concern
in forced continuous applications with low ripple current
at light loads. If the duty cycle drops below the minimum
on-time limit in this situation, a significant amount of
cycle skipping can occur with correspondingly larger
current and voltage ripple.
In addition to providing a logic input to force continuous
operation, the FCB pin provides a means to regulate a
flyback winding output. It can force continuous synchro-
nous operation when needed by the flyback winding,
regardless of the primary output load.
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power (×100%). Per-
cent efficiency can be expressed as:
The secondary output voltage VSEC is normally set as
shown in Figure 5a by the turns ratio N of the transformer:
VSEC (N + 1)VOUT
%Efficiency = 100% – (L1 + L2 + L3 + ...)
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current,
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whereL1, L2, etc. aretheindividuallossesasapercentage
of input power. It is often useful to analyze individual
losses to determine what is limiting the efficiency and
which change would produce the most improvement.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1625 circuits:
4. LTC1625 VIN supply current. The VIN current is the DC
supplycurrenttothecontrollerexcludingMOSFETgate
drive current. Total supply current is typically about
850µA. If EXTVCC is connected to 5V, the LTC1625 will
drawonly330µAfromVIN andtheremaining520µAwill
come from EXTVCC. VIN current results in a small
(<1%) loss which increases with VIN.
1. INTVCC current. This is the sum of the MOSFET driver
and control currents. The driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched on and then off,
a packet of gate charge Qg moves from INTVCC to
ground. The resulting current out of INTVCC is typically
much larger than the control circuit current. In continu-
ous mode, IGATECHG = f(Qg(TOP) + Qg(BOT)).
Other losses including CIN and COUT ESR dissipative
losses, Schottky conduction losses during dead time
and inductor core losses, generally account for less
than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. Whenaloadstepoccurs, VOUT immediatelyshifts
by an amount equal to (∆ILOAD)(ESR), where ESR is the
effective series resistance of COUT, and COUT begins to
charge or discharge. The regulator loop acts on the
resultingfeedbackerrorsignaltoreturnVOUT toitssteady-
state value. During this recovery time VOUT can be moni-
tored for overshoot or ringing which would indicate a
stability problem. The ITH pin external components shown
in Figure 1 will provide adequate compensation for most
applications.
By powering EXTVCC from an output-derived source,
the additional VIN current resulting from the driver and
control currents will be scaled by a factor of Duty Cycle/
Efficiency. For example, in a 20V to 5V application at
400mA load, 10mA of INTVCC current results in ap-
proximately 3mA of VIN current. This reduces the loss
from 10% (if the driver was powered directly from VIN)
to about 3%.
2. DC I2R Losses. Since there is no separate sense resis-
tor, DC I2R losses arise only from the resistances of the
MOSFETs and inductor. In continuous mode the aver-
age output current flows through L, but is “chopped”
between the top MOSFET and the bottom MOSFET. If
A second, more severe transient is caused by connecting
loads with large (>1µF) supply bypass capacitors. The
dischargedbypasscapacitorsareeffectivelyputinparallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive in order
to limit the inrush current to the load.
thetwoMOSFETshaveapproximatelythesameRDS(ON)
,
then the resistance of one MOSFET can simply be
summed with the resistance of L to obtain the DC I2R
loss. For example, if each RDS(ON) = 0.05Ω and RL =
0.15Ω, then the total resistance is 0.2Ω. This results in
losses ranging from 2% to 8% as the output current
increases from 0.5A to 2A for a 5V output. I2R losses
cause the efficiency to drop at high output currents.
Automotive Considerations: Plugging into the
Cigarette Lighter
3. Transition losses apply only to the topside MOSFET,
and only when operating at high input voltages (typi-
cally 20V or greater). Transition losses can be esti-
mated from:
As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during opera-
tion. But before you connect, be advised: you are plug-
ging into the supply from hell. The main battery line in an
Transition Loss = (1.7)(VIN2)(IO(MAX))(CRSS)(f)
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automobile is the source of a number of nasty potential
transients, including load dump, reverse and double
battery.
For 40% ripple current at maximum VIN the inductor
should be:
3.3V
(225kHz)(0.4)(2A)
3.3V
22V
Load dump is the result of a loose battery cable. When the
cablebreaksconnection,thefieldcollapseinthealternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse battery is
just what it says, while double battery is a consequence of
tow truck operators finding that a 24V jump start cranks
cold engines faster than 12V.
L ≥
1–
= 16µH
Choosing a standard value of 15µH results in a maximum
ripple current of:
3.3V
(225kHz)(15µH)
3.3V
22V
∆I
=
1–
= 0.83A
L(MAX)
ThenetworkshowninFigure8isthemoststraightforward
approach to protect a DC/DC converter from the ravages
of an automotive battery line. The series diode prevents
current from flowing during reverse battery, while the
transient suppressor clamps the input voltage during load
dump. Note that the transient suppressor should not
conduct during double-battery operation, but must still
clamptheinputvoltagebelowbreakdownoftheconverter.
Although the LTC1625 has a maximum input voltage of
36V, most applications will be limited to 30V by the
Next, check that the minimum value of the current limit is
acceptable. Assume a junction temperature close to a
70°C ambient with ρ80°C = 1.3.
150mV
(0.042Ω)(1.3)
1
2
I
≥
–
0.83A = 2.3A
LIMIT
ThisiscomfortablyaboveIO(MAX)=2A.Nowdouble-check
the assumed TJ:
MOSFET V(BR)DSS
.
3.3V
22V
2
50A I
12V
PK
P
=
(2.3A) (1.3)(0.042Ω)+
TOP
RATING
V
IN
2
(1.7)(22) (2.3A)(180pF)(225kHz)
TRANSIENT VOLTAGE
SUPPRESSOR
GENERAL INSTRUMENT
1.5KA24A
LTC1625
= 43mW + 77mW = 120mW
PGND
TJ = 70°C + (120mW)(50°C/W) = 76°C
1625 F08
Since ρ(76°C) ρ(80°C), the solution is self-consistent.
Figure 8. Automotive Application Protection
A short circuit to ground will result in a folded back
current of:
Design Example
As a design example, take a supply with the following
specifications: VIN = 12V to 22V (15V nominal), VOUT
30mV
(0.03Ω)(1.1)
1 (15V)(0.5µs)
I
=
+
= 1.2A
SC
2
15µH
=
3.3V, IO(MAX) = 2A, and f = 225kHz. The required RDS(ON)
can immediately be estimated:
with a typical value of RDS(ON) and ρ(50°C) = 1.1. The
resulting power dissipated in the bottom MOSFET is:
120mV
(2A)(1.3)
15V – 3.3V
15V
2
R
=
= 0.046Ω
P
=
(1.2A) (1.1)(0.03Ω) = 37mW
DS(ON)
BOT
A 0.042Ω Siliconix Si4412DY MOSFET (θJA = 50°C/W) is
close to this value.
which is less than under full load conditions.
17
LTC1625
APPLICATIONS INFORMATION
U
W U U
V
IN
12V TO 22V
C
IN
+
1
16
22µF
35V
×2
EXTV
V
IN
CC
C
SS
2
3
15
14
M1
0.1µF
INTV
SYNC
TK
CC
Si4412DY
L1
15µH
RUN/SS
SW
C
C1
V
3.3V
2A
LTC1625
OUT
4
5
13
12
R
470pF
C
OPEN
FCB
TG
10k
I
TH
BOOST
C
VCC
C
C2
D
B
C
B
4.7µF
220pF
CMDSH-3
0.1µF
C
6
11
OUT
100µF
10V
0.065Ω
×2
SGND
INTV
CC
+
+
7
8
10
9
M2
Si4412DY
V
V
BG
OSENSE
PGND
PROG
D1
MBRS140T3
1625 F09
C
C
: AVX TPSE226M035R0300
IN
: AVX TPSD107M010R0065
OUT
L1: SUMIDA CDRH125-150MC
Figure 9. 3.3V/2A Fixed Output at 225kHz
CIN is chosen for an RMS current rating of at least 1A at
temperature. COUT is chosen with an ESR of 0.033Ω for
low output ripple. The output ripple in continuous mode
will be highest at the maximum input voltage and is
approximately:
3) The LTC1625 signal ground pin must return to the (–)
plate of COUT. Connect the (–) plate of COUT to power
ground at the source of the bottom MOSFET
4) Keep the switch node SW away from sensitive small-
signal nodes. Ideally the switch node should be placed
on the opposite side of the power MOSFETs from the
LTC1625.
∆VO = (∆IL(MAX))(ESR) = (0.83A)(0.033Ω) = 27mV
The complete circuit is shown in Figure 9.
5) Connect the INTVCC decoupling capacitor CVCC closely
to the INTVCC pin and the power ground pin. This
capacitor carries the MOSFET gate drive current.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1625. These items are also illustrated graphically in
the layout diagram of Figure 10. Check the following in
your layout:
6) Does the VOSENSE pin connect directly to the (+) plate of
COUT? In adjustable applications, the resistive divider
(R1, R2) must be connected between the (+) plate of
COUT and signal ground. Place the divider near the
LTC1625 in order to keep the high impedance VOSENSE
node short.
1) Connect the TK lead directly to the drain of the topside
MOSFET. Then connect the drain to the (+) plate of CIN.
This capacitor provides the AC current to the top
MOSFET.
7) For applications with multiple switching power con-
vertersconnectedtothesameVIN, ensurethattheinput
filtercapacitancefortheLTC1625isnotsharedwiththe
other converters. AC input current from another con-
verter will cause substantial input voltage ripple that
may interfere with proper operation of the LTC1625. A
few inches of PC trace or wire (≈100nH) between CIN
and VIN is sufficient to prevent sharing.
2) Thepowergroundpinconnectsdirectlytothesourceof
thebottomN-channelMOSFET.Thenconnectthesource
to the anode of the Schottky diode and (–) plate of CIN,
which should have as short lead lengths as possible.
18
LTC1625
U
W U U
APPLICATIONS INFORMATION
OPTIONAL 5V EXTV
CC
+
CONNECTION
16
1
EXTV
V
IN
CC
2
3
15
14
EXT
CLK
C
SS
SYNC
TK
M1
RUN/SS
SW
L1
LTC1625
4
5
13
12
OPEN
C
FCB
TG
C
C
B
V
IN
C1
R
I
TH
BOOST
C
D
B
VCC
6
11
SGND
INTV
CC
+
+
R2
7
8
10
9
D1
M2
C
V
BG
IN
OSENSE
OPEN
V
PGND
–
–
PROG
R1
OUTPUT DIVIDER
REQUIRED
V
OUT
C
OUT
+
WITH V
OPEN
PROG
+
1625 F10
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 10. LTC1625 Layout Diagram
U
TYPICAL APPLICATIONS
5V/1.2A Fixed Output at 225kHz
V
IN
5V TO 28V
+
C
IN
1
16
15µF
EXTV
V
IN
CC
C
SS
35V
2
3
15
14
M1
1/2 Si9936DY
L1
39µH
0.1µF
SYNC
RUN/SS
LTC1625
TK
INTV
CC
SW
C
C
V
OUT
4
5
13
12
R
330pF
C
OPEN
FCB
TG
5V
10k
1.2A
I
TH
BOOST
C
VCC
D
B
C
C
4.7µF
B
OUT
CMDSH-3
+
0.1µF
100µF
10V
6
11
SGND
INTV
CC
+
0.100Ω
7
8
10
9
M2
1/2 Si9936DY
V
V
BG
OSENSE
PGND
PROG
1625 TA02
C
C
: AVX TPSD156M035R0300
IN
: AVX TPSD107M010R0100
OUT
L1: SUMIDA CD104-390MC
19
LTC1625
U
TYPICAL APPLICATIONS
2.5V/2.8A Adjustable Output
R
F
4.7Ω
V
IN
5V TO 28V
C
C
IN
F
+
1
16
22µF
35V
×2
EXTV
V
0.1µF
CC
IN
C
SS
2
3
15
14
M1
0.1µF
SYNC
TK
1/2 Si4920DY
RUN/SS
SW
L1
15µH
C
C1
V
2.5V
2.8A
LTC1625
OUT
4
5
13
12
R
1nF
C
OPEN
OPEN
FCB
TG
10k
I
TH
BOOST
R2
11k
1%
C
VCC
D
C
B
C2
C
B
4.7µF
CMDSH-3
330pF
0.22µF
C
6
11
OUT
100µF
10V
0.065Ω
×2
SGND
INTV
CC
+
+
R1
10k
1%
7
8
10
9
M2
V
V
BG
OSENSE
1/2 Si4920DY
PGND
PROG
D1
MBRS140T3
1625 TA03
C
C
: AVX TPSE226M020R0300
IN
: AVX TPSD107M010R0065
OUT
L1: SUMIDA CDRH125-150MC
3.3V/7A Fixed Output
R
F
4.7Ω
V
IN
5V TO 28V
C
IN
C
F
+
1
16
10µF
30V
×3
0.1µF
EXTV
V
IN
CC
C
SS
2
3
15
14
M1
EXT
CLK
0.1µF
SYNC
TK
FDS6680A
L1
RUN/SS
SW
C
7µH
C1
V
LTC1625
OUT
4
5
13
12
R
2.2nF
C
3.3V
7A
FCB
TG
OPEN
10k
I
BOOST
TH
C
VCC
C
D
C2
B
C
B
4.7µF
220pF
CMDSH-3
0.22µF
C
OUT
150µF
6.3V
0.03Ω
×2
6
11
SGND
INTV
CC
+
+
7
8
10
9
M2
FDS6680A
V
V
BG
OSENSE
PGND
PROG
D1
MBRS140T3
1625 TA05
C
C
: SANYO 30SC10M
OUT
IN
: SANYO 6SA150M
20
LTC1625
U
TYPICAL APPLICATIONS
3.3V/4A Fixed Output with 12V/120mA Auxiliary Output
R
F
4.7Ω
V
IN
6V TO 20V
C
C
IN
F
+
10µF
30V
×2
0.1µF
M1
IRLR3103
V
SEC
12V
•
120mA
T1
8µH
C
S
1
16
0.1µF
1:2.53
EXTV
V
IN
CC
R
S
100k
C
•
SS
D
R4
95.3k
1%
R3
11k
1%
S
2
3
15
14
EXT
CLK
0.1µF
SYNC
TK
SM4003TR*
+
C
SEC
3.3µF
RUN/SS
SW
C
B
35V
LTC1625
4
5
13
12
R
0.22µF
C
FCB
TG
M3
10k
NDT410EL
I
TH
BOOST
R1
4.7k
C
VCC
C
C2
V
3.3V
4A
D
OUT
B
C
4.7µF
C1
220pF
CMDSH-3
470pF
6
11
C
OUT
SGND
INTV
D2
CDMSH-3
CC
+
+
100µF
7
8
10
9
10V
C1
0.01µF
M2
IRLR3103
V
V
BG
OSENSE
0.065Ω
×3
PGND
PROG
D1
MBRS140T3
1625 TA04
C
: SANYO 30SC10M
IN
C
C
: AVX TPSD107M010R0065
OUT
SEC
: AVX TAJB335M035R
T1: BH ELECTRONICS 510-1079
*YES! USE A STANDARD RECOVERY DIODE
12V/2.2A Adjustable Output
R
F
4.7Ω
V
IN
12.5V TO 28V
C
IN
C
F
+
1
16
22µF
35V
×2
0.1µF
EXTV
V
IN
CC
C
SS
2
3
15
14
M1
0.1µF
SYNC
TK
Si4412DY
L1
27µH
RUN/SS
SW
C
C
V
12V
2A
LTC1625
OUT
4
5
13
12
R
470pF
C
FCB
TG
22k
I
BOOST
TH
C
R2
35.7k
1%
VCC
D
B
C
B
C
4.7µF
OUT
CMDSH-3
+
0.1µF
68µF
6
11
SGND
INTV
CC
20V
+
R1
3.92k
1%
0.15Ω
×2
7
8
10
9
M2
Si4412DY
V
V
BG
OSENSE
OPEN
PGND
PROG
1625TA06
C
: AVX TPSE226M020R0300
OUT
IN
C
: AVX TPSE686M020R0150
L1: SUMIDA CDRH127-270MC
21
LTC1625
TYPICAL APPLICATIONS
U
–5V/4.5A Positive to Negative Converter
R
F
4.7Ω
V
IN
5V TO 10V
C
F
0.1µF
1
16
EXTV
V
IN
CC
C
+
IN
M1
FDS6670A
2
3
15
14
220µF
SYNC
TK
16V
L1
RUN/SS
SW
6µH
C
C1
2.2nF
C
B
LTC1625
4
5
13
12
C
R
SS
C
0.22µF
FCB
TG
0.1µF
10k
I
TH
BOOST
C
D
D
1
C2
B
220pF
C
CMDSH-3
MBR140T3
+
OUT
6
7
11
10
470µF
SGND
INTV
CC
6.3V
M2
FDS6670A
V
BG
OSENSE
+
C
VCC
4.7µF
V
OUT
8
9
–5V
V
PGND
PROG
4.5A
1625TA08
C
C
: SANYO 16SV220M
OUT
L1: MAGNETICS Kool-Mµ 77120-A7, 9 TURNS, 17 GAUGE
IN
: SANYO 6SV470M
Single Inductor, Positive Output Buck Boost
R
F
4.7Ω
V
V
I
IN
IN
OUT
6V TO 18V
C
F
18
12
6
4.0
3.3
2.0
0.1µF
C
IN
+
1
16
68µF
20V
x2
EXTV
V
IN
CC
M1
C
SS
2
3
15
14
D2
Si4420DY
0.1µF
SYNC
TK
MBRS340T3
L1
RUN/SS
SW
18µH
4
5
13
12
V
OUT
FCB
TG
C
12V
B
LTC1625
R
C
0.33µF
10k
M4
Si4425DY
I
BOOST
C1
TH
R1
100k
C
470pF
C2
D
C
OUT
100µF
16V
B
D3
BAT85
C
C1
220pF
+
CMDSH-3
2.2nF
6
7
11
10
SGND
INTV
CC
30mΩ
x2
M2
8
2
M3
V
BG
Si4420DY
1
7
OSENSE
Z1
R1
3.92k
Si4420DY
+
D1
MBRS
340T3
MMBZ
5240
10V
C
VCC
1/2
4.7µF
LTC1693-2
8
9
V
PGND
PROG
R2
35.7k
C2
4
6
0.1µF
5
3
D4
BAT85
D5
BAT85
1/2
LTC1693-2
1625TA09
C
C
: SANYO 20S68M
OUT
L1: 7A, 18µH Kool-Mµ 77120-A7, 15 TURNS, 17 GAUGE
IN
: SANYO 16SA100M
22
LTC1625
U
Dimensions in inches (millimeters) unless otherwise noted.
PACKAGE DESCRIPTION
GN Package
16-Lead Plastic SSOP (Narrow 0.150)
(LTC DWG # 05-08-1641)
0.189 – 0.196*
(4.801 – 4.978)
0.009
(0.229)
REF
16 15 14 13 12 11 10 9
0.229 – 0.244
(5.817 – 6.198)
0.150 – 0.157**
(3.810 – 3.988)
1
2
3
4
5
6
7
8
0.015 ± 0.004
(0.38 ± 0.10)
× 45°
0.053 – 0.068
(1.351 – 1.727)
0.004 – 0.0098
(0.102 – 0.249)
0.007 – 0.0098
(0.178 – 0.249)
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.008 – 0.012
(0.203 – 0.305)
0.025
(0.635)
BSC
*
DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
GN16 (SSOP) 0398
** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
S Package
16-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.386 – 0.394*
(9.804 – 10.008)
16
15
14
13
12
11
10
9
0.150 – 0.157**
0.228 – 0.244
(3.810 – 3.988)
(5.791 – 6.197)
5
7
8
1
2
3
4
6
0.010 – 0.020
(0.254 – 0.508)
× 45°
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0.008 – 0.010
(0.203 – 0.254)
0° – 8° TYP
0.050
(1.270)
TYP
0.014 – 0.019
(0.355 – 0.483)
0.016 – 0.050
0.406 – 1.270
S16 0695
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
23
LTC1625
U
TYPICAL APPLICATION
3.3V/1.8A Fixed Output
V
IN
5V TO 28V
C
1
16
IN
+
EXTV
CC
V
IN
15µF
35V
×2
C
SS
2
3
15
14
M1
0.1µF
SYNC
TK
1/2 Si4936DY
RUN/SS
SW
L1
C
27µH
C1
V
LTC1625
OUT
4
5
13
12
R
1nF
C
OPEN
FCB
TG
3.3V
1.8A
10k
I
TH
BOOST
C
VCC
C
D
B
CMDSH-3
C2
C
4.7µF
B
C
100pF
OUT
0.1µF
6
11
+
100µF
10V
SGND
INTV
CC
+
7
8
10
9
0.1Ω
×2
M2
1/2 Si4936DY
V
V
BG
OSENSE
PGND
PROG
D1
MBRS140T3
1625 TA07
C
C
: AVX TPSD156M035R0300
IN
: AVX TPSD107M010R0100
OUT
L1: SUMIDA CDRH125-270MC
RELATED PARTS
PART NUMBER
LTC1435A
DESCRIPTION
High Efficiency Synchronous Step-Down Controller
COMMENTS
Optimized for Low Duty Cycle Battery to CPU Power Applications
PLL Synchronization and Auxiliary Linear Regulator
Power-On Reset and Low-Battery Comparator
LTC1436A-PLL
LTC1438
High Efficiency Low Noise Synchronous Step-Down Controller
Dual High Efficiency Step-Down Controller
LTC1530
High Power Synchronous Step-Down Controller
SO-8 with Current Limit, No R
Frequency Ideal for 5V to 3.3V
Saves Space, Fixed
SENSE
LTC1538-AUX
LTC1649
Dual High Efficiency Step-Down Controller
3.3V Input High Power Step-Down Controller
5V Standby Output and Auxiliary Linear Regulator
2.7V to 5V Input, 90% Efficiency, Ideal for 3.3V to 1.xV – 2.xV
Up to 20A
1625f LT/TP 1298 4K • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1998
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
24
●
●
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
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