LTC1872ES6#TRMPBF [Linear]
LTC1872 - Constant Frequency Current Mode Step-Up DC/DC Controller in SOT-23; Package: SOT; Pins: 6; Temperature Range: -40°C to 85°C;型号: | LTC1872ES6#TRMPBF |
厂家: | Linear |
描述: | LTC1872 - Constant Frequency Current Mode Step-Up DC/DC Controller in SOT-23; Package: SOT; Pins: 6; Temperature Range: -40°C to 85°C 控制器 |
文件: | 总12页 (文件大小:183K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
LTC1872
Constant Frequency
Current Mode Step-Up
DC/DC Controller in SOT-23
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FEATURES
DESCRIPTIO
The LTC®1872 is a constant frequency current mode step-
up DC/DC controller providing excellent AC and DC load
and line regulation. The device incorporates an accurate
undervoltagelockoutfeaturethatshutsdowntheLTC1872
when the input voltage falls below 2.0V.
■
High Efficiency: Over 90%
■
High Output Currents Easily Achieved
■
Wide VIN Range: 2.5V to 9.8V
■
VOUT Limited Only by External Components
■
Constant Frequency 550kHz Operation
Burst ModeTM Operation at Light Load
■
TheLTC1872boastsa±2.5%outputvoltageaccuracyand
consumes only 270µA of quiescent current. For applica-
tionswhereefficiencyisaprimeconsideration,theLTC1872
is configured for Burst Mode operation, which enhances
efficiency at low output current.
■
Current Mode Operation for Excellent Line and Load
Transient Response
Low Quiescent Current: 270µA
Shutdown Mode Draws Only 8µA Supply Current
±2.5% Reference Accuracy
Tiny 6-Lead SOT-23 Package
■
■
■
■
In shutdown, the device draws a mere 8µA. The high
550kHz constant operating frequency allows the use of a
small external inductor.
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APPLICATIO S
The LTC1872 is available in a small footprint 6-lead
SOT-23.
■
Lithium-Ion-Powered Applications
■
Cellular Telephones
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
■
Wireless Modems
■
Portable Computers
■
Scanners
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TYPICAL APPLICATION
V
IN
Efficiency vs Load Current
3.3V
C1
10µF
10V
R1
100
95
90
85
80
75
70
65
0.03Ω
V
V
= 3.3V
= 5V
IN
OUT
147k
1
5
L1
4.7µH
I
/RUN
V
TH
IN
–
LTC1872
220pF
80.6k
V
5V
1A
OUT
2
3
4
6
GND
SENSE
NGATE
+
C2
2× 22µF
6.3V
D1
M1
V
FB
422k
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: MURATA GRM42-2X5R226K6.3
D1: IR10BQ015
L1: MURATA LQN6C4R7M04
M1: IRLMS2002
R1: DALE 0.25W
1872 TA01
1
10
100
1000
LOAD CURRENT (mA)
Figure 1. LTC1872 High Output Current 3.3V to 5V Boost Converter
1872 TA01b
1
LTC1872
W W
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
Input Supply Voltage (VIN).........................–0.3V to 10V
SENSE–, NGATE Voltages ............ –0.3V to (VIN + 0.3V)
VFB, ITH/RUN Voltages ..............................–0.3V to 2.4V
NGATE Peak Output Current (<10µs) ....................... 1A
Storage Ambient Temperature Range ... –65°C to 150°C
Operating Temperature Range (Note 2) .. –40°C to 85°C
Junction Temperature (Note 3)............................. 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
LTC1872ES6
I
/RUN 1
GND 2
6 NGATE
5 V
TH
IN
4 SENSE
–
V
3
FB
S6 PART MARKING
LTMK
S6 PACKAGE
6-LEAD PLASTIC SOT-23
TJMAX = 150°C, θJA = 230°C/ W
Consult factory for parts specified with wider operating temperature ranges.
The ● denotes specifications that apply over the full operating temperature
ELECTRICAL CHARACTERISTICS
range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input DC Supply Current
Normal Operation
Sleep Mode
Shutdown
UVLO
Typicals at V = 4.2V (Note 4)
IN
2.4V ≤ V ≤ 9.8V
270
230
8
420
370
22
µA
µA
µA
µA
IN
2.4V ≤ V ≤ 9.8V
IN
2.4V ≤ V ≤ 9.8V, V /RUN = 0V
IN ITH
V
< UVLO Threshold
6
10
IN
Undervoltage Lockout Threshold
V
V
Falling
Rising
●
●
1.55
1.85
2.00
2.10
2.35
2.40
V
V
IN
IN
Shutdown Threshold (at I /RUN)
0.15
0.25
0.35
0.5
0.55
0.85
V
TH
Start-Up Current Source
V
/RUN = 0V
ITH
µA
Regulated Feedback Voltage
0°C to 70°C(Note 5)
–40°C to 85°C(Note 5)
●
●
0.780
0.770
0.800
0.800
0.820
0.830
V
V
V
Input Current
(Note 5)
10
550
40
50
nA
kHz
ns
FB
Oscillator Frequency
Gate Drive Rise Time
Gate Drive Fall Time
V
C
C
= 0.8V
500
114
650
FB
= 3000pF
= 3000pF
LOAD
LOAD
40
ns
Peak Current Sense Voltage
(Note 6)
120
mV
Note 1: Absolute Maximum Ratings are those values beyond which the life
Note 4: Dynamic supply current is higher due to the gate charge being
of a device may be impaired.
delivered at the switching frequency.
Note 2: The LTC1872E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 5: The LTC1872 is tested in a feedback loop that servos V to the
output of the error amplifier.
Note 6: Guaranteed by design at duty cycle = 30%. Peak current sense
FB
voltage is V /6.67 at duty cycle <40%, and decreases as duty cycle
REF
Note 3: T is calculated from the ambient temperature T and power
increases due to slope compensation as shown in Figure 2.
J
A
dissipation P according to the following formula:
D
T = T + (P • θ °C/W)
J
A
D
JA
2
LTC1872
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TYPICAL PERFORMANCE CHARACTERISTICS
Undervoltage Lockout Trip
Voltage vs Temperature
Reference Voltage
vs Temperature
Normalized Oscillator Frequency
vs Temperature
825
820
815
810
805
800
795
790
785
780
775
10
8
2.24
2.20
2.16
2.12
2.08
2.04
2.00
1.96
1.92
1.88
1.84
V
IN
= 4.2V
V
IN
= 4.2V
V
IN
FALLING
6
4
2
0
–2
–4
–6
–8
–10
–55 –35 –15
5
25 45 65 85 105 125
–55 –35 –15
5
25 45 65 85 105 125
–55 –35 –15
5
25 45 65 85 105 125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
1872 G01
1872 G02
1872 G03
Maximum Current Sense Trip
Voltage vs Duty Cycle
Shutdown Threshold
vs Temperature
600
560
520
480
440
400
360
320
280
240
200
130
120
110
100
90
V
IN
= 4.2V
V
A
= 4.2V
IN
T
= 25°C
80
70
60
50
60 70
–55 –35 –15
5
45
85 105 125
20 30 40 50
80 90 100
25
65
TEMPERATURE (°C)
DUTY CYCLE (%)
1872 G04
1872 G05
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PIN FUNCTIONS
ITH/RUN (Pin 1): This pin performs two functions. It
servesastheerroramplifiercompensationpointaswellas
the run control input. Nominal voltage range for this pin is
0.7V to 1.9V. Forcing this pin below 0.35V causes the
device to be shut down. In shutdown all functions are
disabled and the NGATE pin is held low.
SENSE– (Pin 4): The Negative Input to the Current Com-
parator.
VIN (Pin5):SupplyPin. MustbecloselydecoupledtoGND
Pin 2.
NGATE (Pin 6): Gate Drive for the External N-Channel
MOSFET. This pin swings from 0V to VIN.
GND (Pin 2): Ground Pin.
V
FB (Pin 3): Receives the feedback voltage from an exter-
nal resistive divider across the output.
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LTC1872
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FUNCTIONAL DIAGRA
–
V
SENSE
4
IN
5
+
–
ICMP
V
IN
RS
NGATE
6
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
SLOPE
COMP
R
OSC
Q
S
–
+
FREQ
FOLDBACK
BURST
CMP
OVP
+
–
+
–
0.3V
V
+
SLEEP
REF
0.15V
60mV
V
IN
EAMP
V
REF
+
–
0.8V
0.5µA
V
FB
I
/RUN
1
3
+
TH
V
IN
V
IN
0.3V
–
0.35V
+
SHDN
UV
SHDN
CMP
VOLTAGE
REFERENCE
V
REF
0.8V
–
GND
2
UNDERVOLTAGE
LOCKOUT
1.2V
1872FD
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(Refer to Functional Diagram)
OPERATIO
Main Control Loop
0.8V reference, which in turn causes the
ITH/RUN voltage to increase until the average inductor
current matches the new load current.
TheLTC1872isaconstantfrequencycurrentmodeswitch-
ing regulator. During normal operation, the external
N-channel power MOSFET is turned on each cycle by the
oscillator and turned off when the current comparator
(ICMP) resets the RS latch. The peak inductor current at
whichICMPresetstheRSlatchiscontrolledbythevoltage
on the ITH/RUN pin, which is the output of the error
amplifier EAMP. An external resistive divider connected
between VOUT and ground allows the EAMP to receive an
output feedback voltage VFB. When the load current in-
creases, it causes a slight decrease in VFB relative to the
ThemaincontrolloopisshutdownbypullingtheITH/RUN
pin low. Releasing ITH/RUN allows an internal 0.5µA
current source to charge up the external compensation
network. When the ITH/RUN pin reaches 0.35V, the main
control loop is enabled with the ITH/RUN voltage then
pulled up to its zero current level of approximately 0.7V.
Astheexternalcompensationnetworkcontinuestocharge
up, the corresponding output current trip level follows,
allowing normal operation.
4
LTC1872
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(Refer to Functional Diagram)
OPERATIO
Comparator OVP guards against transient overshoots
>7.5% by turning off the external N-channel power
MOSFET and keeping it off until the fault is removed.
Overvoltage Protection
The overvoltage comparator in the LTC1872 will turn the
external MOSFET off when the feedback voltage has risen
7.5% above the reference voltage of 0.8V. This compara-
tor has a typical hysteresis of 20mV.
Burst Mode Operation
The LTC1872 enters Burst Mode operation at low load
currents. In this mode, the peak current of the inductor is
set as if VITH/RUN = 1V (at low duty cycles) even though
the voltage at the ITH/RUN pin is at a lower value. If the
inductor’saveragecurrentisgreaterthantheloadrequire-
ment, the voltage at the ITH/RUN pin will drop. When the
ITH/RUN voltage goes below 0.85V, the sleep signal goes
high, turning off the external MOSFET. The sleep signal
goes low when the ITH/RUN voltage goes above 0.925V
and the LTC1872 resumes normal operation. The next
oscillator cycle will turn the external MOSFET on and the
switching cycle repeats.
Slope Compensation and Inductor’s Peak Current
The inductor’s peak current is determined by:
V
ITH − 0.7
IPK
=
10 RSENSE
(
)
when the LTC1872 is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope com-
pensation begins and effectively reduces the peak induc-
torcurrent. Theamountofreductionisgivenbythecurves
in Figure 2.
Undervoltage Lockout
Short-Circuit Protection
TopreventoperationoftheN-channelMOSFETbelowsafe
input voltage levels, an undervoltage lockout is incorpo-
rated into the LTC1872. When the input supply voltage
drops below approximately 2.0V, the N-channel MOSFET
and all circuitry is turned off except the undervoltage
block, which draws only several microamperes.
Sincethepowerswitchinaboostconverterisnotinseries
with the power path from input to load, turning off the
switch provides no protection from a short-circuit at the
output. External means such as a fuse in series with the
boost inductor must be employed to handle this fault
condition.
110
100
90
80
70
60
50
I
= 0.4I
PK
RIPPLE
AT 5% DUTY CYCLE
= 0.2I
40
30
20
10
I
RIPPLE
PK
AT 5% DUTY CYCLE
V
IN
= 4.2V
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
1872 F02
Figure 2. Maximum Output Current vs Duty Cycle
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LTC1872
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APPLICATIONS INFORMATION
The basic LTC1872 application circuit is shown in
Figure 1. External component selection is driven by the
load requirement and begins with the selection of L1 and
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripplecurrent. However, thisisattheexpenseofefficiency
due to an increase in MOSFET gate charge losses.
R
SENSE (= R1). Next, the power MOSFET and the output
diodeD1isselectedfollowedbyCIN(=C1)andCOUT(=C2).
RSENSE Selection for Output Current
The inductance value also has a direct effect on ripple
current. The ripple current, IRIPPLE, decreases with higher
inductance or frequency and increases with higher VOUT
The inductor’s peak-to-peak ripple current is given by:
RSENSE is chosen based on the required output current.
Withthecurrentcomparatormonitoringthevoltagedevel-
oped across RSENSE, the threshold of the comparator
determines the inductor’s peak current. The output cur-
rent the LTC1872 can provide is given by:
.
V
VOUT + VD − V
IN
IN
IRIPPLE
=
VOUT + VD
f L
( )
0.12
RSENSE
IRIPPLE
V
IN
VOUT + VD
IOUT
=
−
2
wherefistheoperatingfrequency.Acceptinglargervalues
of IRIPPLE allows the use of low inductances, but results in
higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is:
where IRIPPLE is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section) and VD is the
forward drop of the output diode at the full rated output
current.
VOUT + VD
I
RIPPLE = 0.4 IOUT MAX
(
)
A reasonable starting point for setting ripple current is:
V
IN
VOUT + VD
In Burst Mode operation, the ripple current is normally set
such that the inductor current is continuous during the
burst periods. Therefore, the peak-to-peak ripple current
must not exceed:
I
RIPPLE = O.4 IOUT
(
)(
)
V
IN
Rearranging the above equation, it becomes:
1
V
0.03
RSENSE
IN
RSENSE
=
IRIPPLE
≤
VOUT + VD
IOUT
10
( )(
)
for Duty Cycle < 40%
This implies a minimum inductance of:
However,foroperationthatisabove40%dutycycle,slope
compensation’s effect has to be taken into consideration
to select the appropriate value to provide the required
amount of current. Using the scaling factor (SF, in %) in
Figure 2, the value of RSENSE is:
V
VOUT + VD − V
IN
IN
LMIN
=
VOUT + VD
0.03
RSENSE
f
A smaller value than LMIN could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
V
SF
IN
RSENSE
=
VOUT + VD
10 IOUT 100
( )(
)(
)
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APPLICATIONS INFORMATION
Itisimportanttoadequatelyspecifythediodepeakcurrent
and average power dissipation so as not to exceed the
diode ratings.
Inductor Selection
When selecting the inductor, keep in mind that inductor
saturation current has to be greater than the current limit
set by the current sense resistor. Also, keep in mind that
the DC resistance of the inductor will affect the efficiency.
OfftheshelfinductorsareavailablefromMurata,Coilcraft,
Toko, Panasonic, Coiltronics and many other suppliers.
Schottky diodes are recommended for low forward drop
and fast switching times. Remember to keep lead length
short and observe proper grounding (see Board Layout
Checklist) to avoid ringing and increased dissipation.
Power MOSFET Selection
CIN and COUT Selection
The main selection criteria for the power MOSFET are the
To prevent large input voltage ripple, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current for a boost
converter is approximately equal to:
threshold voltage VGS(TH), the “on” resistance RDS(ON)
,
reverse transfer capacitance CRSS and total gate charge.
Since the LTC1872 is designed for operation down to low
input voltages, a logic level threshold MOSFET (RDS(ON)
guaranteed at VGS = 2.5V) is required for applications that
workclosetothisvoltage.WhentheseMOSFETsareused,
makesurethattheinputsupplytotheLTC1872islessthan
the absolute maximum VGS rating, typically 8V.
CIN Required IRMS ≈ 0.3 IRIPPLE
(
)
where IRIPPLE is as defined in the Inductor Value Calcula-
tion section.
Note that capacitor manufacturer’s ripple current ratings
are often based on 2000 hours of life. This makes it
advisable to further derate the capacitor, or to choose a
capacitor rated at a higher temperature than required.
Several capacitors may be paralleled to meet the size or
height requirements in the design. Due to the high operat-
ingfrequencyoftheLTC1872,ceramiccapacitorscanalso
be used for CIN. Always consult the manufacturer if there
is any question.
The required minimum RDS(ON) of the MOSFET is gov-
erned by its allowable power dissipation given by:
PP
RDS(ON)
2
1+ δp
DC I
(
)
(
)
IN
where PP is the allowable power dissipation and δp is the
temperature dependency of RDS(ON). (1 + δp) is generally
given for a MOSFET in the form of a normalized RDS(ON) vs
temperature curve, but δp = 0.005/°C can be used as an
approximation for low voltage MOSFETs. DC is the maxi-
mum operating duty cycle of the LTC1872.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:
Output Diode Selection
VOUT + VD IRIPPLE
Under normal load conditions, the average current con-
ducted by the diode in a boost converter is equal to the
output load current:
∆VOUT ≈ IO •
+
•
V
IN
2
1
2
2
1
ID(avg) = IOUT
ESR2 +
2πfCOUT
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APPLICATIONS INFORMATION
where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the induc-
tor.
Setting Output Voltage
The LTC1872 develops a 0.8V reference voltage between
thefeedback(Pin3)terminalandground(seeFigure4).By
selecting resistor R1, a constant current is caused to flow
through R1 and R2 to set the overall output voltage. The
regulated output voltage is determined by:
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higherprice.TheoutputcapacitorRMScurrentisapproxi-
mately equal to:
R2
R1
VOUT = 0.8V 1+
105
V
IPK • DC −DC2
REF
100
95
90
85
80
75
where IPK is the peak inductor current and DC is the switch
duty cycle.
V
ITH
Whenusingelectrolyticoutputcapacitors, iftherippleand
ESR requirements are met, there is likely to be far more
capacitance than required.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surface mount configurations. An excellent choice of
tantalum capacitors is the AVX TPS and KEMET T510
series of surface mount tantalum capacitors. Also,
ceramic capacitors in X5R pr X7R dielectrics offer excel-
lent performance.
2.0
2.2
2.4
2.6
2.8
3.0
INPUT VOLTAGE (V)
1872 F03
Figure 3. Line Regulation of VREF and VITH
V
OUT
R2
R1
LTC1872
3
V
FB
Low Supply Operation
1872 F04
Although the LTC1872 can function down to approxi-
mately 2.0V, the maximum allowable output current is
reducedwhenVIN decreasesbelow3V. Figure3showsthe
amount of change as the supply is reduced down to 2V.
Also shown in Figure 3 is the effect of VIN on VREF as VIN
goes below 2.3V.
Figure 4. Setting Output Voltage
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APPLICATIONS INFORMATION
Formostapplications, an80kresistorissuggestedforR1.
To prevent stray pickup, locate resistors R1 and R2 close
to LTC1872.
2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each time
a MOSFET gate is switched from low to high to low
again, a packet of charge, dQ, moves from VIN to
ground. The resulting dQ/dt is a current out of VIN
which is typically much larger than the contoller’s DC
supply current. In continuous mode, IGATECHG = f(Qp).
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current sense resistor. The
MOSFET RDS(ON) multiplied by duty cycle times the
average output current squared can be summed with
I2RlossesintheinductorESRinserieswiththecurrent
sense resistor.
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
oftenusefultoanalyzeindividuallossestodeterminewhat
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (η1 + η2 + η3 + ...)
where η1, η2, etc. are the individual losses as a percent-
age of input power.
4. The output diode is a major source of power loss at
high currents. The diode loss is calculated by multiply-
ing the forward voltage by the load current.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1872 circuits: 1) LTC1872 DC bias current,
2) MOSFET gate charge current, 3) I2R losses and 4)
voltage drop of the output diode.
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
1. The VIN current is the DC supply current, given in the
electricalcharacteristics, thatexcludesMOSFETdriver
and control currents. VIN current results in a small loss
which increases with VIN.
Transition Loss = 2(VIN)2IIN(MAX) RSS
(f)
C
Other losses, including CIN and COUT ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
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APPLICATIONS INFORMATION
PC Board Layout Checklist
4. Connect the end of RSENSE as close to VIN (Pin 5) as
possible. The VIN pin is the SENSE+ of the current
comparator.
5. The trace from SENSE– (Pin 4) to the Sense resistor
should be kept short. The trace should connect close
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1872. These items are illustrated graphically in the
layout diagram in Figure 5. Check the following in your
layout:
to RSENSE
.
1. The Schottky diode should be closely connected
between the output capacitor and the drain of the
external MOSFET.
6. Keep the switching node NGATE away from sensitive
small signal nodes.
7. The VFB pin should connect directly to the feedback
resistors. The resistive divider R1 and R2 must be
connected between the (+) plate of COUT and signal
ground.
2. The (+) plate of CIN should connect to the sense
resistor as closely as possible. This capacitor provides
AC current to the inductor.
3. The input decoupling capacitor (0.1µF) should be
connected closely between VIN (Pin 5) and ground
(Pin 2).
V
IN
1
2
3
6
5
4
I
/RUN NGATE
LTC1872
TH
M1
L1
R
S
R
ITH
GND
V
IN
+
0.1µF
C
IN
D1
–
V
SENSE
C
ITH
FB
V
OUT
+
R2
C
OUT
R1
BOLD LINES INDICATE HIGH CURRENT PATHS
1872 F05
Figure 5. LTC1872 Layout Diagram (See PC Board Layout Checklist)
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LTC1872
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TYPICAL APPLICATIO
LTC1872 12V/500mA Boost Converter
V
IN
3V TO 9.8V
C1
10µF
10V
R1
0.033Ω
1
5
L1
I
/RUN
V
TH
IN
–
10µH
LTC1872
10k
V
2
3
4
6
OUT
GND
SENSE
NGATE
12V
+
C2
47µF
16V
220pF
D1
M1
V
FB
1.1M
1872 TA02
78.7k
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: AVX TPSE476M016R0150
D1: IR10BQ015
L1: COILTRONICS UP2B-100
M1: Si9804DV
R1: DALE 0.25W
LTC1872 Three-Cell White LED Driver
V
= 3 AA CELLS ≈ 2.7V TO 4.8V
IN
C1
R1
0.27Ω
10µF
10V
AA
AA
AA
1
5
L1
I
/RUN
V
TH
IN
150µH
V
≈ 28.8V
OUT
LTC1872
10k
220pF
(WITH 8 LEDs)
2
3
4
6
–
GND
SENSE
NGATE
+
C2
15µF
35V
C3
15mA
D0
M1
V
0.1µF
CERAMIC
FB
D1
D2
1 TO 8
WHITE
LEDs
•
•
•
D8
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: AVX TPSD156M035R0300
D0: MOTOROLA MBR0540
L1: COILCRAFT DO1608C-154
M1: Si9804
R1: DALE 0.25W
53.6Ω
1872 TA04
D1-D7: CMD333UWC
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
11
LTC1872
U
TYPICAL APPLICATIO
LTC1872 –2.5V to 3.3V/0.5A Boost Converter
LTC1872 2.7V to 9.8V Input
to 3.3V/1.2A Output SEPIC Converter
R1
0.034Ω
C2
V
IN
2× 100µF
2.7V TO 9.8V
C
+
IN
R
CS
0.03Ω
1
5
10V
10µF
L1
4.7µH
I
/RUN
V
IN
C
C1
TH
10V, X5R
R
C1
10k
220pF
LTC1872
10k
V
3.3V
0.5A
V
1
5
OUT
OUT
2
3
4
6
I
/RUN
V
–
TH
IN
–
3.3V/1.2A
GND
SENSE
NGATE
+
C01
180µF
4V, SP
D1
MBRM120
L1A
L1B
220pF
C1
LTC1872
D1
332k
V
M1
FB
2
3
4
6
GND
SENSE
NGATE
CS
4.7µF
10V
V
FB
R
f2
80.6k
+
U1
0.1µF
CERAMIC
100µF
10V
R
f1
252k
M1
1872 TA05
80.6k
180k
C
01
, CS; TOKO, MURATA OR TAIYO YUDEN
: PANASONIC EEFUE0G181R
FOR V
= 5V CHANGE
IN
OUT
V
IN
–2.5V
C
R
TO 427kΩ AND
f1
1872 TA03
L1: BH ELECTRONICS 511-1012
M1: IRLMS2002
C
TO 150µF, 6V PANASONIC
01
C1, C2: AVX TPSE107M010R0100
D1: MOTOROLA MBR2045CT
L1: COILTRONICS UP2B-4R7
M1: Si9804DV
SP TYPE CAPACITOR
R1: DALE 0.25W
R
: DALE OR IRC
CS
U1: PANASONIC 2SB709A
U
PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted.
S6 Package
6-Lead Plastic SOT-23
(LTC DWG # 05-08-1634)
2.6 – 3.0
(0.110 – 0.118)
2.80 – 3.00
(0.110 – 0.118)
(NOTE 3)
1.50 – 1.75
(0.059 – 0.069)
0.00 – 0.15
(0.00 – 0.006)
0.90 – 1.45
(0.035 – 0.057)
0.35 – 0.55
(0.014 – 0.022)
0.35 – 0.50
(0.014 – 0.020)
SIX PLACES (NOTE 2)
0.90 – 1.30
(0.035 – 0.051)
0.95
(0.037)
REF
0.09 – 0.20
(0.004 – 0.008)
(NOTE 2)
1.90
(0.074)
REF
NOTE:
S6 SOT-23 0898
1. DIMENSIONS ARE IN MILLIMETERS
2. DIMENSIONS ARE INCLUSIVE OF PLATING
3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
4. MOLD FLASH SHALL NOT EXCEED 0.254mm
5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ)
RELATED PARTS
PART NUMBER
LT1304
DESCRIPTION
COMMENTS
Micropower DC/DC Converter with Low-Battery Detector
1.7MHz, Single Cell Micropower DC/DC Converter
1.4MHz, Single Cell DC/DC Converter in 5-Lead SOT-23
Low Voltage Current Mode PWM Controller
High Power DC/DC Step-Up Controller
120µA Quiescent Current, 1.5V ≤ V ≤ 8V
IN
LT1610
30µA Quiescent Current, V Down to 1V
IN
LT1613
Internally Compensated, V Down to 1V
IN
LT1619
8-Lead MSOP Package, 1.9V ≤ V ≤ 18V
IN
LT1680
Operation Up to 60V, Fixed Frequency Current Mode
8-Pin N-Channel Drive, 3.5V ≤ V ≤ 36V
LTC1624
LT1615
High Efficiency SO-8 N-Channel Switching Regulator Controller
Micropower Step-Up DC/DC Converter in SOT-23
IN
20µA Quiescent Current, V Down to 1V
IN
LTC1700
LTC1772
LTC3401/LTC3402
No R
Synchronous Current Mode DC/DC Step-Up Controller
95% Efficient, 0.9V ≤ V ≤ 5V, 550kHz Operation
IN
SENSE
Constant Frequency Current Mode Step-Down DC/DC Controller
1A/2A, 3MHz Micropower Synchronous Boost Converter
V
2.5V to 9.8V, I
up to 4A, SOT-23 Package
OUT
IN
10-Lead MSOP Package, 0.5V ≤ V ≤ 5V
IN
1872f LT/TP 0301 4K • PRINTED IN USA
12 LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
●
●
LINEAR TECHNOLOGY CORPORATION 2000
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
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