LTC1872_技术文档

LTC1872

Constant Frequency

Current Mode Step-Up

DC/DC Controller in SOT-23

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FEATURES

DESCRIPTIO

The LTC^®1872 is a constant frequency current mode step-

up DC/DC controller providing excellent AC and DC load

and line regulation. The device incorporates an accurate

undervoltagelockoutfeaturethatshutsdowntheLTC1872

when the input voltage falls below 2.0V.

■

High Efficiency: Over 90%

■

High Output Currents Easily Achieved

■

Wide V_INRange: 2.5V to 9.8V

■

V_OUTLimited Only by External Components

■

Constant Frequency 550kHz Operation

Burst Mode^TMOperation at Light Load

■

TheLTC1872boastsa±2.5%outputvoltageaccuracyand

consumes only 270µA of quiescent current. For applica-

tionswhereefficiencyisaprimeconsideration,theLTC1872

is configured for Burst Mode operation, which enhances

efficiency at low output current.

■

Current Mode Operation for Excellent Line and Load

Transient Response

Low Quiescent Current: 270µA

Shutdown Mode Draws Only 8µA Supply Current

±2.5% Reference Accuracy

Tiny 6-Lead SOT-23 Package

■

In shutdown, the device draws a mere 8µA. The high

550kHz constant operating frequency allows the use of a

small external inductor.

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APPLICATIO S

The LTC1872 is available in a small footprint 6-lead

SOT-23.

■

Lithium-Ion-Powered Applications

■

Cellular Telephones

, LTC and LT are registered trademarks of Linear Technology Corporation.

Burst Mode is a trademark of Linear Technology Corporation.

■

Wireless Modems

■

Portable Computers

■

Scanners

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TYPICAL APPLICATION

V

IN

Efficiency vs Load Current

3.3V

C1

10µF

10V

R1

100

95

90

85

80

75

70

65

0.03Ω

V

= 3.3V

= 5V

IN

OUT

147k

1

5

L1

4.7µH

I

/RUN

V

TH

IN

–

LTC1872

220pF

80.6k

V

5V

1A

OUT

2

3

4

6

GND

SENSE

NGATE

+

C2

2× 22µF

6.3V

D1

M1

V

FB

422k

C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT

C2: MURATA GRM42-2X5R226K6.3

D1: IR10BQ015

L1: MURATA LQN6C4R7M04

M1: IRLMS2002

R1: DALE 0.25W

1872 TA01

1

10

100

1000

LOAD CURRENT (mA)

Figure 1. LTC1872 High Output Current 3.3V to 5V Boost Converter

1872 TA01b

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LTC1872

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ABSOLUTE MAXIMUM RATINGS

PACKAGE/ORDER INFORMATION

(Note 1)

Input Supply Voltage (V_IN).........................–0.3V to 10V

SENSE^–, NGATE Voltages ............ –0.3V to (V_IN+ 0.3V)

V_FB, I_TH/RUN Voltages ..............................–0.3V to 2.4V

NGATE Peak Output Current (<10µs) ....................... 1A

Storage Ambient Temperature Range ... –65°C to 150°C

Operating Temperature Range (Note 2) .. –40°C to 85°C

Junction Temperature (Note 3)............................. 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

ORDER PART

NUMBER

TOP VIEW

LTC1872ES6

I

/RUN 1

GND 2

6 NGATE

5 V

TH

IN

4 SENSE

–

V

3

FB

S6 PART MARKING

LTMK

S6 PACKAGE

6-LEAD PLASTIC SOT-23

T_JMAX= 150°C, θ_JA= 230°C/ W

Consult factory for parts specified with wider operating temperature ranges.

The ● denotes specifications that apply over the full operating temperature

ELECTRICAL CHARACTERISTICS

range, otherwise specifications are at T_A= 25°C. V_IN= 4.2V unless otherwise specified. (Note 2)

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

Input DC Supply Current

Normal Operation

Sleep Mode

Shutdown

UVLO

Typicals at V = 4.2V (Note 4)

IN

2.4V ≤ V ≤ 9.8V

270

230

8

420

370

22

µA

IN

2.4V ≤ V ≤ 9.8V

IN

2.4V ≤ V ≤ 9.8V, V /RUN = 0V

IN ITH

V

< UVLO Threshold

6

10

IN

Undervoltage Lockout Threshold

V

Falling

Rising

●

1.55

1.85

2.00

2.10

2.35

2.40

V

IN

Shutdown Threshold (at I /RUN)

0.15

0.25

0.35

0.5

0.55

0.85

V

TH

Start-Up Current Source

V

/RUN = 0V

ITH

µA

Regulated Feedback Voltage

0°C to 70°C(Note 5)

–40°C to 85°C(Note 5)

●

0.780

0.770

0.800

0.820

0.830

V

Input Current

(Note 5)

10

550

40

50

nA

kHz

ns

FB

Oscillator Frequency

Gate Drive Rise Time

Gate Drive Fall Time

V

C

= 0.8V

500

114

650

FB

= 3000pF

LOAD

40

ns

Peak Current Sense Voltage

(Note 6)

120

mV

Note 1: Absolute Maximum Ratings are those values beyond which the life

Note 4: Dynamic supply current is higher due to the gate charge being

of a device may be impaired.

delivered at the switching frequency.

Note 2: The LTC1872E is guaranteed to meet performance specifications

from 0°C to 70°C. Specifications over the –40°C to 85°C operating

temperature range are assured by design, characterization and correlation

with statistical process controls.

Note 5: The LTC1872 is tested in a feedback loop that servos V to the

output of the error amplifier.

Note 6: Guaranteed by design at duty cycle = 30%. Peak current sense

FB

voltage is V /6.67 at duty cycle <40%, and decreases as duty cycle

REF

Note 3: T is calculated from the ambient temperature T and power

increases due to slope compensation as shown in Figure 2.

J

A

dissipation P according to the following formula:

D

T = T + (P • θ °C/W)

J

A

D

JA

2

LTC1872

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TYPICAL PERFORMANCE CHARACTERISTICS

Undervoltage Lockout Trip

Voltage vs Temperature

Reference Voltage

vs Temperature

Normalized Oscillator Frequency

vs Temperature

825

820

815

810

805

800

795

790

785

780

775

10

8

2.24

2.20

2.16

2.12

2.08

2.04

2.00

1.96

1.92

1.88

1.84

V

IN

= 4.2V

V

IN

= 4.2V

V

IN

FALLING

6

4

2

0

–2

–4

–6

–8

–10

–55 –35 –15

5

25 45 65 85 105 125

–55 –35 –15

5

25 45 65 85 105 125

–55 –35 –15

5

25 45 65 85 105 125

TEMPERATURE (°C)

1872 G01

1872 G02

1872 G03

Maximum Current Sense Trip

Voltage vs Duty Cycle

Shutdown Threshold

vs Temperature

600

560

520

480

440

400

360

320

280

240

200

130

120

110

100

90

V

IN

= 4.2V

V

A

= 4.2V

IN

T

= 25°C

80

70

60

50

60 70

–55 –35 –15

5

45

85 105 125

20 30 40 50

80 90 100

25

65

TEMPERATURE (°C)

DUTY CYCLE (%)

1872 G04

1872 G05

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PIN FUNCTIONS

I_TH/RUN (Pin 1): This pin performs two functions. It

servesastheerroramplifiercompensationpointaswellas

the run control input. Nominal voltage range for this pin is

0.7V to 1.9V. Forcing this pin below 0.35V causes the

device to be shut down. In shutdown all functions are

disabled and the NGATE pin is held low.

SENSE^–(Pin 4): The Negative Input to the Current Com-

parator.

V_IN(Pin5):SupplyPin. MustbecloselydecoupledtoGND

Pin 2.

NGATE (Pin 6): Gate Drive for the External N-Channel

MOSFET. This pin swings from 0V to V_IN.

GND (Pin 2): Ground Pin.

V

_FB(Pin 3): Receives the feedback voltage from an exter-

nal resistive divider across the output.

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LTC1872

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FUNCTIONAL DIAGRA

–

V

SENSE

4

IN

5

+

–

ICMP

V

IN

RS

NGATE

6

SWITCHING

LOGIC AND

BLANKING

CIRCUIT

SLOPE

COMP

R

OSC

Q

S

–

+

FREQ

FOLDBACK

BURST

CMP

OVP

+

–

+

–

0.3V

V

+

SLEEP

REF

0.15V

60mV

V

IN

EAMP

V

REF

+

–

0.8V

0.5µA

V

FB

I

/RUN

1

3

+

TH

V

IN

V

IN

0.3V

–

0.35V

+

SHDN

UV

SHDN

CMP

VOLTAGE

REFERENCE

V

REF

0.8V

–

GND

2

UNDERVOLTAGE

LOCKOUT

1.2V

1872FD

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(Refer to Functional Diagram)

OPERATIO

Main Control Loop

0.8V reference, which in turn causes the

I_TH/RUN voltage to increase until the average inductor

current matches the new load current.

TheLTC1872isaconstantfrequencycurrentmodeswitch-

ing regulator. During normal operation, the external

N-channel power MOSFET is turned on each cycle by the

oscillator and turned off when the current comparator

(ICMP) resets the RS latch. The peak inductor current at

whichICMPresetstheRSlatchiscontrolledbythevoltage

on the I_TH/RUN pin, which is the output of the error

amplifier EAMP. An external resistive divider connected

between V_OUTand ground allows the EAMP to receive an

output feedback voltage V_FB. When the load current in-

creases, it causes a slight decrease in V_FBrelative to the

ThemaincontrolloopisshutdownbypullingtheI_TH/RUN

pin low. Releasing I_TH/RUN allows an internal 0.5µA

current source to charge up the external compensation

network. When the I_TH/RUN pin reaches 0.35V, the main

control loop is enabled with the I_TH/RUN voltage then

pulled up to its zero current level of approximately 0.7V.

Astheexternalcompensationnetworkcontinuestocharge

up, the corresponding output current trip level follows,

allowing normal operation.

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LTC1872

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(Refer to Functional Diagram)

OPERATIO

Comparator OVP guards against transient overshoots

>7.5% by turning off the external N-channel power

MOSFET and keeping it off until the fault is removed.

Overvoltage Protection

The overvoltage comparator in the LTC1872 will turn the

external MOSFET off when the feedback voltage has risen

7.5% above the reference voltage of 0.8V. This compara-

tor has a typical hysteresis of 20mV.

Burst Mode Operation

The LTC1872 enters Burst Mode operation at low load

currents. In this mode, the peak current of the inductor is

set as if V_ITH/RUN = 1V (at low duty cycles) even though

the voltage at the I_TH/RUN pin is at a lower value. If the

inductor’saveragecurrentisgreaterthantheloadrequire-

ment, the voltage at the I_TH/RUN pin will drop. When the

I_TH/RUN voltage goes below 0.85V, the sleep signal goes

high, turning off the external MOSFET. The sleep signal

goes low when the I_TH/RUN voltage goes above 0.925V

and the LTC1872 resumes normal operation. The next

oscillator cycle will turn the external MOSFET on and the

switching cycle repeats.

Slope Compensation and Inductor’s Peak Current

The inductor’s peak current is determined by:

V

_ITH− 0.7

I_PK

=

10 R_SENSE

(

)

when the LTC1872 is operating below 40% duty cycle.

However, once the duty cycle exceeds 40%, slope com-

pensation begins and effectively reduces the peak induc-

torcurrent. Theamountofreductionisgivenbythecurves

in Figure 2.

Undervoltage Lockout

Short-Circuit Protection

TopreventoperationoftheN-channelMOSFETbelowsafe

input voltage levels, an undervoltage lockout is incorpo-

rated into the LTC1872. When the input supply voltage

drops below approximately 2.0V, the N-channel MOSFET

and all circuitry is turned off except the undervoltage

block, which draws only several microamperes.

Sincethepowerswitchinaboostconverterisnotinseries

with the power path from input to load, turning off the

switch provides no protection from a short-circuit at the

output. External means such as a fuse in series with the

boost inductor must be employed to handle this fault

condition.

110

100

90

80

70

60

50

I

= 0.4I

PK

RIPPLE

AT 5% DUTY CYCLE

= 0.2I

40

30

20

10

I

RIPPLE

PK

AT 5% DUTY CYCLE

V

IN

= 4.2V

0

10 20 30 40 50 60 70 80 90 100

DUTY CYCLE (%)

1872 F02

Figure 2. Maximum Output Current vs Duty Cycle

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APPLICATIONS INFORMATION

The basic LTC1872 application circuit is shown in

Figure 1. External component selection is driven by the

load requirement and begins with the selection of L1 and

Inductor Value Calculation

The operating frequency and inductor selection are inter-

related in that higher operating frequencies permit the use

of a smaller inductor for the same amount of inductor

ripplecurrent. However, thisisattheexpenseofefficiency

due to an increase in MOSFET gate charge losses.

R

_SENSE(= R1). Next, the power MOSFET and the output

diodeD1isselectedfollowedbyC_IN(=C1)andC_OUT(=C2).

R_SENSESelection for Output Current

The inductance value also has a direct effect on ripple

current. The ripple current, I_RIPPLE, decreases with higher

inductance or frequency and increases with higher V_OUT

The inductor’s peak-to-peak ripple current is given by:

R_SENSEis chosen based on the required output current.

Withthecurrentcomparatormonitoringthevoltagedevel-

oped across R_SENSE, the threshold of the comparator

determines the inductor’s peak current. The output cur-

rent the LTC1872 can provide is given by:

.

V

V_OUT+ V_D− V

IN

I_RIPPLE

=

V_OUT+ V_D

f L

( )

0.12

R_SENSE

I_RIPPLE

V

IN

V_OUT+ V_D

I_OUT

=

−

2

wherefistheoperatingfrequency.Acceptinglargervalues

of I_RIPPLEallows the use of low inductances, but results in

higher output voltage ripple and greater core losses. A

reasonable starting point for setting ripple current is:

where I_RIPPLEis the inductor peak-to-peak ripple current

(see Inductor Value Calculation section) and V_Dis the

forward drop of the output diode at the full rated output

current.

V_OUT+ V_D

I

_RIPPLE= 0.4 I_{OUT MAX}

(

)

A reasonable starting point for setting ripple current is:

V

IN

V_OUT+ V_D

In Burst Mode operation, the ripple current is normally set

such that the inductor current is continuous during the

burst periods. Therefore, the peak-to-peak ripple current

must not exceed:

I

_RIPPLE= O.4 I_OUT

(

)(

)

V

IN

Rearranging the above equation, it becomes:

1

V

0.03

R_SENSE

IN

R_SENSE

=

I_RIPPLE

≤

V_OUT+ V_D

I_OUT

10

( )(

)

for Duty Cycle < 40%

This implies a minimum inductance of:

However,foroperationthatisabove40%dutycycle,slope

compensation’s effect has to be taken into consideration

to select the appropriate value to provide the required

amount of current. Using the scaling factor (SF, in %) in

Figure 2, the value of R_SENSEis:

V

V_OUT+ V_D− V

IN

L_MIN

=

V_OUT+ V_D

0.03

R_SENSE

f

A smaller value than L_MINcould be used in the circuit;

however, the inductor current will not be continuous

during burst periods.

V

SF

IN

R_SENSE

=

V_OUT+ V_D

10 I_OUT100

( )(

)(

)

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APPLICATIONS INFORMATION

Itisimportanttoadequatelyspecifythediodepeakcurrent

and average power dissipation so as not to exceed the

diode ratings.

Inductor Selection

When selecting the inductor, keep in mind that inductor

saturation current has to be greater than the current limit

set by the current sense resistor. Also, keep in mind that

the DC resistance of the inductor will affect the efficiency.

OfftheshelfinductorsareavailablefromMurata,Coilcraft,

Toko, Panasonic, Coiltronics and many other suppliers.

Schottky diodes are recommended for low forward drop

and fast switching times. Remember to keep lead length

short and observe proper grounding (see Board Layout

Checklist) to avoid ringing and increased dissipation.

Power MOSFET Selection

C_INand C_OUTSelection

The main selection criteria for the power MOSFET are the

To prevent large input voltage ripple, a low ESR input

capacitor sized for the maximum RMS current must be

used. The maximum RMS capacitor current for a boost

converter is approximately equal to:

threshold voltage V_GS(TH), the “on” resistance R_DS(ON)

,

reverse transfer capacitance C_RSSand total gate charge.

Since the LTC1872 is designed for operation down to low

input voltages, a logic level threshold MOSFET (R_DS(ON)

guaranteed at V_GS= 2.5V) is required for applications that

workclosetothisvoltage.WhentheseMOSFETsareused,

makesurethattheinputsupplytotheLTC1872islessthan

the absolute maximum V_GSrating, typically 8V.

C_INRequired I_RMS≈ 0.3 I_RIPPLE

(

)

where I_RIPPLEis as defined in the Inductor Value Calcula-

tion section.

Note that capacitor manufacturer’s ripple current ratings

are often based on 2000 hours of life. This makes it

advisable to further derate the capacitor, or to choose a

capacitor rated at a higher temperature than required.

Several capacitors may be paralleled to meet the size or

height requirements in the design. Due to the high operat-

ingfrequencyoftheLTC1872,ceramiccapacitorscanalso

be used for C_IN. Always consult the manufacturer if there

is any question.

The required minimum R_DS(ON)of the MOSFET is gov-

erned by its allowable power dissipation given by:

P_P

R_DS(ON)

2

1+ δp

DC I

(

)

(

)

IN

where P_Pis the allowable power dissipation and δp is the

temperature dependency of R_DS(ON). (1 + δp) is generally

given for a MOSFET in the form of a normalized R_DS(ON)vs

temperature curve, but δp = 0.005/°C can be used as an

approximation for low voltage MOSFETs. DC is the maxi-

mum operating duty cycle of the LTC1872.

The selection of C_OUTis driven by the required effective

series resistance (ESR). Typically, once the ESR require-

ment is satisfied, the capacitance is adequate for filtering.

The output ripple (∆V_OUT) is approximated by:

Output Diode Selection

V_OUT+ V_DI_RIPPLE

Under normal load conditions, the average current con-

ducted by the diode in a boost converter is equal to the

output load current:

∆V_OUT≈ I_O•

+

•

V

IN

2

1

2

1

I_D(avg)= I_OUT

ESR²+

2πfC_OUT

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APPLICATIONS INFORMATION

where f is the operating frequency, C_OUTis the output

capacitance and I_RIPPLEis the ripple current in the induc-

tor.

Setting Output Voltage

The LTC1872 develops a 0.8V reference voltage between

thefeedback(Pin3)terminalandground(seeFigure4).By

selecting resistor R1, a constant current is caused to flow

through R1 and R2 to set the overall output voltage. The

regulated output voltage is determined by:

Manufacturers such as Nichicon, United Chemicon and

Sanyoshouldbeconsideredforhighperformancethrough-

hole capacitors. The OS-CON semiconductor dielectric

capacitor available from Sanyo has the lowest ESR (size)

product of any aluminum electrolytic at a somewhat

higherprice.TheoutputcapacitorRMScurrentisapproxi-

mately equal to:

R2

R1

V_OUT= 0.8V 1+

105

V

I_PK• DC −DC²

REF

100

95

90

85

80

75

where I_PKis the peak inductor current and DC is the switch

duty cycle.

V

ITH

Whenusingelectrolyticoutputcapacitors, iftherippleand

ESR requirements are met, there is likely to be far more

capacitance than required.

In surface mount applications, multiple capacitors may

have to be paralleled to meet the ESR or RMS current

handling requirements of the application. Aluminum elec-

trolytic and dry tantalum capacitors are both available in

surface mount configurations. An excellent choice of

tantalum capacitors is the AVX TPS and KEMET T510

series of surface mount tantalum capacitors. Also,

ceramic capacitors in X5R pr X7R dielectrics offer excel-

lent performance.

2.0

2.2

2.4

2.6

2.8

3.0

INPUT VOLTAGE (V)

1872 F03

Figure 3. Line Regulation of V_REFand V_ITH

V

OUT

R2

R1

LTC1872

3

V

FB

Low Supply Operation

1872 F04

Although the LTC1872 can function down to approxi-

mately 2.0V, the maximum allowable output current is

reducedwhenV_INdecreasesbelow3V. Figure3showsthe

amount of change as the supply is reduced down to 2V.

Also shown in Figure 3 is the effect of V_INon V_REFas V_IN

goes below 2.3V.

Figure 4. Setting Output Voltage

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APPLICATIONS INFORMATION

Formostapplications, an80kresistorissuggestedforR1.

To prevent stray pickup, locate resistors R1 and R2 close

to LTC1872.

2. MOSFET gate charge current results from switching

the gate capacitance of the power MOSFET. Each time

a MOSFET gate is switched from low to high to low

again, a packet of charge, dQ, moves from V_INto

ground. The resulting dQ/dt is a current out of V_IN

which is typically much larger than the contoller’s DC

supply current. In continuous mode, I_GATECHG= f(Qp).

3. I²R losses are predicted from the DC resistances of the

MOSFET, inductor and current sense resistor. The

MOSFET R_DS(ON)multiplied by duty cycle times the

average output current squared can be summed with

I²RlossesintheinductorESRinserieswiththecurrent

sense resistor.

Efficiency Considerations

The efficiency of a switching regulator is equal to the

output power divided by the input power times 100%. It is

oftenusefultoanalyzeindividuallossestodeterminewhat

is limiting the efficiency and which change would produce

the most improvement. Efficiency can be expressed as:

Efficiency = 100% – (η1 + η2 + η3 + ...)

where η1, η2, etc. are the individual losses as a percent-

age of input power.

4. The output diode is a major source of power loss at

high currents. The diode loss is calculated by multiply-

ing the forward voltage by the load current.

Although all dissipative elements in the circuit produce

losses, four main sources usually account for most of the

losses in LTC1872 circuits: 1) LTC1872 DC bias current,

2) MOSFET gate charge current, 3) I²R losses and 4)

voltage drop of the output diode.

5. Transition losses apply to the external MOSFET and

increase at higher operating frequencies and input

voltages. Transition losses can be estimated from:

1. The V_INcurrent is the DC supply current, given in the

electricalcharacteristics, thatexcludesMOSFETdriver

and control currents. V_INcurrent results in a small loss

which increases with V_IN.

Transition Loss = 2(V_IN)²I_{IN(MAX) RSS}

(f)

C

Other losses, including C_INand C_OUTESR dissipative

losses, and inductor core losses, generally account for

less than 2% total additional loss.

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APPLICATIONS INFORMATION

PC Board Layout Checklist

4. Connect the end of R_SENSEas close to V_IN(Pin 5) as

possible. The V_INpin is the SENSE⁺of the current

comparator.

5. The trace from SENSE^–(Pin 4) to the Sense resistor

should be kept short. The trace should connect close

When laying out the printed circuit board, the following

checklist should be used to ensure proper operation of the

LTC1872. These items are illustrated graphically in the

layout diagram in Figure 5. Check the following in your

layout:

to R_SENSE

.

1. The Schottky diode should be closely connected

between the output capacitor and the drain of the

external MOSFET.

6. Keep the switching node NGATE away from sensitive

small signal nodes.

7. The V_FBpin should connect directly to the feedback

resistors. The resistive divider R1 and R2 must be

connected between the (+) plate of C_OUTand signal

ground.

2. The (+) plate of C_INshould connect to the sense

resistor as closely as possible. This capacitor provides

AC current to the inductor.

3. The input decoupling capacitor (0.1µF) should be

connected closely between V_IN(Pin 5) and ground

(Pin 2).

V

IN

1

2

3

6

5

4

I

/RUN NGATE

LTC1872

TH

M1

L1

R

S

R

ITH

GND

V

IN

+

0.1µF

C

IN

D1

–

V

SENSE

C

ITH

FB

V

OUT

+

R2

C

OUT

R1

BOLD LINES INDICATE HIGH CURRENT PATHS

1872 F05

Figure 5. LTC1872 Layout Diagram (See PC Board Layout Checklist)

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LTC1872

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TYPICAL APPLICATIO

LTC1872 12V/500mA Boost Converter

V

IN

3V TO 9.8V

C1

10µF

10V

R1

0.033Ω

1

5

L1

I

/RUN

V

TH

IN

–

10µH

LTC1872

10k

V

2

3

4

6

OUT

GND

SENSE

NGATE

12V

+

C2

47µF

16V

220pF

D1

M1

V

FB

1.1M

1872 TA02

78.7k

C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT

C2: AVX TPSE476M016R0150

D1: IR10BQ015

L1: COILTRONICS UP2B-100

M1: Si9804DV

R1: DALE 0.25W

LTC1872 Three-Cell White LED Driver

V

= 3 AA CELLS ≈ 2.7V TO 4.8V

IN

C1

R1

0.27Ω

10µF

10V

AA

1

5

L1

I

/RUN

V

TH

IN

150µH

V

≈ 28.8V

OUT

LTC1872

10k

220pF

(WITH 8 LEDs)

2

3

4

6

–

GND

SENSE

NGATE

+

C2

15µF

35V

C3

15mA

D0

M1

V

0.1µF

CERAMIC

FB

D1

D2

1 TO 8

WHITE

LEDs

•

D8

C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT

C2: AVX TPSD156M035R0300

D0: MOTOROLA MBR0540

L1: COILCRAFT DO1608C-154

M1: Si9804

R1: DALE 0.25W

53.6Ω

1872 TA04

D1-D7: CMD333UWC

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.

11

LTC1872

U

TYPICAL APPLICATIO

LTC1872 –2.5V to 3.3V/0.5A Boost Converter

LTC1872 2.7V to 9.8V Input

to 3.3V/1.2A Output SEPIC Converter

R1

0.034Ω

C2

V

IN

2× 100µF

2.7V TO 9.8V

C

+

IN

R

CS

0.03Ω

1

5

10V

10µF

L1

4.7µH

I

/RUN

V

IN

C

C1

TH

10V, X5R

R

C1

10k

220pF

LTC1872

10k

V

3.3V

0.5A

V

1

5

OUT

2

3

4

6

I

/RUN

V

–

TH

IN

–

3.3V/1.2A

GND

SENSE

NGATE

+

C01

180µF

4V, SP

D1

MBRM120

L1A

L1B

220pF

C1

LTC1872

D1

332k

V

M1

FB

2

3

4

6

GND

SENSE

NGATE

CS

4.7µF

10V

V

FB

R

f2

80.6k

+

U1

0.1µF

CERAMIC

100µF

10V

R

f1

252k

M1

1872 TA05

80.6k

180k

C

01

, CS; TOKO, MURATA OR TAIYO YUDEN

: PANASONIC EEFUE0G181R

FOR V

= 5V CHANGE

IN

OUT

V

IN

–2.5V

C

R

TO 427kΩ AND

f1

1872 TA03

L1: BH ELECTRONICS 511-1012

M1: IRLMS2002

C

TO 150µF, 6V PANASONIC

01

C1, C2: AVX TPSE107M010R0100

D1: MOTOROLA MBR2045CT

L1: COILTRONICS UP2B-4R7

M1: Si9804DV

SP TYPE CAPACITOR

R1: DALE 0.25W

R

: DALE OR IRC

CS

U1: PANASONIC 2SB709A

U

PACKAGE DESCRIPTION ^{Dimensions in inches (millimeters) unless otherwise noted.}

S6 Package

6-Lead Plastic SOT-23

(LTC DWG # 05-08-1634)

2.6 – 3.0

(0.110 – 0.118)

2.80 – 3.00

(0.110 – 0.118)

(NOTE 3)

1.50 – 1.75

(0.059 – 0.069)

0.00 – 0.15

(0.00 – 0.006)

0.90 – 1.45

(0.035 – 0.057)

0.35 – 0.55

(0.014 – 0.022)

0.35 – 0.50

(0.014 – 0.020)

SIX PLACES (NOTE 2)

0.90 – 1.30

(0.035 – 0.051)

0.95

(0.037)

REF

0.09 – 0.20

(0.004 – 0.008)

(NOTE 2)

1.90

(0.074)

REF

NOTE:

S6 SOT-23 0898

1. DIMENSIONS ARE IN MILLIMETERS

2. DIMENSIONS ARE INCLUSIVE OF PLATING

3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR

4. MOLD FLASH SHALL NOT EXCEED 0.254mm

5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ)

RELATED PARTS

PART NUMBER

LT1304

DESCRIPTION

COMMENTS

Micropower DC/DC Converter with Low-Battery Detector

1.7MHz, Single Cell Micropower DC/DC Converter

1.4MHz, Single Cell DC/DC Converter in 5-Lead SOT-23

Low Voltage Current Mode PWM Controller

High Power DC/DC Step-Up Controller

120µA Quiescent Current, 1.5V ≤ V ≤ 8V

IN

LT1610

30µA Quiescent Current, V Down to 1V

IN

LT1613

Internally Compensated, V Down to 1V

IN

LT1619

8-Lead MSOP Package, 1.9V ≤ V ≤ 18V

IN

LT1680

Operation Up to 60V, Fixed Frequency Current Mode

8-Pin N-Channel Drive, 3.5V ≤ V ≤ 36V

LTC1624

LT1615

High Efficiency SO-8 N-Channel Switching Regulator Controller

Micropower Step-Up DC/DC Converter in SOT-23

IN

20µA Quiescent Current, V Down to 1V

IN

LTC1700

LTC1772

LTC3401/LTC3402

No R

Synchronous Current Mode DC/DC Step-Up Controller

95% Efficient, 0.9V ≤ V ≤ 5V, 550kHz Operation

IN

SENSE

Constant Frequency Current Mode Step-Down DC/DC Controller

1A/2A, 3MHz Micropower Synchronous Boost Converter

V

2.5V to 9.8V, I

up to 4A, SOT-23 Package

OUT

IN

10-Lead MSOP Package, 0.5V ≤ V ≤ 5V

IN

1872f LT/TP 0301 4K • PRINTED IN USA

12 ^{LinearTechnology Corporation}

1630 McCarthy Blvd., Milpitas, CA 95035-7417

●

LINEAR TECHNOLOGY CORPORATION 2000

(408)432-1900 FAX:(408)434-0507 www.linear-tech.com


型号：	LTC1872
厂家：	Linear
描述：	Constant Frequency Current Mode Step-Up DC/DC Controller in SOT-23 恒定频率电流模式升压型DC / DC控制器，采用SOT -23 控制器
文件：	总12页 (文件大小：183K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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