LTC1872BES6#TRMPBF [Linear]
LTC1872B - Constant Frequency Current Mode Step-Up DC/DC Controller in ThinSOT; Package: SOT; Pins: 6; Temperature Range: -40°C to 85°C;
| 型号： | LTC1872BES6#TRMPBF |
| 厂家： | 
Linear |
| 描述： | LTC1872B - Constant Frequency Current Mode Step-Up DC/DC Controller in ThinSOT; Package: SOT; Pins: 6; Temperature Range: -40°C to 85°C 开关 光电二极管 控制器 |
| 文件： | 总12页 (文件大小：195K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC1872B
Constant Frequency
Current Mode Step-Up
DC/DC Controller in ThinSOT
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FEATURES
DESCRIPTIO
Burst ModeTM Operation Disabled for Lower Output
The LTC®1872B is a constant frequency current mode
step-up DC/DC controller providing excellent AC and DC
load and line regulation. The device incorporates an accu-
rate undervoltage lockout feature that shuts down the
LTC1872B when the input voltage falls below 2.0V.
■
Ripple at Light Loads
■
High Efficiency: Over 90%
■
High Output Currents Easily Achieved
■
Wide VIN Range: 2.5V to 9.8V
■
VOUT Limited Only by External Components
The LTC1872B provides a ±2.5% output voltage accuracy
and consumes only 270µA of quiescent current. In shut-
down, the device draws a mere 8µA.
■
Constant Frequency 550kHz Operation
■
Current Mode Operation for Excellent Line and Load
Transient Response
■
■
High constant operating frequency of 550kHz allows the
use of a small external inductor. The constant frequency
operation is maintained down to very light loads, resulting
in less low frequency noise generation over a wide load
current range.
Shutdown Mode Draws Only 8µA Supply Current
Low Profile (1mm) ThinSOTTM Package
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APPLICATIO S
■
Optical Communications
The LTC1872B is available in a 6-lead low profile (1mm)
ThinSOT package. For a Burst Mode operation enabled
versionoftheLTC1872B, pleaserefertotheLTC1872data
sheet.
■
Lithium-Ion-Powered Applications
■
Cellular Telephones
■
Wireless Devices
■
Portable Computers
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode and ThinSOT are trademarks of Linear Technology Corporation.
■
Scanners
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TYPICAL APPLICATION
Typical Efficiency vs Load Current*
V
IN
3.3V
100
95
90
85
80
75
70
65
C1
10µF
10V
R1
0.03Ω
V
V
= 3.3V
OUT
IN
= 5V
147k
1
5
L1
I
/RUN
V
TH
IN
–
4.7µH
LTC1872B
GND
220pF
80.6k
V
OUT
2
3
4
6
5V
1A
SENSE
C2
4× 10µF
10V
D1
M1
V
NGATE
FB
422k
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: MURATA GRM42-2X5R106K010AL
D1: IR10BQ015
L1: MURATA LQN6C4R7M04
M1: Si2302DS
R1: DALE 0.25W
1
10
100
1000
LOAD CURRENT (mA)
1872B TA01
1872B TA01b
Figure 1. LTC1872B High Output Current 3.3V to 5V Boost Converter
*Output ripple waveforms for the circuit of Figure 1 appear in Figure 2.
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LTC1872B
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
Input Supply Voltage (VIN).........................–0.3V to 10V
SENSE–, NGATE Voltages ............ –0.3V to (VIN + 0.3V)
VFB, ITH/RUN Voltages ..............................–0.3V to 2.4V
NGATE Peak Output Current (<10µs) ....................... 1A
Storage Ambient Temperature Range ... –65°C to 150°C
Operating Temperature Range (Note 2) .. –40°C to 85°C
Junction Temperature (Note 3)............................. 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
LTC1872BES6
I
/RUN 1
GND 2
6 NGATE
5 V
TH
IN
4 SENSE
–
V
3
FB
S6 PART MARKING
LTXY
S6 PACKAGE
6-LEAD PLASTIC SOT-23
TJMAX = 150°C, θJA = 230°C/ W
Consult LTC marketing for parts specified with wider operating temperature ranges.
The ● denotes specifications that apply over the full operating temperature
ELECTRICAL CHARACTERISTICS
range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input DC Supply Current
Normal Operation
Sleep Mode
Shutdown
UVLO
Typicals at V = 4.2V (Note 4)
IN
2.4V ≤ V ≤ 9.8V
270
230
8
420
370
22
µA
µA
µA
µA
IN
2.4V ≤ V ≤ 9.8V
IN
2.4V ≤ V ≤ 9.8V, V /RUN = 0V
IN ITH
V
< UVLO Threshold
6
10
IN
Undervoltage Lockout Threshold
V
V
Falling
Rising
●
●
1.55
1.85
2.00
2.10
2.35
2.40
V
V
IN
IN
Shutdown Threshold (at I /RUN)
0.15
0.25
0.35
0.5
0.55
0.85
V
TH
Start-Up Current Source
V
/RUN = 0V
ITH
µA
Regulated Feedback Voltage
0°C to 70°C(Note 5)
–40°C to 85°C(Note 5)
●
●
0.780
0.770
0.800
0.800
0.820
0.830
V
V
V
Input Current
(Note 5)
10
550
40
50
nA
kHz
ns
FB
Oscillator Frequency
Gate Drive Rise Time
Gate Drive Fall Time
V
C
C
= 0.8V
500
114
650
FB
= 3000pF
= 3000pF
LOAD
LOAD
40
ns
Peak Current Sense Voltage
(Note 6)
120
mV
Note 1: Absolute Maximum Ratings are those values beyond which the life
Note 4: Dynamic supply current is higher due to the gate charge being
of a device may be impaired.
delivered at the switching frequency.
Note 2: The LTC1872BE is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 5: The LTC1872B is tested in a feedback loop that servos V to the
output of the error amplifier.
Note 6: Guaranteed by design at duty cycle = 30%. Peak current sense
FB
voltage is V /6.67 at duty cycle <40%, and decreases as duty cycle
REF
Note 3: T is calculated from the ambient temperature T and power
increases due to slope compensation as shown in Figure 3.
J
A
dissipation P according to the following formula:
D
T = T + (P • θ °C/W)
J
A
D
JA
2
LTC1872B
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TYPICAL PERFORMANCE CHARACTERISTICS
Undervoltage Lockout Trip
Voltage vs Temperature
Reference Voltage
vs Temperature
Normalized Oscillator Frequency
vs Temperature
825
820
815
810
805
800
795
790
785
780
775
10
8
2.24
2.20
2.16
2.12
2.08
2.04
2.00
1.96
1.92
1.88
1.84
V
IN
= 4.2V
V
IN
= 4.2V
V
IN
FALLING
6
4
2
0
–2
–4
–6
–8
–10
–55 –35 –15
5
25 45 65 85 105 125
–55 –35 –15
5
25 45 65 85 105 125
–55 –35 –15
5
25 45 65 85 105 125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
1872B G01
1872B G02
1872B G03
Maximum Current Sense Trip
Voltage vs Duty Cycle
Shutdown Threshold
vs Temperature
600
560
520
480
440
400
360
320
280
240
200
130
120
110
100
90
V
IN
= 4.2V
V
A
= 4.2V
IN
T
= 25°C
80
70
60
50
60 70
–55 –35 –15
5
45
85 105 125
20 30 40 50
80 90 100
25
65
TEMPERATURE (°C)
DUTY CYCLE (%)
187B2 G04
1872B G05
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PIN FUNCTIONS
ITH/RUN (Pin 1): This pin performs two functions. It
servesastheerroramplifiercompensationpointaswellas
the run control input. Nominal voltage range for this pin is
0.7V to 1.9V. Forcing this pin below 0.35V causes the
device to be shut down. In shutdown all functions are
disabled and the NGATE pin is held low.
SENSE– (Pin 4): The Negative Input to the Current Com-
parator.
VIN (Pin5):SupplyPin. MustbecloselydecoupledtoGND
Pin 2.
NGATE (Pin 6): Gate Drive for the External N-Channel
MOSFET. This pin swings from 0V to VIN.
GND (Pin 2): Ground Pin.
V
FB (Pin 3): Receives the feedback voltage from an exter-
nal resistive divider across the output.
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LTC1872B
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FUNCTIONAL DIAGRA
–
V
SENSE
4
IN
5
15mV
+
–
ICMP
V
IN
RS
NGATE
6
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
SLOPE
COMP
R
OSC
Q
S
–
+
FREQ
FOLDBACK
OVP
+
–
0.3V
V
+
REF
60mV
EAMP
V
REF
0.8V
+
–
0.5µA
V
FB
I
TH
/RUN
3
+
1
V
IN
V
IN
–
0.35V
+
SHDN
UV
SHDN
CMP
VOLTAGE
REFERENCE
V
REF
0.8V
–
GND
2
UNDERVOLTAGE
LOCKOUT
1.2V
1872B FD
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(Refer to Functional Diagram)
OPERATIO
Main Control Loop
pulled up to its zero current level of approximately 0.7V.
Astheexternalcompensationnetworkcontinuestocharge
up, the corresponding output current trip level follows,
allowing normal operation.
The LTC1872B is a constant frequency current mode
switchingregulator.Duringnormaloperation,theexternal
N-channel power MOSFET is turned on each cycle by the
oscillator and turned off when the current comparator
(ICMP) resets the RS latch. The peak inductor current at
whichICMPresetstheRSlatchiscontrolledbythevoltage
on the ITH/RUN pin, which is the output of the error
amplifier EAMP. An external resistive divider connected
between VOUT and ground allows the EAMP to receive an
output feedback voltage VFB. When the load current in-
creases, it causes a slight decrease in VFB relative to the
0.8V reference, which in turn causes the
ITH/RUN voltage to increase until the average inductor
current matches the new load current.
Comparator OVP guards against transient overshoots
>7.5% by turning off the external N-channel power
MOSFET and keeping it off until the fault is removed.
Low Load Current Operation
Under very light load current conditions, the ITH/RUN pin
voltage will be very close to the zero current level of 0.85V.
As the load current decreases further, an internal offset at
the current comparator input will assure that the current
comparator remains tripped (even at zero load current)
and the regulator will start to skip cycles, as it must, in
order to maintain regulation. This behavior allows the
regulator to maintain constant frequency down to very
light loads, resulting in less low frequency noise genera-
tion over a wide load current range.
ThemaincontrolloopisshutdownbypullingtheITH/RUN
pin low. Releasing ITH/RUN allows an internal 0.5µA
current source to charge up the external compensation
network. When the ITH/RUN pin reaches 0.35V, the main
control loop is enabled with the ITH/RUN voltage then
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LTC1872B
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(Refer to Functional Diagram)
OPERATIO
Slope Compensation and Inductor’s Peak Current
The inductor’s peak current is determined by:
Figure 2 illustrates this result for the circuit of Figure 1
using both an LTC1872 in Burst Mode operation and an
LTC1872B (non-Burst Mode operation). At an output
current of 50mA, the Burst Mode operation part exhibits
an output ripple of approximately 80mVP-P, whereas the
non-Burst Mode operation part has an output ripple of
≈45mVP-P. At lower output current levels, the improve-
ment is even greater. This comes at a trade off of slightly
lower efficiency for the non-Burst Mode operation part.
Also notice the constant frequency operation of the
LTC1872B, even at 5% of maximum output current.
V
10 R
ITH − 0.7
IPK
=
(
)
SENSE
when the LTC1872B is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope com-
pensation begins and effectively reduces the peak induc-
torcurrent. Theamountofreductionisgivenbythecurves
in Figure 3.
110
100
90
Undervoltage Lockout
TopreventoperationoftheN-channelMOSFETbelowsafe
input voltage levels, an undervoltage lockout is incorpo-
rated into the LTC1872B. When the input supply voltage
drops below approximately 2.0V, the N-channel MOSFET
and all circuitry is turned off except the undervoltage
block, which draws only several microamperes.
80
70
60
50
I
= 0.4I
PK
RIPPLE
AT 5% DUTY CYCLE
= 0.2I
40
30
20
10
I
RIPPLE
PK
AT 5% DUTY CYCLE
Overvoltage Protection
V
IN
= 4.2V
0
10 20 30 40 50 60 70 80 90 100
The overvoltage comparator in the LTC1872B will turn the
external MOSFET off when the feedback voltage has risen
7.5% above the reference voltage of 0.8V. This compara-
tor has a typical hysteresis of 20mV.
DUTY CYCLE (%)
1872B F03
Figure 3. Maximum Output Current vs Duty Cycle
20mV AC/DIV
20mV AC/DIV
VIN = 3.3V
VOUT = 5V
IOUT = 50mA
5µs/DIV
1872B F02a
VIN = 3.3V
VOUT = 5V
IOUT = 50mA
5µs/DIV
1872B F02b
(2a) VOUT Ripple for Figure 1 Circuit
Using LTC1872 Burst Mode Operation
(2b) VOUT Ripple for Figure 1 Circuit Using
LTC1872B Non-Burst Mode Operation
Figure 2. Output Ripple Waveforms for the Circuit of Figure 1
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LTC1872B
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(Refer to Functional Diagram)
OPERATIO
Short-Circuit Protection
output. External means such as a fuse in series with the
boost inductor must be employed to handle this fault
condition.
Sincethepowerswitchinaboostconverterisnotinseries
with the power path from input to load, turning off the
switch provides no protection from a short-circuit at the
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APPLICATIONS INFORMATION
The basic LTC1872B application circuit is shown in
Figure 1. External component selection is driven by the
load requirement and begins with the selection of L1 and
V
SF
IN
+ VD
RSENSE
=
10 I
100 V
( )( OUT)(
)
OUT
R
SENSE (= R1). Next, the power MOSFET and the output
Inductor Value Calculation
diodeD1isselectedfollowedbyCIN(=C1)andCOUT(=C2).
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripplecurrent. However, thisisattheexpenseofefficiency
due to an increase in MOSFET gate charge losses.
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
Withthecurrentcomparatormonitoringthevoltagedevel-
oped across RSENSE, the threshold of the comparator
determines the inductor’s peak current. The output cur-
rent the LTC1872B can provide is given by:
The inductance value also has a direct effect on ripple
current. The ripple current, IRIPPLE, decreases with higher
inductance or frequency and increases with higher VOUT
The inductor’s peak-to-peak ripple current is given by:
.
0.12
RSENSE
IRIPPLE
V
IN
VOUT + VD
IOUT
=
−
2
V
f L
( )
VOUT + VD − V
IN
IN
IRIPPLE
=
where IRIPPLE is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section) and VD is the
forward drop of the output diode at the full rated output
current.
VOUT + VD
wherefistheoperatingfrequency.Acceptinglargervalues
of IRIPPLE allows the use of low inductances, but results in
higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is:
A reasonable starting point for setting ripple current is:
VOUT + VD
IRIPPLE = O.4 I
(
)( OUT )
VOUT + VD
V
IN
I
RIPPLE = 0.4 I
(
)
OUT MAX
(
)
V
IN
Rearranging the above equation, it becomes:
In Burst Mode operation, the ripple current is normally set
such that the inductor current is continuous during the
burst periods. Therefore, the peak-to-peak ripple current
must not exceed:
1
I
V
IN
VOUT + VD
RSENSE
=
10
( )(
)
OUT
for Duty Cycle < 40%
0.03
RSENSE
IRIPPLE
≤
However,foroperationthatisabove40%dutycycle,slope
compensation’s effect has to be taken into consideration
to select the appropriate value to provide the required
amount of current. Using the scaling factor (SF, in %) in
Figure 3, the value of RSENSE is:
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Output Diode Selection
This implies a minimum inductance of:
Under normal load conditions, the average current con-
ducted by the diode in a boost converter is equal to the
output load current:
V
0.03
RSENSE
VOUT + VD − V
IN
IN
LMIN
=
VOUT + VD
f
ID(avg) = IOUT
A smaller value than LMIN could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
Itisimportanttoadequatelyspecifythediodepeakcurrent
and average power dissipation so as not to exceed the
diode ratings.
Inductor Selection
Schottky diodes are recommended for low forward drop
and fast switching times. Remember to keep lead length
short and observe proper grounding (see Board Layout
Checklist) to avoid ringing and increased dissipation.
When selecting the inductor, keep in mind that inductor
saturation current has to be greater than the current limit
set by the current sense resistor. Also, keep in mind that
the DC resistance of the inductor will affect the efficiency.
OfftheshelfinductorsareavailablefromMurata,Coilcraft,
Toko, Panasonic, Coiltronics and many other suppliers.
CIN and COUT Selection
To prevent large input voltage ripple, a low ESR input
capacitor sized for the maximum RMS current must be
used. The maximum RMS capacitor current for a boost
converter is approximately equal to:
Power MOSFET Selection
The main selection criteria for the power MOSFET are the
threshold voltage VGS(TH), the “on” resistance RDS(ON)
,
CIN Required IRMS ≈ 0.3 I
(
)
RIPPLE
reverse transfer capacitance CRSS and total gate charge.
where IRIPPLE is as defined in the Inductor Value Calcula-
tion section.
SincetheLTC1872Bisdesignedforoperationdowntolow
input voltages, a logic level threshold MOSFET (RDS(ON)
guaranteed at VGS = 2.5V) is required for applications that
workclosetothisvoltage.WhentheseMOSFETsareused,
make sure that the input supply to the LTC1872B is less
than the absolute maximum VGS rating, typically 8V.
Note that capacitor manufacturer’s ripple current ratings
are often based on 2000 hours of life. This makes it
advisable to further derate the capacitor, or to choose a
capacitor rated at a higher temperature than required.
Several capacitors may be paralleled to meet the size or
height requirements in the design. Due to the high operat-
ing frequency of the LTC1872B, ceramic capacitors can
also be used for CIN. Always consult the manufacturer if
there is any question.
The required minimum RDS(ON) of the MOSFET is gov-
erned by its allowable power dissipation given by:
PP
RDS(ON)
2
1+ δp
DC I
(
)
(
)
IN
where PP is the allowable power dissipation and δp is the
temperature dependency of RDS(ON). (1 + δp) is generally
given for a MOSFET in the form of a normalized RDS(ON) vs
temperature curve, but δp = 0.005/°C can be used as an
approximation for low voltage MOSFETs. DC is the maxi-
mum operating duty cycle of the LTC1872B.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:
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LTC1872B
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APPLICATIONS INFORMATION
amount of change as the supply is reduced down to 2V.
Also shown in Figure 4 is the effect of VIN on VREF as VIN
goes below 2.3V.
VOUT + VD IRIPPLE
∆VOUT ≈ IO •
+
•
V
2
IN
1
2
Setting Output Voltage
2
1
ESR2 +
TheLTC1872Bdevelopsa0.8Vreferencevoltagebetween
thefeedback(Pin3)terminalandground(seeFigure5).By
selecting resistor R1, a constant current is caused to flow
through R1 and R2 to set the overall output voltage. The
regulated output voltage is determined by:
2πfCOUT
where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the induc-
tor.
R2
VOUT = 0.8V 1+
R1
Manufacturers such as Nichicon, United Chemicon and
Sanyoshouldbeconsideredforhighperformancethrough-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higherprice.TheoutputcapacitorRMScurrentisapproxi-
mately equal to:
Formostapplications, an80kresistorissuggestedforR1.
To prevent stray pickup, locate resistors R1 and R2 close
to LTC1872B.
105
V
REF
IPK • DC −DC2
100
95
90
85
80
75
V
where IPK is the peak inductor current and DC is the switch
duty cycle.
ITH
Whenusingelectrolyticoutputcapacitors, iftherippleand
ESR requirements are met, there is likely to be far more
capacitance than required.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
surface mount configurations. An excellent choice of
tantalum capacitors is the AVX TPS and KEMET T510
series of surface mount tantalum capacitors. Also,
ceramic capacitors in X5R pr X7R dielectrics offer excel-
lent performance.
2.0
2.2
2.4
2.6
2.8
3.0
INPUT VOLTAGE (V)
1872B F04
Figure 4. Line Regulation of VREF and VITH
V
OUT
R2
R1
LTC1872B
3
V
FB
Low Supply Operation
1872B F05
Although the LTC1872B can function down to approxi-
mately 2.0V, the maximum allowable output current is
reducedwhenVIN decreasesbelow3V. Figure4showsthe
Figure 5. Setting Output Voltage
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APPLICATIONS INFORMATION
Efficiency Considerations
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
oftenusefultoanalyzeindividuallossestodeterminewhat
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Transition Loss = 2(VIN)2IIN(MAX) RSS
(f)
C
Other losses, including CIN and COUT ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
Efficiency = 100% – (η1 + η2 + η3 + ...)
where η1, η2, etc. are the individual losses as a percent-
age of input power.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1872B. These items are illustrated graphically in the
layout diagram in Figure 6. Check the following in your
layout:
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1872B circuits: 1) LTC1872B DC bias cur-
rent, 2) MOSFET gate charge current, 3) I2R losses and 4)
voltage drop of the output diode.
1. The Schottky diode should be closely connected
between the output capacitor and the drain of the
external MOSFET.
1. The VIN current is the DC supply current, given in the
electricalcharacteristics, thatexcludesMOSFETdriver
and control currents. VIN current results in a small loss
which increases with VIN.
2. The (+) plate of CIN should connect to the sense
resistor as closely as possible. This capacitor provides
AC current to the inductor.
2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each time
a MOSFET gate is switched from low to high to low
again, a packet of charge, dQ, moves from VIN to
ground. The resulting dQ/dt is a current out of VIN
which is typically much larger than the contoller’s DC
supply current. In continuous mode, IGATECHG = f(Qp).
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current sense resistor. The
MOSFET RDS(ON) multiplied by duty cycle times the
average output current squared can be summed with
I2RlossesintheinductorESRinserieswiththecurrent
sense resistor.
3. The input decoupling capacitor (0.1µF) should be
connected closely between VIN (Pin 5) and ground
(Pin 2).
4. Connect the end of RSENSE as close to VIN (Pin 5) as
possible. The VIN pin is the SENSE+ of the current
comparator.
5. The trace from SENSE– (Pin 4) to the Sense resistor
should be kept short. The trace should connect close
to RSENSE
.
6. Keep the switching node NGATE away from sensitive
small signal nodes.
4. The output diode is a major source of power loss at
high currents. The diode loss is calculated by multiply-
ing the forward voltage by the load current.
7. The VFB pin should connect directly to the feedback
resistors. The resistive divider R1 and R2 must be
connected between the (+) plate of COUT and signal
ground.
9
LTC1872B
U
W U U
APPLICATIONS INFORMATION
V
V
IN
1
6
5
4
I
/RUN NGATE
LTC1872B
TH
M1
L1
R
S
2
3
R
ITH
GND
V
IN
+
0.1µF
C
IN
D1
–
V
SENSE
C
FB
ITH
OUT
+
R2
C
OUT
R1
BOLD LINES INDICATE HIGH CURRENT PATHS
1872B F06
Figure 6. LTC1872B Layout Diagram (See PC Board Layout Checklist)
U
TYPICAL APPLICATIO S
LTC1872B 3-Cell White LED Driver
V
IN
= 3 AA CELLS ≈ 2.7V TO 4.8V
C1
10µF
10V
R1
0.27Ω
AA
AA
AA
1
5
L1
I
/RUN
V
TH
IN
–
150µH
V
≈ 28.8V
OUT
LTC1872B
GND
10k
220pF
(WITH 8 LEDs)
2
3
4
6
SENSE
+
C2
15µF
35V
C3
15mA
D0
M1
V
NGATE
0.1µF
CERAMIC
FB
D1
D2
1 TO 8
WHITE
LEDs
•
•
•
D8
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: AVX TPSD156M035R0300
D0: MOTOROLA MBR0540
L1: COILCRAFT DO1608C-154
M1: Si9804
R1: DALE 0.25W
53.6Ω
1872B TA04
D1-D7: CMD333UWC
10
LTC1872B
U
TYPICAL APPLICATIO S
LTC1872B 12V/500mA Boost Converter
V
IN
3V TO 9.8V
C1
R1
0.033Ω
10µF
10V
1
5
L1
I
/RUN
V
TH
IN
–
10µH
LTC1872B
GND
10k
V
2
3
4
6
OUT
SENSE
12V
+
C2
47µF
16V
220pF
D1
M1
V
NGATE
FB
1.1M
1872B TA02
78.7k
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: AVX TPSE476M016R0150
D1: IR10BQ015
L1: COILTRONICS UP2B-100
M1: Si9804DV
R1: DALE 0.25W
U
PACKAGE DESCRIPTION
S6 Package
6-Lead Plastic SOT-23
(Reference LTC DWG # 05-08-1634)
(Reference LTC DWG # 05-08-1636)
2.80 – 3.10
(.110 – .118)
(NOTE 3)
SOT-23
(Original)
SOT-23
(ThinSOT)
.90 – 1.45
1.00 MAX
A
A1
A2
L
(.035 – .057)
(.039 MAX)
.00 – 0.15
(.00 – .006)
.01 – .10
(.0004 – .004)
2.60 – 3.00
1.50 – 1.75
(.102 – .118) (.059 – .069)
(NOTE 3)
.90 – 1.30
(.035 – .051)
.80 – .90
(.031 – .035)
PIN ONE ID
.35 – .55
(.014 – .021)
.30 – .50 REF
(.012 – .019 REF)
.95
(.037)
REF
.25 – .50
(.010 – .020)
(6PLCS, NOTE 2)
.20
(.008)
A2
A
DATUM ‘A’
1.90
(.074)
REF
L
.09 – .20
(.004 – .008)
(NOTE 2)
A1
S6 SOT-23 0401
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
3. DRAWING NOT TO SCALE
4. DIMENSIONS ARE INCLUSIVE OF PLATING
5. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
6. MOLD FLASH SHALL NOT EXCEED .254mm
7. PACKAGE EIAJ REFERENCE IS:
SC-74A (EIAJ) FOR ORIGINAL
JEDEL MO-193 FOR THIN
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
11
LTC1872B
U
TYPICAL APPLICATIO S
LTC1872B –2.5V to 3.3V/0.5A Boost Converter
R1
0.034Ω
C2
2× 100µF
+
1
5
10V
L1
I
/RUN
V
TH
IN
–
4.7µH
LTC1872B
GND
10k
V
3.3V
0.5A
OUT
2
3
4
6
SENSE
220pF
C1
D1
332k
V
NGATE
M1
FB
+
U1
0.1µF
CERAMIC
100µF
10V
80.6k
180k
V
IN
–2.5V
1872B TA03
C1, C2: AVX TPSE107M010R0100
D1: MOTOROLA MBR2045CT
L1: COILTRONICS UP2B-4R7
M1: Si9804DV
R1: DALE 0.25W
U1: PANASONIC 2SB709A
LTC1872B 2.7V to 9.8V Input to 3.3V/1.2A Output SEPIC Converter
V
IN
2.7V TO 9.8V
C
IN
R
CS
0.03Ω
10µF
C
C1
220pF
10V, X5R
R
C1
10k
V
1
5
OUT
I
/RUN
TH
V
IN
3.3V/1.2A
+
C
O1
D1
L1A
L1B
LTC1872B
GND
180µF
MBRM120
2
3
4
6
4V, SP
–
SENSE
CS
4.7µF
10V
V
NGATE
FB
R
f2
80.6k
R
f1
252k
M1
1872B TA05
C
O1
, CS; TOKO, MURATA OR TAIYO YUDEN
: PANASONIC EEFUE0G181R
FOR V
= 5V CHANGE
IN
C
OUT
R
TO 427kΩ AND
f1
L1: BH ELECTRONICS 511-1012
M1: IRLMS2002
C
TO 150µF, 6V PANASONIC
O1
SP TYPE CAPACITOR
R
: DALE OR IRC
CS
RELATED PARTS
PART NUMBER
LT1304
DESCRIPTION
COMMENTS
120µA Quiescent Current, 1.5V ≤ V ≤ 8V
Micropower DC/DC Converter with Low-Battery Detector
1.7MHz, Single Cell Micropower DC/DC Converter
1.4MHz, Single Cell DC/DC Converter in 5-Lead ThinSOT
Low Voltage Current Mode PWM Controller
IN
LT1610
30µA Quiescent Current, V Down to 1V
IN
LT1613
Internally Compensated, V Down to 1V
IN
LT1619
8-Lead MSOP Package, 1.9V ≤ V ≤ 18V
IN
LT1680
High Power DC/DC Step-Up Controller
Operation Up to 60V, Fixed Frequency Current Mode
LTC1624
LT1615
High Efficiency SO-8 N-Channel Switching Regulator Controller
Micropower Step-Up DC/DC Converter in ThinSOT
8-Pin N-Channel Drive, 3.5V ≤ V ≤ 36V
IN
20µA Quiescent Current, V Down to 1V
IN
LTC1700
LTC1772
LTC1872
No R
Synchronous Current Mode DC/DC Step-Up Controller
95% Efficient, 0.9V ≤ V ≤ 5V, 550kHz Operation
IN
SENSE
Constant Frequency Current Mode Step-Down DC/DC Controller
V
2.5V to 9.8V, I
up to 4A, ThinSOT Package
OUT
IN
Constant Frequency Current Mode Step-Up DC/DC Controller in ThinSOT With Burst Mode Operation for Higher Efficiency at Light
Load Current
LTC3401/LTC3402 1A/2A, 3MHz Micropower Synchronous Boost Converter
10-Lead MSOP Package, 0.5V ≤ V ≤ 5V
IN
1872bf LT/TP 0601 2K • PRINTED IN USA
12 LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
●
●
LINEAR TECHNOLOGY CORPORATION 2001
(408)432-1900 FAX:(408)434-0507 www.linear-tech.com
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