LTC3406B-1.2 [Linear]
1.5MHz, 600mA Synchronous Step-Down Regulator in ThinSOT; 为1.5MHz ， 600mA同步降压型稳压器采用ThinSOT
| 型号： | LTC3406B-1.2 |
| 厂家： | 
Linear |
| 描述： | 1.5MHz, 600mA Synchronous Step-Down Regulator in ThinSOT |
| 文件： | 总12页 (文件大小：267K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC3406B-1.2
1.5MHz, 600mA
Synchronous Step-Down
Regulator in ThinSOT
U
FEATURES
DESCRIPTIO
The LTC®3406B-1.2 is a high efficiency monolithic syn-
chronous buck regulator using a constant frequency,
current mode architecture. Supply current with no load is
300µA dropping to <1µA in shutdown. The 2.5V to 5.5V
inputvoltagerangemakestheLTC3406B-1.2ideallysuited
forsingleLi-Ionbattery-poweredapplications. 100%duty
cycle provides low dropout operation, extending battery
lifeinportablesystems.PWMpulseskippingmodeopera-
tion provides very low output ripple voltage for noise
sensitive applications.
■
High Efficiency: Up to 96%
■
600mA Output Current at VIN = 3V
■
2.5V to 5.5V Input Voltage Range
■
1.5MHz Constant Frequency Operation
■
No Schottky Diode Required
■
Low Quiescent Current: 300µA
■
Shutdown Mode Draws <1µA Supply Current
■
Current Mode Operation for Excellent Line and
Load Transient Response
■
Overtemperature Protected
Low Profile (1mm) ThinSOTTM Package
■
Switching frequency is internally set at 1.5MHz, allowing
the use of small surface mount inductors and capacitors.
The internal synchronous switch increases efficiency and
eliminates the need for an external Schottky diode. The
LTC3406B-1.2isavailableinalowprofile(1mm)ThinSOT
package.
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APPLICATIO S
■
Cellular Telephones
■
Personal Information Appliances
■
Wireless and DSL Modems
, LTC and LT are registered trademarks of Linear Technology Corporation. All other
trademarks are the property of their respective owners. ThinSOT is a trademark of Linear
Technology Corporation. Protected by U.S. Patents including 5481178, 6580258, 6304066,
6127815, 6498466, 6611131.
■
Digital Still Cameras
■
MP3 Players
■
Portable Instruments
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TYPICAL APPLICATIO
Efficiency and Power Loss
100
90
80
70
60
50
40
30
20
10
1
High Efficiency Step-Down Converter
2.2µH
EFFICIENCY
V
V
IN
OUT
V
SW
LTC3406B-1.2
RUN
IN
0.1
2.7V TO 5.5V
C
1.2V
C
OUT
IN
10µF 600mA
4.7µF
CER
CER
0.01
0.001
0.0001
V
OUT
3406B12 TA01a
GND
POWER LOSS
V
V
V
= 2.7V
= 3.6V
= 4.2V
IN
IN
IN
0.1
1000
1
10
100
LOAD CURRENT (mA)
3406B12 TA01b
sn3406b12 3406b12fs
1
LTC3406B-1.2
W W U W
U W
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ABSOLUTE AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Supply Voltage .................................. –0.3V to 6V
RUN, VOUT Voltages................................... –0.3V to VIN
SW Voltage (DC) ......................... –0.3V to (VIN + 0.3V)
P-Channel Switch Source Current (DC) ............. 800mA
N-Channel Switch Sink Current (DC) ................. 800mA
Peak SW Sink and Source Current (VIN = 3V)........ 1.3A
Operating Temperature Range (Note 2) .. –40°C to 85°C
Junction Temperature (Notes 3, 5) ...................... 125°C
Storage Temperature Range ................ –65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
ORDER PART
TOP VIEW
NUMBER
RUN 1
GND 2
SW 3
5 V
4 V
OUT
IN
LTC3406BES5-1.2
S5 PART MARKING
LTBMR
S5 PACKAGE
5-LEAD PLASTIC TSOT-23
TJMAX = 125°C, θJA = 250°C/ W, θJC = 90°C/ W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
The ● denotes specifications which apply over the full operating
ELECTRICAL CHARACTERISTICS
temperature range, otherwise specifications are TA = 25°C. VIN = 3.6V unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
MIN
1.164
2.5
TYP
1.2
6.25
0.04
1
MAX
1.236
10
UNITS
V
Regulated Output Voltage
Output Overvoltage Lockout
Output Voltage Line Regulation
Peak Inductor Current
●
●
V
%
OUT
∆V
∆V
∆V
= V
– V
OVL
OVL
OVL OUT
V
V
= 2.5V to 5.5V
0.4
%/V
A
OUT
IN
IN
I
= 3V, V
= 1.08V, Duty Cycle < 35%
0.75
2.5
1.25
PK
OUT
V
V
Output Voltage Load Regulation
Input Voltage Range
0.5
%
LOADREG
IN
●
●
5.5
V
I
Input DC Bias Current
(Note 4)
S
V
V
= 1.08V
300
0.1
400
1
µA
µA
OUT
RUN
Shutdown
= 0V, V = 5.5V
IN
f
Oscillator Frequency
V
V
= 1.2V
= 0V
1.2
0.3
1.5
210
1.8
MHz
kHz
OSC
OUT
OUT
R
R
R
R
of P-Channel FET
of N-Channel FET
I
I
= 100mA
0.4
0.35
±0.01
1
0.5
0.45
±1
Ω
Ω
PFET
NFET
LSW
DS(ON)
SW
SW
= –100mA
DS(ON)
I
SW Leakage
V
= 0V, V = 0V or 5V, V = 5V
µA
V
RUN
SW
IN
V
RUN Threshold
RUN Leakage Current
●
●
1.5
±1
RUN
I
±0.01
µA
RUN
sn3406b12 3406b12fs
2
LTC3406B-1.2
ELECTRICAL CHARACTERISTICS
Note 1: Absolute Maximum Ratings are those values beyond which the life
Note 4: Dynamic supply current is higher due to the gate charge being
of a device may be impaired.
delivered at the switching frequency.
Note 2: The LTC3406BE-1.2 is guaranteed to meet performance
specifications from 0°C to 70°C. Specifications over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls.
Note 5: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
Note 3: T is calculated from the ambient temperature T and power
J
A
dissipation P according to the following formula:
D
LTC3406B-1.2: T = T + (P )(250°C/W)
J
A
D
U W
TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1)
Reference Voltage vs
Temperature
Efficiency vs Input Voltage
Efficiency vs Output Current
95
90
85
80
75
70
65
60
55
50
1.228
1.218
1.208
1.198
1.188
1.178
1.168
100
90
80
70
60
50
40
30
20
10
V
T
= 1.2V
V
= 3.6V
OUT
= 25°C
IN
A
V
= 2.7V
IN
I
= 100mA
OUT
I
= 600mA
OUT
V
= 3.6V
IN
I
= 10mA
V
= 4.2V
OUT
IN
2
3
4
5
6
–50 –25
25
50
TEMPERATURE (°C)
75
100 125
0
0.1
1000
1
10
100
INPUT VOLTAGE (V)
OUTPUT CURRENT (mA)
3406B12 G01
3406B12 G03
3406B12 GO2
Oscillator Frequency vs
Temperature
Oscillator Frequency vs
Supply Voltage
Output Voltage vs Load Current
1.70
1.65
1.60
1.55
1.50
1.45
1.40
1.35
1.30
1.8
1.7
1.6
1.5
1.4
1.3
1.2
1.224
1.214
1.204
1.194
1.184
1.174
T
= 25°C
V
= 3.6V
A
IN
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
75
2
3
4
5
6
0
700
9001000
800
100 200 300 400 500 600
SUPPLY VOLTAGE (V)
LOAD CURRENT (mA)
3406B12 G05
3406B12 G06
3406B12 G04
sn3406b12 3406b12fs
3
LTC3406B-1.2
TYPICAL PERFOR A CE CHARACTERISTICS
U W
(From Figure 1)
Dynamic Supply Current vs
Supply Voltage
RDS(ON) vs Input Voltage
RDS(ON) vs Temperature
0.7
0.6
0.7
0.6
400
380
360
340
320
300
280
260
240
220
200
T
A
= 25°C
I
= 0A
LOAD
A
V
IN
= 2.7V
T
= 25°C
V
IN
= 3.6V
V
IN
= 4.2V
0.5
0.4
0.3
0.2
0.1
0.5
0.4
0.3
0.2
0.1
MAIN
SWITCH
SYNCHRONOUS
SWITCH
MAIN SWITCH
SYNCHRONOUS SWITCH
0
0
50
100 125
–50 –25
0
25
75
5
7
0
1
2
3
4
6
2
3
4
5
6
TEMPERATURE (°C)
INPUT VOLTAGE (V)
SUPPLY VOLTAGE (V)
3406B12 G08
3406B12 G07
3406B12 G09
Dynamic Supply Current vs
Temperature
Switch Leakage vs Temperature
340
320
300
280
260
240
220
200
300
250
200
150
V
I
= 3.6V
= 0A
V
= 5.5V
IN
LOAD
IN
RUN = 0V
100
50
0
MAIN SWITCH
SYNCHRONOUS SWITCH
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
75
50
TEMPERATURE (°C)
100 125
–50 –25
0
25
75
3406B12 G11
3406B12 G10
Switch Leakage vs Input Voltage
Discontinuous Operation
120
100
80
60
40
20
0
RUN = 0V
T
= 25°C
A
SW
SYNCHRONOUS
SWITCH
2V/DIV
V
OUT
10mV/DIV
AC COUPLED
MAIN
SWITCH
I
L
200mA/DIV
3406B12 G13
1µs/DIV
V
= 3.6V
= 50mA
IN
I
LOAD
0
2
3
4
5
6
1
INPUT VOLTAGE (V)
3406B12 G12
sn3406b12 3406b12fs
4
LTC3406B-1.2
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1a Except for the Resistive Divider Resistor Values)
Start-Up from Shutdown
Load Step
Load Step
V
OUT
V
OUT
RUN
2V/DIV
100mV/DIV
100mV/DIV
AC COUPLED
AC COUPLED
V
OUT
I
1V/DIV
I
L
L
500mA/DIV
500mA/DIV
I
L
500mA/DIV
I
I
LOAD
500mA/DIV
LOAD
500mA/DIV
3406B12 G15
3406B12 G14
3406B12 G16
25µs/DIV
= 0mA TO 600mA
LOAD
50µs/DIV
25µs/DIV
= 50mA TO 600mA
V = 3.6V
IN
V
I
= 3.6V
LOAD
V
I
= 3.6V
LOAD
IN
IN
I
= 600mA
Load Step
Load Step
V
V
OUT
OUT
100mV/DIV
100mV/DIV
AC COUPLED
AC COUPLED
I
I
L
L
500mA/DIV
500mA/DIV
I
I
LOAD
LOAD
500mA/DIV
500mA/DIV
3406B12 G17
3406B12 G18
25µs/DIV
25µs/DIV
= 200mA TO 600mA
V
I
= 3.6V
LOAD
V
I
= 3.6V
LOAD
IN
IN
= 100mA TO 600mA
U
U
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PI FU CTIO S
RUN (Pin 1): Run Control Input. Forcing this pin above
1.5V enables the part. Forcing this pin below 0.3V shuts
down the device. In shutdown, all functions are disabled
drawing <1µA supply current. Do not leave RUN floating.
VIN (Pin 4): Main Supply Pin. Must be closely decoupled
to GND, Pin 2, with a 2.2µF or greater ceramic capacitor.
VOUT (Pin 5): Output Voltage Feedback Pin. An internal
resistive divider divides the output voltage down for com-
parison to the internal reference voltage.
GND (Pin 2): Ground Pin.
SW (Pin 3): Switch Node Connection to Inductor. This pin
connects to the drains of the internal main and synchro-
nous power MOSFET switches.
sn3406b12 3406b12fs
5
LTC3406B-1.2
U
U
W
FU CTIO AL DIAGRA
SLOPE
COMP
OSC
OSC
V
IN
4
FREQ
–
+
SHIFT
V
OUT
5
+
–
5Ω
0.8V
+
–
60k
I
COMP
EA
FB
Q
Q
S
R
120k
SWITCHING
LOGIC
AND
RS LATCH
V
ANTI-
SHOOT-
THRU
IN
BLANKING
CIRCUIT
–
OV
SW
3
OVDET
+
RUN
1
0.8V + ∆V
OVL
0.8V REF
+
–
SHUTDOWN
I
RCMP
2
GND
3406B12 BD
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OPERATIO
(Refer to Functional Diagram)
2.2µH*
V
V
as indicated by the current reversal comparator IRCMP, or
the beginning of the next clock cycle. The comparator
OVDET guards against transient overshoots >6.25% by
turning the main switch off and keeping it off until the fault
is removed.
IN
OUT
4
3
2.7V
TO 5.5V
1.2V
V
SW
IN
†
C
C
**
4.7µF
OUT
600mA
IN
10µF
LTC3406B-1.2
CER
CER
1
5
3406B12 F01
RUN
V
OUT
GND
2
*MURATA LQH3C2R2M24
**TAIYO YUDEN JMK212BJ475MG
Pulse Skipping Mode Operation
†TAIYO YUDEN JMK316BJ106ML
At light loads, the inductor current may reach zero or re-
verse on each pulse. The bottom MOSFET is turned off by
the current reversal comparator, IRCMP, and the switch
voltage will ring. This is discontinuous mode operation,
and is normal behavior for the switching regulator. At very
lightloads,theLTC3406B-1.2willautomaticallyskippulses
inpulseskippingmodeoperationtomaintainoutputregu-
lation. Refer to LTC3406-1.2 data sheet if Burst Mode op-
eration is preferred.
Figure 1. Typical Application
Main Control Loop
The LTC3406B-1.2 uses a constant frequency, current
mode step-down architecture. Both the main (P-channel
MOSFET)andsynchronous(N-channelMOSFET)switches
are internal. During normal operation, the internal top
power MOSFET is turned on each cycle when the oscillator
sets the RS latch, and turned off when the current com-
parator, ICOMP, resets the RS latch. The peak inductor
current at which ICOMP resets the RS latch, is controlled by
the output of error amplifier EA. When the load current
increases, it causes a slight decrease in the feedback
voltage, FB, relative to the 0.8V reference, which in turn
causes the EA amplifier’s output voltage to increase until
the average inductor current matches the new load cur-
rent. While the top MOSFET is off, the bottom MOSFET is
turnedonuntileithertheinductorcurrentstartstoreverse,
Short-Circuit Protection
Whentheoutputisshortedtoground, thefrequencyofthe
oscillator is reduced to about 210kHz, 1/7 the nominal
frequency. This frequency foldback ensures that the in-
ductorcurrenthasmoretimetodecay, therebypreventing
runaway. The oscillator’s frequency will progressively
increase to 1.5MHz when VOUT rises above 0V.
sn3406b12 3406b12fs
6
LTC3406B-1.2
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APPLICATIO S I FOR ATIO
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The basic LTC3406B-1.2 application circuit is shown in
Figure 1. External component selection is driven by the
load requirement and begins with the selection of L fol-
Table 1. Representative Surface Mount Inductors
PART
NUMBER
VALUE
(µH)
DCR
MAX DC
SIZE
3
(Ω MAX) CURRENT (A) W × L × H (mm )
Sumida
CDRH3D16
1.5
2.2
3.3
4.7
0.043
0.075
0.110
0.162
1.55
1.20
1.10
0.90
3.8 × 3.8 × 1.8
lowed by CIN and COUT
.
Inductor Selection
For most applications, the value of the inductor will fall in
the range of 1µH to 4.7µH. Its value is chosen based on the
desired ripple current. Large value inductors lower ripple
current and small value inductors result in higher ripple
currents. Higher VIN or VOUT also increases the ripple
currentasshowninequation1. Areasonablestartingpoint
for setting ripple current is ∆IL = 240mA (40% of 600mA).
Sumida
CMD4D06
2.2
3.3
4.7
0.116
0.174
0.216
0.950
0.770
0.750
3.5 × 4.3 × 0.8
Panasonic
ELT5KT
3.3
4.7
0.17
0.20
1.00
0.95
4.5 × 5.4 × 1.2
2.5 × 3.2 × 2.0
Murata
LQH3C
1.0
2.2
4.7
0.060
0.097
0.150
1.00
0.79
0.65
⎛
VOUT
V
IN
⎞
1
∆IL =
VOUT 1−
CIN and COUT Selection
⎜
⎟
(1)
f L
( )( )
⎝
⎠
Incontinuousmode,thesourcecurrentofthetopMOSFET
is a square wave of duty cycle VOUT/VIN. To prevent large
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
RMS capacitor current is given by:
The DC current rating of the inductor should be at least
equal to the maximum load current plus half the ripple
current to prevent core saturation. Thus, a 720mA rated
inductorshouldbeenoughformostapplications(600mA
+ 120mA). For better efficiency, choose a low DC-resis-
tance inductor.
1/2
]
V
V − V
OUT
(
)
[
OUT IN
CIN requiredIRMS ≅ IOMAX
V
IN
Inductor Core Selection
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is com-
monlyusedfordesignbecauseevensignificantdeviations
do not offer much relief. Note that the capacitor
manufacturer’s ripple current ratings are often based on
2000hoursoflife.Thismakesitadvisabletofurtherderate
the capacitor, or choose a capacitor rated at a higher
temperature than required. Always consult the manufac-
turer if there is any question.
Different core materials and shapes will change the size/
current and price/current relationship of an inductor.
Toroid or shielded pot cores in ferrite or permalloy mate-
rials are small and don’t radiate much energy, but gener-
ally cost more than powdered iron core inductors with
similarelectricalcharacteristics. Thechoiceofwhichstyle
inductor to use often depends more on the price vs size
requirements and any radiated field/EMI requirements
than on what the LTC3406B-1.2 requires to operate. Table
1 shows some typical surface mount inductors that work
well in LTC3406B-1.2 applications.
The selection of COUT is driven by the required effective
series resistance (ESR).
sn3406b12 3406b12fs
7
LTC3406B-1.2
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APPLICATIO S I FOR ATIO
Typically, once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.Theoutputripple∆VOUT isdeter-
mined by:
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage charac-
teristics of all the ceramics for a given value and size.
⎛
1
⎞
Efficiency Considerations
∆VOUT ≅ ∆I ESR +
⎜
⎟
L
⎝
8fCOUT
⎠
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
oftenusefultoanalyzeindividuallossestodeterminewhat
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. For a fixed output
voltage, the output ripple is highest at maximum input
voltage since ∆IL increases with input voltage.
Efficiency = 100% – (L1 + L2 + L3 + ...)
Aluminum electrolytic and dry tantalum capacitors are
bothavailableinsurfacemountconfigurations.Inthecase
oftantalum,itiscriticalthatthecapacitorsaresurgetested
for use in switching power supplies. An excellent choice is
the AVX TPS series of surface mount tantalum. These are
specially constructed and tested for low ESR so they give
the lowest ESR for a given volume. Other capacitor types
include Sanyo POSCAP, Kemet T510 and T495 series, and
Sprague 593D and 595D series. Consult the manufacturer
for other specific recommendations.
whereL1, L2, etc. aretheindividuallossesasapercentage
of input power.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of the
losses in LTC3406B-1.2 circuits: VIN quiescent current
and I2R losses. The VIN quiescent current loss dominates
the efficiency loss at very low load currents whereas the
I2R loss dominates the efficiency loss at medium to high
load currents. In a typical efficiency plot, the efficiency
curveatverylowloadcurrentscanbemisleadingsincethe
actual power lost is of no consequence as illustrated in
Figure 2.
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. Because the
LTC3406B-1.2’s control loop does not depend on the
output capacitor’s ESR for stable operation, ceramic ca-
pacitors can be used freely to achieve very low output
ripple and small circuit size.
1
0.1
0.01
0.001
However, care must be taken when ceramic capacitors are
usedattheinputandtheoutput.Whenaceramiccapacitor
is used at the input and the power is supplied by a wall
adapter through long wires, a load step at the output can
induce ringing at the input, VIN. At best, this ringing can
couple to the output and be mistaken as loop instability. At
worst, a sudden inrush of current through the long wires
can potentially cause a voltage spike at VIN, large enough
to damage the part.
V
V
V
= 2.7V
= 3.6V
= 4.2V
IN
IN
IN
0.0001
0.1
1
10
100
1000
LOAD CURRENT (mA)
3406B12 F02
Figure 2. Power Loss vs Load Current
sn3406b12 3406b12fs
8
LTC3406B-1.2
W U U
APPLICATIO S I FOR ATIO
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1. The VIN quiescent current is due to two components:
the DC bias current as given in the electrical character-
istics and the internal main switch and synchronous
switch gate charge currents. The gate charge current
results from switching the gate capacitance of the
internal power MOSFET switches. Each time the gate is
switched from high to low to high again, a packet of
charge, dQ, moves from VIN to ground. The resulting
dQ/dtisthecurrentoutofVINthatistypicallylargerthan
To avoid the LTC3406B-1.2 from exceeding the maximum
junction temperature, the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated exceeds the
maximum junction temperature of the part. The tempera-
ture rise is given by:
TR = (PD)(θJA)
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
the DC bias current. In continuous mode, IGATECHG
=
f(QT + QB) where QT and QB are the gate charges of the
internal top and bottom switches. Both the DC bias and
gate charge losses are proportional to VIN and thus
their effects will be more pronounced at higher supply
voltages.
The junction temperature, TJ, is given by:
TJ = TA + TR
where TA is the ambient temperature.
2. I2R losses are calculated from the resistances of the
internal switches, RSW, and external inductor RL. In
continuous mode, the average output current flowing
through inductor L is “chopped” between the main
switch and the synchronous switch. Thus, the series
resistance looking into the SW pin is a function of both
top and bottom MOSFET RDS(ON) and the duty cycle
(DC) as follows:
As an example, consider the LTC3406B-1.2 with an input
voltage of 2.7V, a load current of 600mA and an ambient
temperature of 70°C. From the typical performance graph
of switch resistance, the RDS(ON) at 70°C is approximately
0.52Ω for the P-channel switch and 0.42Ω for the
N-channel switch. Using equation (2) to find the series
resistance looking into the SW pin gives:
RSW = 0.52Ω(0.44) + 0.42Ω(0.56) = 0.46Ω
Therefore, power dissipated by the part is:
PD = ILOAD2 • RSW = 165.6mW
(2)
RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
beobtainedfromtheTypicalPerformanceCharateristics
curves. Thus, to obtain I2R losses, simply add RSW to
RL and multiply the result by the square of the average
output current.
For the SOT-23 package, the θJA is 250°C/W. Thus, the
junction temperature of the regulator is:
TJ = 70°C + (0.1656)(250) = 111.4°C
Other losses including CIN and COUT ESR dissipative
losses and inductor core losses generally account for less
than 2% total additional loss.
which is below the maximum junction temperature of
125°C.
Note that at higher supply voltages, the junction tempera-
ture is lower due to reduced switch resistance (RSW).
Thermal Considerations
In most applications the LTC3406B-1.2 does not dissi-
pate much heat due to its high efficiency. But, in applica-
tionswheretheLTC3406B-1.2isrunningathighambient
temperature with low supply voltage, the heat dissipated
may exceed the maximum junction temperature of the
part. If the junction temperature reaches approximately
150°C,bothpowerswitcheswillbeturnedoffandtheSW
node will become high impedance.
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to (∆ILOAD • ESR), where ESR is the effective series
resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT, which generates a feedback error signal.
sn3406b12 3406b12fs
9
LTC3406B-1.2
W U U
U
APPLICATIO S I FOR ATIO
The regulator loop then acts to return VOUT to its steady-
state value. During this recovery time VOUT can be moni-
toredforovershootorringingthatwouldindicateastability
problem. For a detailed explanation of switching control
loop theory, see Application Note 76.
3. Keepthe(–)platesofCIN andCOUT ascloseaspossible.
Design Example
As a design example, assume the LTC3406B-1.2 is used
in a single lithium-ion battery-powered cellular phone
application. The VIN will be operating from a maximum of
4.2V down to about 2.7V. The load current requirement
is a maximum of 0.6A but most of the time it will be in
standbymode, requiringonly2mA. Efficiencyatbothlow
and high load currents is important. With this informa-
tion we can calculate L using equation (1),
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
dischargedbypasscapacitorsareeffectivelyputinparallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25 • CLOAD).
Thus, a 10µF capacitor charging to 3.3V would require a
250µs rise time, limiting the charging current to about
130mA.
1
⎛
1.2V⎞
V ⎠
IN
L =
1.2V 1−
⎜
⎟
(3)
f ∆I
( )(
⎝
)
L
Substituting VIN = 4.2V, ∆IL = 240mA and f = 1.5MHz in
equation (3) gives:
PC Board Layout Checklist
1.2V
1.5MHz(240mA)
1.2V
4.2V
⎛
⎜
⎝
⎞
⎟
⎠
L =
1−
= 2.38µH
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3406B-1.2. These items are also illustrated graphi-
callyin Figures3and4. Checkthefollowinginyourlayout:
A 2.2µH inductor works well for this application. For best
efficiency choose a 720mA or greater inductor with less
than 0.2Ω series resistance.
1. The power traces, consisting of the GND trace, the SW
trace and the VIN trace should be kept short, direct and
wide.
CIN will require an RMS current rating of at least 0.3A ≅
ILOAD(MAX)/2 at temperature and COUT will require an ESR
of less than 0.25Ω. In most cases, a ceramic capacitor will
satisfy this requirement.
2. Does the (+) plate of CIN connect to VIN as closely as
possible? This capacitor provides the AC current to the
internal power MOSFETs.
VIA TO V
OUT
V
IN
VIA TO V
1
IN
RUN
LTC3406B-1.2
2
3
5
4
PIN 1
GND
V
OUT
–
+
C
V
OUT
OUT
LTC3406B-1.2
V
OUT
SW
V
IN
L1
SW
L1
C
IN
V
IN
C
OUT
C
IN
3406B12 F03
GND
BOLD LINES INDICATE HIGH CURRENT PATHS
3406B12 F04
Figure 3. LTC3406B-1.2 Layout Diagram
Figure 4. LTC3406B-1.2 Suggested Layout
sn3406b12 3406b12fs
10
LTC3406B-1.2
U
TYPICAL APPLICATIO S
Single Li-Ion 1.2V/600mA Regulator for Lowest Profile, ≤1mm High
†
2.2µH
4
3
V
V
OUT
1.2V
IN
2.7V TO 4.2V
V
SW
IN
C
**
IN
C
*
LTC3406B-1.2
RUN
OUT1
4.7µF
1
10µF
CER
CER
5
V
OUT
GND
3406B12 TA02
2
*MURATA GRM219R60JI06KE19B
**AVX06036D475MAT
†
FDK MIPW3226D2R2M
Efficiency vs Output Current
Load Step
Load Step
100
90
80
70
60
50
40
30
20
10
V
OUT
V
OUT
100mV/DIV
100mV/DIV
AC COUPLED
AC COUPLED
I
L
V
IN
= 2.7V
I
L
500mA/DIV
500mA/DIV
V
IN
= 4.2V
I
I
LOAD
LOAD
500mA/DIV
500mA/DIV
3406B12 TA04
3406B12 TA05
20µs/DIV
20µs/DIV
V
IN
= 3.6V
V
I
= 3.6V
LOAD
V
I
= 3.6V
LOAD
IN
IN
= 0mA TO 600mA
= 200mA TO 600mA
0.1
1
10
100
1000
OUTPUT CURRENT (mA)
3406B12 TA03
U
PACKAGE DESCRIPTIO
S5 Package
5-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1635)
0.62
MAX
0.95
REF
2.90 BSC
(NOTE 4)
1.22 REF
1.50 – 1.75
(NOTE 4)
2.80 BSC
1.4 MIN
3.85 MAX 2.62 REF
PIN ONE
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45 TYP
5 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.09 – 0.20
(NOTE 3)
0.20 BSC
DATUM ‘A’
0.01 – 0.10
1.00 MAX
0.30 – 0.50 REF
1.90 BSC
S5 TSOT-23 0302
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
sn3406b12 3406b12fs
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tationthattheinterconnectionofitscircuitsasdescribedhereinwillnotinfringeonexistingpatentrights.
11
LTC3406B-1.2
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
90% Efficiency, V = 3.6V to 25V, V
LT1616
500mA (I ), 1.4MHz, High Efficiency Step-Down
= 1.25V, I = 1.9mA,
Q
OUT
IN
OUT
DC/DC Converter
I
= <1µA, ThinSOT Package
SD
LT1676
450mA (I ), 100kHz, High Efficiency Step-Down
DC/DC Converter
90% Efficiency, V = 7.4V to 60V, V
= 1.24V, I = 3.2mA,
Q
OUT
IN
OUT
I
= 2.5µA, S8 Package
SD
LTC1701/LT1701B
LT1776
750mA (I ), 1MHz, High Efficiency Step-Down
DC/DC Converter
90% Efficiency, V = 2.5V to 5V, V
= 1.25V, I = 135µA,
OUT Q
OUT
IN
I
= <1µA, ThinSOT Package
SD
500mA (I ), 200kHz, High Efficiency Step-Down
90% Efficiency, V = 7.4V to 40V, V
= 1.24V, I = 3.2mA,
Q
OUT
IN
OUT
OUT
DC/DC Converter
I
= 30µA, N8, S8 Packages
SD
LTC1877
600mA (I ), 550kHz, Synchronous Step-Down
95% Efficiency, V = 2.7V to 10V, V
= 0.8V, I = 10µA,
Q
OUT
IN
DC/DC Converter
I
= <1µA, MS8 Package
SD
LTC1878
600mA (I ), 550kHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, V = 2.7V to 6V, V
= 0.8V, I = 10µA,
OUT Q
OUT
IN
I
= <1µA, MS8 Package
SD
LTC1879
1.2A (I ), 550kHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, V = 2.7V to 10V, V
= 0.8V, I = 15µA,
OUT
IN
OUT
Q
I
= <1µA, TSSOP-16 Package
SD
LTC3403
600mA (I ), 1.5MHz, Synchronous Step-Down
DC/DC Converter with Bypass Transistor
96% Efficiency, V = 2.5V to 5.5V, V = Dynamically Adjustable,
OUT
I = 20µA, I = <1µA, DFN Package
Q SD
OUT
IN
LTC3404
600mA (I ), 1.4MHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, V = 2.7V to 6V, V
= 0.8V, I = 10µA,
OUT Q
OUT
IN
I
= <1µA, MS8 Package
SD
LTC3405/LTC3405A
LTC3406
300mA (I ), 1.5MHz, Synchronous Step-Down
DC/DC Converter
96% Efficiency, V = 2.5V to 5.5V, V
= 0.8V, I = 20µA,
Q
OUT
IN
OUT
OUT
OUT
OUT
OUT
I
= <1µA, ThinSOT Package
SD
600mA (I ), 1.5MHz, Synchronous Step-Down
96% Efficiency, V = 2.5V to 5.5V, V
= 0.6V, I = 20µA,
Q
OUT
IN
DC/DC Converter
I
= <1µA, ThinSOT Package
SD
LTC3411
1.25A (I ), 4MHz, Synchronous Step-Down
95% Efficiency, V = 2.5V to 5.5V, V
= 0.8V, I = 60µA,
Q
OUT
IN
DC/DC Converter
I
= <1µA, MS Package
SD
LTC3412
2.5A (I ), 4MHz, Synchronous Step-Down
95% Efficiency, V = 2.5V to 5.5V, V
= 0.8V, I = 60µA,
Q
OUT
IN
DC/DC Converter
I
= <1µA, TSSOP-16E Package
SD
LTC3440
600mA (I ), 2MHz, Synchronous Buck-Boost
95% Efficiency, V = 2.5V to 5.5V, V
= 2.5V, I = 25µA,
Q
OUT
IN
DC/DC Converter
I
= <1µA, MS Package
SD
sn3406b12 3406b12fs
LT/TP 1004 1K • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
12
●
●
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
©LINEAR TECHNOLOGY CORPORATION 2004
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