LTC3606BIDDPBF [Linear]
800mA Synchronous Step-Down DC/DC with Average Input Current Limit; 800毫安同步降压型DC / DC与平均输入电流限制
| 型号： | LTC3606BIDDPBF |
| 厂家： | 
Linear |
| 描述： | 800mA Synchronous Step-Down DC/DC with Average Input Current Limit |
| 文件： | 总20页 (文件大小：314K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LTC3606B
800mA Synchronous
Step-Down DC/DC with
Average Input Current Limit
FEATURES
DESCRIPTION
The LTC®3606B is an 800mA monolithic synchronous
buck regulator using a constant frequency current mode
architecture.
n
Programmable Average Input Current Limit:
±±5 Accuracꢀ
n
Step-Down Output: Up to 965 Efficiencꢀ
n
Low Noise Pulse-Skipping Operation at Light Loads
The input supply voltage range is 2.5V to 5.5V, making it
ideal for Li-Ion and USB powered applications. 100% duty
cyclecapabilityprovideslowdropoutoperation,extending
the run time in battery-operated systems. Low output
voltages are supported with the 0.6V feedback reference
voltage. The LTC3606B can supply 800mA output current.
n
Input Voltage Range: 2.±V to ±.±V
n
Output Voltage Range: 0.6V to ±V
n
2.2±MHz Constant-Frequencꢀ Operation
n
Power Good Output Voltage Monitor
n
Low Dropout Operation: 1005 Dutꢀ Cꢀcle
n
Internal Soft-Start
n
The LTC3606B’s programmable average input current
limit is ideal for USB applications and for point-of-load
power supplies because the LTC3606B’s limited input
current will still allow its output to deliver high peak load
currentswithoutcollapsingtheinputsupply.Theoperating
frequency is internally set at 2.25MHz allowing the use of
smallsurfacemountinductors. Internalsoft-startreduces
in-rushcurrentduringstart-up. TheLTC3606Bisavailable
in an 8-Lead 3mm × 3mm DFN package.
Current Mode Operation for Excellent Line and Load
Transient Response
2% Output Voltage Accuracy
n
n
Short-Circuit Protected
Shutdown Current ≤ 1μA
Available in Small Thermally Enhanced 8-Lead
3mm × 3mm DFN Package
n
n
APPLICATIONS
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S.Patents, including 5481178, 6127815,
6304066, 6498466, 6580258, 6611131.
n
High Peak Load Current Applications
n
USB Powered Devices
n
Supercapacitor Charging
Radio Transmitters and Other Handheld Devices
n
TYPICAL APPLICATION
Monolithic Buck Regulator with Input Current Limit
1.5ꢀH
GSM Pulse Load
V
OUT
V
3.4V
V
OUT
IN
V
SW
3.4V AT
800mA
IN
TO 5.5V
200mV/DIV
LTC3606B
499k
RUN
C
V
IN
IN
+
2.2mF
10ꢀF
AC-COUPLED
1V/DIV
PGOOD
RLIM
s2
1210k
SuperCap
V
FB
PGOOD
I
OUT
GND
255k
500mA/DIV
116k
1000pF
3606B TA01
I
IN
500mA/DIV
I
= 475mA
3606B TA01b
LIM
1ms/DIV
V
LOAD
= 5V, 500mA COMPLIANT
IN
I
= 0A to 2.2A
3606bfa
1
LTC3606B
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
TOP VIEW
Input Supply Voltage (V )........................... –0.3V to 6V
IN
1
2
3
4
8
7
6
5
GND
RLIM
GND
SW
V
FB
V ...................................................–0.3V to V + 0.3V
FB
IN
RUN
9
GND
RUN, RLIM.......................................–0.3V to V + 0.3V
IN
PGOOD
SW...................................................–0.3V to V + 0.3V
IN
V
IN
PGOOD.............................................–0.3V to V + 0.3V
IN
P-Channel SW Source Current (DC) (Note 2)..............1A
N-Channel SW Source Current (DC) (Note 2) .............1A
Peak SW Source and Sink Current (Note 2)............. 2.7A
Operating Junction Temperature Range
DD PACKAGE
8-LEAD (3mm s 3mm) PLASTIC DFN
T
JMAX
= 125°C, θ = 40°C/W
JA
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
(Notes 3, 6, 8)........................................–40°C to 125°C
Storage Temperature Range .................. –65°C to 125°C
Reflow Peak Body Temperature ............................260°C
ORDER INFORMATION
LEAD FREE FINISH
LTC3606BEDD#PBF
LTC3606BIDD#PBF
TAPE AND REEL
PART MARKING*
LFMB
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3606BEDD#TRPBF
LTC3606BIDD#TRPBF
–40°C to 85°C
–40°C to 125°C
8-Lead (3mm × 3mm) Plastic DFN
8-Lead (3mm × 3mm) Plastic DFN
LFMB
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3606bfa
2
LTC3606B
ELECTRICAL CHARACTERISTICS The l denotes the specifications which applꢀ over the full operating
junction temperature range, otherwise specifications are at TA = 2±°C, VIN = ±V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
5.5
2.5
30
UNITS
l
l
l
V
V
V
Operating Voltage Range
Undervoltage Lockout
2.5
V
V
IN
IN
IN
V
V
Low to High
IN
2.1
UV
I
FB
Feedback Pin Input Current
Feedback Voltage
nA
l
l
V
LTC3606BE, –40°C < T < 85°C (Note 7)
LTC3606BI, –40°C < T < 125°C (Note 7)
0.588
0.582
0.600
0.600
0.612
0.618
V
V
FBREG
J
J
ΔV
ΔV
V
V
Line Regulation
Load Regulation
V
= 2.5V to 5.5V (Note 7)
0.01
0.5
0.25
%/V
%
LINEREG
FB
IN
I
= 0mA to 800mA (Note 7)
LOADREG
FB
LOAD
I
S
Supply Current
Active Mode (Note 4)
Shutdown
V
V
= 0.95 × V
420
650
1
ꢀA
ꢀA
FB
FBREG
= 0V, V = 5.5V
RUN
IN
l
f
I
I
Oscillator Frequency
V
V
= V
1.8
2.25
2.7
MHz
mA
OSC
FB
IN
FBREG
Peak Switch Current Limit
Input Average Current Limit
= 5V, V < V , Duty Cycle <35%
FBREG
1800
2400
LIM(PEAK)
INLIM
FB
RLIM = 116k
RLIM = 116k, LTC3606BE
RLIM = 116k, LTC3606BI
450
437
427
475
475
475
500
513
523
mA
mA
mA
l
l
R
Main Switch On-Resistance (Note 5)
Synchronous Switch On-Resistance (Note 5)
V
V
= 5V, I = 100mA
0.27
0.25
Ω
Ω
DS(ON)
IN
IN
SW
= 5V, I = 100mA
SW
I
t
Switch Leakage Current
Soft-Start Time
V
V
= 5V, V = 0V
RUN
0.01
0.95
1
1
ꢀA
ms
V
SW(LKG)
IN
from 0.06V to 0.54V
0.3
0.4
1.3
1.2
1
SOFTSTART
FB
l
l
V
RUN Threshold High
RUN Leakage Current
Power Good Threshold
RUN
RUN
I
0V ≤ V
≤ 5V
0.01
ꢀA
RUN
PGOOD
Entering Window
V
FB
V
FB
Ramping Up
Ramping Down
–5
5
–7
7
%
%
Leaving Window
V
FB
V
FB
Ramping Up
Ramping Down
9
–9
11
–11
%
%
PGOOD Blanking Power Good Blanking Time
PGOOD Rising and Falling, V = 5V
90
15
ꢀs
Ω
IN
R
Power Good Pull-Down On-Resistance
PGOOD Leakage Current
8
30
1
PGOOD
PGOOD
I
V
= 5V
ꢀA
PGOOD
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note ±: The switch on-resistance is guaranteed by correlation to wafer
level measurements.
Note 6: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
Note 7: The converter is tested in a proprietary test mode that connects
the output of the error amplifier to the SW pin, which is connected to an
external servo loop.
Note 2: Guaranteed by long term current density limitations.
Note 3: The LTC3606BE is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
junction temperature range are assured by design, characterization and
correlation with statistical process controls. The LTC3606BI is guaranteed
to meet specified performance over the full –40°C to 125°C operating
junction temperature range.
Note 8: T is calculated from the ambient temperature T and the power
J
A
Note 4: Dynamic supply current is higher due to the internal gate charge
dissipation as follows: T = T + (P )(θ °C/W)
J A D JA
being delivered at the switching frequency.
3606bfa
3
LTC3606B
TA = 2±°C, VIN = ±V, unless otherwise noted.
TYPICAL PERFORMANCE CHARACTERISTICS
Pulse-Skipping Mode Operation
Supplꢀ Current vs Temperature
Efficiencꢀ vs Input Voltage
100
550
500
450
400
350
300
250
200
RUN = V
LOAD
IN
SW
2V/DIV
90
80
70
60
50
I
= 0A
V
= 5.5V
IN
V
OUT
50mV/DIV
AC-
V
= 2.7V
IN
COUPLED
I
I
I
= 100mA
= 400mA
= 800mA
I
I
I
= 10mA
= 1mA
= 0.1mA
40
30
20
10
0
OUT
OUT
OUT
OUT
OUT
OUT
I
L
100mA/DIV
3606B G01
5ꢀs/DIV
V
= 3.3V
4
OUT
V
V
LOAD
= 5V
IN
= 3.3V
= 5mA
OUT
I
3.5
4.5
(V)
5
5.5
–50 –25
0
25
50
75 100 125
V
TEMPERATURE (°C)
IN
3606B G03
3606B G02
Oscillator Frequencꢀ
vs Temperature
Switch Leakage vs Input Voltage
Regulated Voltage vs Temperature
1.5
1.0
2.5
2.4
2.3
2.2
2.1
2.0
1.9
1.8
1000
800
600
400
200
0
0.5
MAIN SWITCH
0
–0.5
–1.0
–1.5
V
V
V
V
= 2.7V
SYNCHRONOUS SWITCH
IN
IN
IN
IN
= 3.6V
= 4.2V
= 5V
–50 –25
0
25
50
75 100 125
–50 –25
0
25
50
75 100 125
2.5
3
3.5
4
4.5
5
5.5
TEMPERATURE (°C)
TEMPERATURE (°C)
V
(V)
IN
3606B G04
3606B G05
3606B G06
Switch On-Resistance
vs Input Voltage
Switch On-Resistance
vs Temperature
Efficiencꢀ vs Load Current
0.5
0.4
0.3
0.2
0.1
0
0.7
0.6
0.5
0.4
0.3
0.2
0.1
600
500
400
300
200
100
V
V
V
= 2.7V
= 3.6V
= 5V
V
= 3.3V
IN
IN
IN
OUT
MAIN SWITCH
90
80
70
60
50
40
30
20
10
0
MAIN SWITCH
V
V
V
= 3.6V
= 4.2V
= 5V
IN
IN
IN
SYNCHRONOUS SWITCH
SYNCHRONOUS SWITCH
25 50
–0.1
–50 –25
0
75 100 125
2.5
3
3.5
4
4.5
5
5.5
0.0001
0.001
0.01
0.1
1
TEMPERATURE (°C)
V
(V)
OUTPUT CURRENT (A)
IN
3606B G09
3606B G07
3606B G11
3606bfa
4
LTC3606B
TA = 2±°C, VIN = ±V, unless otherwise noted.
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiencꢀ vs Load Current
Load Regulation
Line Regulation
0.6
0.4
100
90
80
70
60
50
40
30
20
10
0
3.0
2.5
2.0
1.5
1.0
0.5
0
V
LOAD
= 1.8V
= 100mA
V
= 1.2V
OUT
OUT
I
0.2
0
–0.2
–0.4
–0.6
V
V
V
V
= 2.7V
= 3.6V
= 4.2V
= 5V
IN
IN
IN
IN
V
V
V
= 1.8V
= 2.5V
= 3.3V
OUT
OUT
OUT
–0.5
–1.0
2.5
3.0
3.5
4.0
(V)
4.5
5.0
5.5
0.0001
0.001
0.01
0.1
1
0
100 200 300 400 500 600 700 800
V
OUTPUT CURRENT (A)
LOAD CURRENT (mA)
IN
3606B G16
3606B G13
3606B G15
VRLIM vs Input Current
Start-Up from Shutdown
Start-Up from Shutdown
1.2
1.0
0.8
0.6
0.4
0.2
0
I
= 475mA
LIM
LIM
RUN
2V/DIV
RUN
R
= 116k
2V/DIV
V
OUT
2V/DIV
V
OUT
R
LIM
1V/DIV
1V/DIV
I
L
I
IN
250mA/DIV
500mA/DIV
3606B G17
3606B G18
200ꢀs/DIV
2ms/DIV
V
R
C
= 5V, V
= 3.3V
OUT
V
R
C
= 5V, V
= 3.4V
OUT
IN
IN
L
= 7ꢁ
= NO LOAD, C = 4.4mF
L
LOAD
0
100
200
300
(mA)
400
500
600
= 4.7ꢀF
= 2200pF, I
= 500mA
LIM
LOAD
LIM
I
IN
3606B G18b
Average Input Current Limit
vs Temperature
Load Step
Load Step
8
6
V
IN
= 5V
= 475mA
V
OUT
I
LIM
V
OUT
200mV/DIV
AC-COUPLED
200mV/DIV
4
AC-COUPLED
2
I
L
I
L
1A/DIV
0
1A/DIV
–2
–4
–6
–8
I
LOAD
I
LOAD
1A/DIV
1A/DIV
3606B G20
3606B G21
20ꢀs/DIV
= 3.3V
20ꢀs/DIV
= 1.8V
V
I
OUT
= 5V, V
OUT
LOAD
V
I
= 5V, V
OUT
LOAD
OUT
IN
IN
= 0A TO 800mA
= 80mA TO 800mA
–50
125
–25
0
25
50
75 100
C
= 100ꢀF, C = 20pF
C
= 100ꢀF, C = 20pF
F
F
TEMPERATURE (°C)
3606B G19
3606bfa
5
LTC3606B
PIN FUNCTIONS
GND (Pins 1, 3, Exposed Pad Pin 9): Ground. Connect to
PGOOD(Pin6):Open-DrainLogicOutput.PGOODispulled
the (–) terminal of C , and the (–) terminal of C . The
to ground if the voltage on the V pin is not within power
OUT
IN
FB
Exposed Pad must be soldered to PCB.
good threshold.
RLIM (Pin 2): Average Input Current Limit Program Pin.
Place a resistor and capacitor in parallel from this pin to
ground.
RUN (Pin 7): Regulator Enable. Forcing this pin to V
enablesregulator, whileforcingittoGNDcausesregulator
to shut down.
IN
SW (Pin 4): Regulator Switch Node Connection to the
V
(Pin 8): Regulator Output Feedback. Receives the
FB
Inductor. This pin swings from V to GND.
IN
feedback voltage from the external resistive divider
across the regulator output. Nominal voltage for this pin
is 0.6V.
V
(Pin ±): Main Power Supply. Must be closely de-
IN
coupled to GND.
FUNCTIONAL DIAGRAM
2
RLIM
+
–
0.6V REF
OSC
7
RUN
1V
OSC
MIN
CLAMP
5
V
IN
SLOPE
COMP
–
–
+
I
TH
V
FB
8
EA
SLEEP
–
–
+
I
0.6V
COMP
V
+
SLEEP
S
R
Q
RS
LATCH
SOFT-START
+
I
Q
SWITCHING
LOGIC
AND
BLANKING
CIRCUIT
COMP
–
–
+
ANTI
SHOOT-
THRU
0.654V
4
9
SW
V
FB
–
+
PGOOD
6
0.546V
+
–
I
RCMP
GND
SHUTDOWN
3606B FD
3606bfa
6
LTC3606B
OPERATION
The LTC3606B uses a constant-frequency, current mode
architecture. The operating frequency is set at 2.25MHz.
at constant-frequency down to very low currents, where it
will begin skipping pulses to maintain output regulation.
This mode of operation exhibits low output ripple as well
as low audio noise and reduced RF interference while
providing reasonable low current efficiency.
The output voltage is set by an external resistor divider
returned to the V pins. An error amplifier compares the
FB
dividedoutputvoltagewithareferencevoltageof0.6Vand
regulates the peak inductor current accordingly.
Dropout Operation
The LTC3606B continuously monitors the input current
via the voltage drop across the R
When the input supply voltage decreases toward the
output voltage the duty cycle increases to 100%, which
is the dropout condition. In dropout, the PMOS switch is
turnedoncontinuouslywiththeoutputvoltagebeingequal
to the input voltage minus the voltage drops across the
internal P-channel MOSFET and the inductor.
of the internal
DS(ON)
P-channel MOSFET. When the input current exceeds the
programmedinputcurrentlimitsetbyanexternalresistor,
R
LIM
, theregulator’sinputcurrentislimited. Theregulator
output voltage will drop to meet output current demand
and to maintain constant input current.
An important design consideration is that the R
DS(ON)
Main Control Loop
of the P-channel switch increases with decreasing input
supplyvoltage(seetheTypicalPerformanceCharacteristics
section). Therefore, the user should calculate the worst-
case power dissipation when the LTC3606B is used at
100% duty cycle with low input voltage (see Thermal
Considerations in the Applications Information section).
Duringnormaloperation,thetoppowerswitch(P-channel
MOSFET) is turned on at the beginning of a clock cycle
when the V voltage is below the reference voltage. The
FB
current into the inductor and the load increases until the
peak inductor current (controlled by I ) is reached. The
TH
RS latch turns off the synchronous switch and energy
stored in the inductor is discharged through the bottom
switch (N-channel MOSFET) into the load until the next
clock cycle begins, or until the inductor current begins to
Soft-Start
Inordertominimizetheinrushcurrentontheinputbypass
capacitor, the LTC3606B slowly ramps up the output
voltage during start-up. Whenever the RUN pin is pulled
high, the corresponding output will ramp from zero to
full-scale over a time period of approximately 750ꢀs. This
prevents the LTC3606B from having to quickly charge the
output capacitor and thus supplying an excessive amount
of instantaneous current.
reverse (sensed by the I
comparator).
RCMP
The peak inductor current is controlled by the internally
compensated I voltage, which is the output of the error
TH
amplifier.ThisamplifierregulatestheV pintotheinternal
FB
0.6V reference by adjusting the peak inductor current
accordingly.
When the output is loaded heavily, for example, with
millifarad of capacitance, it may take longer than 750ꢀs to
charge the output to regulation. If the output is still low
after the soft-start time, the LTC3606B will try to quickly
charge the output capacitor. In this case, the input current
limit (after it engages) can prevent excessive amount of
instantaneous current that is required to quickly charge
the output. See the Start-Up from Shutdown curve
When the input current limit is engaged, the peak inductor
current will be lowered, which then reduces the switching
duty cycle and V . This allows the input voltage to stay
OUT
regulated when its programmed current output capability
is met.
Light Load Operation
TheLTC3606Bwillautomaticallytransitionfromcontinuous
operation to the pulse-skipping operation when the load
current is low. The inductor current is not fixed during the
pulse-skippingmodewhichallowstheLTC3606Btoswitch
(C = 4.4mF)in the Typical Performance Characteristics
L
section. After input current limit is engaged, the output
slowly ramps up to regulation while limited by its 500mA
of input current.
3606bfa
7
LTC3606B
OPERATION
Short-Circuit Protection
Programming Input Current Limit
When either regulator output is shorted to ground, the
corresponding internal N-channel switch is forced on for
a longer time period for each cycle in order to allow the
inductor to discharge, thus preventing inductor current
runaway. This technique has the effect of decreasing
switching frequency. Once the short is removed, normal
operation resumes and the regulator output will return to
its nominal voltage.
SelectionofoneexternalR resistorwillprogramtheinput
LIM
current limit. The current limit can be programmed from
200mA up to I
LIM
current. As the input current increases,
PEAK
R
voltage will follow. When R
reaches the internal
LIM
comparator threshold of 1V, the power PFET on-time will
be shortened, thereby, limiting the input current.
Use the following equation to select the R
resistance
LIM
that corresponds to the input current limit.
Input Current Limit
R
LIM
= 55k / I
DC
Internal current sense circuitry measures the inductor
current through the voltage drop across the power PFET
switchandforcesthesamevoltageacrossthesmallsense
PFET. The voltage across the small sense PFET generates
a current representing 1/55,000th of the inductor current
during the on-cycle. The current out of RLIM pin is the
representation of the inductor current, which can be
expressed in the following equation.
I
istheinputcurrent(atV )tobelimited.Thefollowingare
DC IN
some R
values with the corresponding current limit.
LIM
R
LIM
I
DC
91.6k
110k
600mA
500mA
400mA
137.5k
Selection of C Capacitance
LIM
I
= I
• D1 • K1
OUT
RLIM
Since I
current is a function of the inductor current,
RLIM
where D1 = V /V is the duty cycle.
OUT1 IN
its dependency on the duty cycle cannot be ignored. Thus,
a C capacitor is needed to integrate the I current
LIM
RLIM
K1istheratioR
(powerPFET)/R (sensePFET).
DS(ON)
DS(ON)
andsmoothouttransientcurrents.TheLTC3606Bisstable
with any size capacitance >100pF at the RLIM pin.
The ratio of the power PFET to the sense PFET is trimmed
to within 2%.
Each application input current limit will call for different
Given that both PFETs are carefully laid out and matched,
their temperature and voltage coefficient effects will be
similar and their terms be canceled out in the equation. In
that case, the constant K1 will only be dependent on area
C
valuetooptimizeitsresponsetime.UsingalargeC
LIM
LIM
capacitor requires longer time for the RLIM pin voltage to
charge.Forexample,considertheapplication500mAinput
currentlimit,5Vinputand1A,2.5Voutputwitha50%duty
cycle. When an instantaneous 1A output pulse is applied,
the current out of the RLIM pin becomes 1A/55k = 18.2ꢀA
during the 50% on-time or 9.1ꢀA full duty cycle. With a
scaling, which is trimmed to within 2%. Thus, the I
RLIM
current will track the input current very well over varying
temperature and V .
IN
The RLIM pin can be grounded to disable input current
limit function.
C
capacitor of 1ꢀF, R of 116k, and using I = CdV/dt,
LIM
LIM
it will take 110ms for C to charge from 0V to 1V. This is
LIM
the time after which the LTC3606B will start input current
limiting. Any current within this time must be considered
in each application to determine if it is tolerable.
3606bfa
8
LTC3606B
OPERATION
Figure 1a shows V (I ) current below input current
and the output must deliver the required current load.
This may cause the input voltage to droop if the current
compliance is exceeded. Depending on how long this time
IN IN
limit with a C capacitor of 0.1ꢀF. When the load pulse
LIM
is applied, under the specified condition, I
current is
LIM
1.1A/55k • 0.66 = 13.2ꢀA, where 0.66 is the duty cycle.
It will take a little more than 7.5ms to charge the C
is, the V supply decoupling capacitor can provide some
IN
ofthiscurrentbeforeV droopstoomuch. Inapplications
LIM
IN
capacitor from 0V to 1V, after which the LTC3606B begins
to limit input current. The I current is not limited during
with a bigger V supply decoupling capacitor and where
IN
V supply is allow to droop closer to dropout, the C
IN LIM
IN
this 7.5ms time and is more than 725mA. This current
transient may cause the input supply to temporarily
droop if the supply current compliance is exceeded, but
recovers after the input current limit engages. The output
will continue to deliver the required current load while the
output voltage droops to allow the input voltage to remain
regulated during input current limit.
capacitor can be increased slightly. This will delay the
start of input current limit and artificially regulated V
OUT
before input current limit is engaged. In this case, within
the 577ꢀs load pulse, the V voltage will stay artificially
OUT
regulated for 92ꢀs out of the total 577ꢀs before the input
current limit activates. This approach may be used if a
faster recovery on the output is desired.
For applications with short load pulse duration, a smaller
LIM
Selecting a very small C
will speed up response time
LIM
C
capacitor may be the better choice as in the example
but it can put the device within threshold of interfering
with normal operation and input current limit in every
few switching cycles. This may be undesirable in terms
of noise. Use 2πRC >> 100/clock frequency (2.25MHz) as
shown in Figure 1b. In this example, a 577ꢀs, 0A to 2A
outputpulseisappliedonceevery4.7ms. AC capacitor
LIM
of 2.2nF requires 92ꢀs for V
to charge from 0V to 1V.
During this 92ꢀs, the input current limit is not yet engaged
RLIM
a starting point, R being R , C being C
.
LIM
LIM
V
V
OUT
OUT
200mV/DIV
2V/DIV
V
IN
I
IN
AC-COUPLED
1V/DIV
500mA/DIV
V
RLIM
1V/DIV
I
OUT
500mA/DIV
I
I
IN
L
500mA/DIV
1A/DIV
3606B F01b
3606B F01a
1ms/DIV
50ms/DIV
V
= 5V, 500mA COMPLIANT
V
= 5V, 500mA COMPLIANT
IN
IN
R
I
= 116k, C
= 2200pF
R
I
= 116k, C
= 0.1ꢀF
LIM
LIM
= 0A to 2A, C
LIM
LIM
= 2.2mF, V
= 3.3V
OUT
= 0A to 1.1A, C
= 2.2mF, V = 3.3V
OUT
LOAD
LIM
OUT
LOAD
LIM
OUT
I
= 475mA
I
= 475mA
Figure 1a. Input Current Limit Within 100ms Load Pulses
Figure 1b. Input Current Limit Within
±77μs, 2A Repeating Load Pulses
3606bfa
9
LTC3606B
APPLICATIONS INFORMATION
AgeneralLTC3606BapplicationcircuitisshowninFigure2.
Externalcomponentselectionisdrivenbytheloadrequire-
ment, andbeginswiththeselectionoftheinductorL. Once
Inductor Core Selection
Different core materials and shapes will change the size/
currentandprice/currentrelationshipofaninductor.Toroid
or shielded pot cores in ferrite or permalloy materials are
small and do not radiate much energy, but generally cost
more than powdered iron core inductors with similar
electrical characteristics. The choice of which style
inductor to use often depends more on the price versus
sizerequirements,andanyradiatedfield/EMIrequirements,
than on what the LTC3606B requires to operate. Table 1
shows some typical surface mount inductors that work
well in LTC3606B applications.
the inductor is chosen, C and C
can be selected.
IN
OUT
Inductor Selection
Although the inductor does not influence the operat-
ing frequency, the inductor value has a direct effect on
ripple current. The inductor ripple current ΔI decreases
L
with higher inductance and increases with higher V or
IN
V
:
OUT
ꢂ
ꢅ
VOUT
fO •L
V
V
ꢄ
OUT ꢇ
IN
ꢀIL =
• 1ꢁ
(1)
ꢃ
ꢆ
Table 1. Representative Surface Mount Inductors
MANU-
MAX DC
Accepting larger values of ΔI allows the use of low
FACTURER PART NUMBER VALUE CURRENT DCR
HEIGHT
L
inductances, but results in higher output voltage ripple,
greater core losses, and lower output current capability.
A reasonable starting point for setting ripple current is
40%ofthemaximumoutputloadcurrent.So,fora800mA
Coilcraft
LPS4012-152ML 1.5ꢀH 2200mA 0.070Ω 1.2mm
LPS4012-222ML 2.2ꢀH 1750mA 0.100Ω 1.2mm
LPS4012-332ML 3.3ꢀH 1450mA 0.100Ω 1.2mm
LPS4012-472ML 4.7ꢀH 1450mA 0.170Ω 1.2mm
LPS4018-222ML 2.2ꢀH 2300mA 0.070Ω 1.8mm
LPS4018-332ML 3.3ꢀH 2000mA 0.080Ω 1.8mm
LPS4018-472ML 4.7ꢀH 1800mA 0.125Ω 1.8mm
regulator, ΔI = 320mA (40% of 800mA).
L
The inductor value will also have an effect on Burst Mode
operation. The transition to low current operation begins
when the peak inductor current falls below a level set by
the internal burst clamp. Lower inductor values result in
higher ripple current which causes the transition to occur
at lower load currents. This causes a dip in efficiency in
the upper range of low current operation. Furthermore,
lower inductance values will cause the bursts to occur
with increased frequency.
FDK
FDKMIPF2520D
FDKMIPF2520D
FDKMIPF2520D
4.7ꢀH 1100mA 0.11Ω
3.3ꢀH 1200mA 0.1Ω
2.2ꢀH 1300mA 0.08Ω
1mm
1mm
1mm
Murata
LQH32CN4R7M23 4.7ꢀH 450mA
4.7ꢀH 950mA
0.2Ω
0.2Ω
2mm
1.2mm
2mm
Panasonic ELT5KT4R7M
Sumida CDRH2D18/LD
4.7ꢀH 630mA 0.086Ω
CDH38D11SNP- 3.3μH 1560mA 0.115Ω 1.2mm
3R3M
CDH38D11SNP- 2.2μH 1900mA 0.082Ω 1.2mm
2R2M
Taiyo Yuden CB2016T2R2M
CB2012T2R2M
2.2ꢀH 510mA
2.2ꢀH 530mA
3.3ꢀH 410mA
2.2ꢀH 1100mA
4.7ꢀH 750mA
0.13Ω 1.6mm
0.33Ω 1.25mm
0.27Ω 1.6mm
L1
V
IN
V
V
SW
OUT
IN
CB2016T3R3M
2.5V TO 5.5V
LTC3606B
C
NR30102R2M
0.1Ω
1mm
1mm
F
R
PGD
RUN
NR30104R7M
0.19Ω
C
C
OUT
IN
PGOOD
RLIM
TDK
VLF3010AT4R7- 4.7ꢀH 700mA
0.28Ω
1mm
1mm
1mm
R2
V
MR70
FB
PGOOD
VLF3010AT3R3- 3.3ꢀH 870mA
MR87
0.17Ω
GND
R1
VLF3010AT2R2- 2.2ꢀH 1000mA 0.12Ω
M1R0
R
C
LIM
LIM
3606B F02
VLF4012AT-2R2 2.2ꢀH 1500mA 0.076Ω 1.2mm
M1R5
Figure 2. LTC3606B General Schematic
VLF5012ST-3R3 3.3μH 1700mA 0.095Ω 1.2mm
M1R7
VLF5014ST-2R2 2.2ꢀH 2300mA 0.059Ω 1.4mm
M2R3
3606bfa
10
LTC3606B
APPLICATIONS INFORMATION
Input Capacitor (C ) Selection
voltage, the output ripple is highest at maximum input
IN
voltage since ΔI increases with input voltage.
L
In continuous mode, the input current of the converter is a
square wave with a duty cycle of approximately V /V .
If tantalum capacitors are used, it is critical that the
capacitors are surge tested for use in switching power
supplies. An excellent choice is the AVX TPS series of
surface mount tantalum. These are specially constructed
and tested for low ESR so they give the lowest ESR for a
givenvolume.OthercapacitortypesincludeSanyoPOSCAP,
Kemet T510 and T495 series, and Sprague 593D and
595D series. Consult the manufacturer for other specific
recommendations.
OUT IN
Topreventlargevoltagetransients, alowequivalentseries
resistance (ESR) input capacitor sized for the maximum
RMS current must be used. The maximum RMS capacitor
current is given by:
VOUT(V ꢁ VOUT
)
IN
IRMS ꢀIMAX
V
IN
Where the maximum average output current I
equals
MAX
the peak current minus half the peak-to-peak ripple cur-
rent, I = I – ΔI /2. This formula has a maximum at
Using Ceramic Input and Output Capacitors
MAX LIM
L
V = 2V , where I = I /2. This simple worst-case
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them
ideal for switching regulator applications. Because the
LTC3606B control loop does not depend on the output
capacitor’s ESR for stable operation, ceramic capacitors
can be used freely to achieve very low output ripple and
small circuit size.
IN
OUT
RMS OUT
is commonly used to design because even significant
deviations do not offer much relief. Note that capacitor
manufacturer’s ripple current ratings are often based on
only2000hourslifetime.Thismakesitadvisabletofurther
deratethecapacitor,orchooseacapacitorratedatahigher
temperaturethanrequired.Severalcapacitorsmayalsobe
paralleled to meet the size or height requirements of the
design. An additional 0.1ꢀF to 1ꢀF ceramic capacitor is
However, care must be taken when ceramic capacitors are
used at the input. When a ceramic capacitor is used at the
input and the power is supplied by a wall adapter through
long wires, a load step at the output can induce ringing at
also recommended on V for high frequency decoupling
IN
when not using an all-ceramic capacitor solution.
Output Capacitor (C ) Selection
OUT
theinput, V . Atbest, thisringingcancoupletotheoutput
IN
The selection of C
is driven by the required effective
and be mistaken as loop instability. At worst, a sudden
OUT
series resistance (ESR). Typically, once the ESR require-
inrush of current through the long wires can potentially
ment for C
has been met, the RMS current rating
cause a voltage spike at V , large enough to damage the
OUT
IN
generally far exceeds the I
requirement. The
part. For more information, see Application Note 88.
RIPPLE(P-P)
is determined by:
output ripple ΔV
OUT
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage
characteristics of all the ceramics for a given value and
size.
ꢂ
ꢅ
ꢇ
ꢆ
1
ꢀVOUT ꢁ ꢀI ESR+
L ꢄ
8fOCOUT
ꢃ
wheref =operatingfrequency,C
=outputcapacitance
O
OUT
and ΔI = ripple current in the inductor. For a fixed output
L
3606bfa
11
LTC3606B
APPLICATIONS INFORMATION
Setting the Output Voltage
The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance. For a detailed
explanationofoptimizingthecompensationcomponents,
including a review of control loop theory, refer to
Application Note 76.
The LTC3606B regulates the V pin to 0.6V during
FB
regulation. Thus, the output voltage is set by a resistive
divider, Figure 2, according to the following formula:
R2
R1
VOUT =0.6V 1+
(2)
Insomeapplications,amoreseveretransientcanbecaused
byswitchinginloadswithlarge(>1ꢀF)inputcapacitors.The
discharged input capacitors are effectively put in parallel
Keeping the current small (<10ꢀA) in these resistors
maximizes efficiency, but making it too small may allow
stray capacitance to cause noise problems or reduce the
phase margin of the error amp loop.
with C , causing a rapid drop in V . No regulator can
OUT
OUT
deliverenoughcurrenttopreventthisproblemiftheswitch
connectingtheloadhaslowresistanceandisdrivenquickly.
Thesolutionistolimittheturn-onspeedoftheloadswitch
driver. A Hot Swap™ controller is designed specifically for
this purpose and usually incorporates current limiting,
short-circuit protection, and soft-starting.
To improve the frequency response of the main control
loop, a feedback capacitor (C ) may also be used. Great
F
care should be taken to route the V line away from noise
FB
sources, such as the inductor or the SW line.
Checking Transient Response
Efficiencꢀ Considerations
The regulator loop response can be checked by looking
at the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
a load step occurs, V
immediately shifts by an amount
OUT
equal to ΔI
• ESR, where ESR is the effective series
LOAD
resistance of C . ΔI
also begins to charge or dis-
OUT
LOAD
chargeC generatingafeedbackerrorsignalusedbythe
OUT
% Efficiency = 100% – (L1 + L2 + L3 + ...)
regulator to return V
to its steady-state value. During
can be monitored for overshoot
OUT
this recovery time, V
whereL1,L2,etc.,aretheindividuallossesasapercentage
of input power.
OUT
or ringing that would indicate a stability problem.
The initial output voltage step may not be within the
bandwidth of the feedback loop, so the standard second
order overshoot/DC ratio cannot be used to determine
Although all dissipative elements in the circuit produce
losses, four sources usually account for the losses in
LTC3606B circuits: 1) V quiescent current, 2) switching
IN
2
the phase margin. In addition, feedback capacitors (C )
losses, 3) I R losses, 4) other system losses.
F
can be added to improve the high frequency response, as
1. The V current is the DC supply current given in the
IN
shown in Figure 2. Capacitor C provides phase lead by
F
Electrical Characteristics which excludes MOSFET
creating a high frequency zero with R2 which improves
driver and control currents. V current results in a
IN
the phase margin.
small (<0.1%) loss that increases with V , even at
IN
no load.
3606bfa
12
LTC3606B
APPLICATIONS INFORMATION
2. The switching current is the sum of the MOSFET driver
andcontrolcurrents.TheMOSFETdrivercurrentresults
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
Thermal Considerations
In a majority of applications, the LTC3606B does not
dissipate much heat due to its high efficiency. In the
unlikely event that the junction temperature somehow
reachesapproximately150°C,bothpowerswitcheswillbe
turned off and the SW node will become high impedance.
The goal of the following thermal analysis is to determine
whetherthepowerdissipatedcausesenoughtemperature
risetoexceedthemaximumjunctiontemperature(125°C)
of the part. The temperature rise is given by:
from V to ground. The resulting dQ/dt is a current
IN
out of V that is typically much larger than the DC bias
IN
current. In continuous mode, I
= f (Q + Q ),
GATECHG
O T B
where Q and Q are the gate charges of the internal
T
B
top and bottom MOSFET switches. The gate charge
losses are proportional to V and thus their effects
IN
T
= P • θ
D JA
will be more pronounced at higher supply voltages.
RISE
2
where P is the power dissipated by the regulator and θ
3. I R losses are calculated from the DC resistances of
D
JA
is the thermal resistance from the junction of the die to
the internal switches, R , and external inductor, R .
SW
L
the ambient temperature. The junction temperature, T ,
is given by:
In continuous mode, the average output current flows
throughinductorL,butis“chopped”betweentheinternal
top and bottom switches. Thus, the series resistance
looking into the SW pin is a function of both top and
J
T = T
J
+ T
AMBIENT
RISE
As a worst-case example, consider the case when the
LTC3606B is in dropout at an input voltage of 2.7V with
a load current of 800mA and an ambient temperature of
70°C.FromtheTypicalPerformanceCharacteristicsgraph
bottom MOSFET R
follows:
and the duty cycle (DC) as
DS(ON)
R
SW
= (R ) • (DC) + (R ) • (1– DC)
DS(ON)TOP DS(ON)BOT
of Switch Resistance, the R
of the switch is 0.33Ω.
TheR
forboththetopandbottomMOSFETscanbe
DS(ON)
DS(ON)
Therefore, the power dissipated is:
obtained from the Typical Performance Characteristics
2
2
curves. Thus, to obtain I R losses:
P = I
• R
= 212mV
D
OUT
DS(ON)
2
2
I R losses = I
• (R + R )
SW L
OUT
Given that the thermal resistance of a properly soldered
DFN package is approximately 40°C/W, the junction
temperature of an LTC3606B device operating in a 70°C
ambient temperature is approximately:
4. Other “hidden” losses, such as copper trace and
internal battery resistances, can account for additional
efficiency degradations in portable systems. It is very
important to include these “system” level losses in
the design of a system. The internal battery and fuse
resistancelossescanbeminimizedbymakingsurethat
T = (0.212W • 40°C/W) + 70°C = 78.5°C
J
which is well below the absolute maximum junction
temperature of 125°C.
C has adequate charge storage and very low ESR at
IN
the switching frequency. Other losses, including diode
conduction losses during dead-time, and inductor
core losses, generally account for less than 2% total
additional loss.
3606bfa
13
LTC3606B
APPLICATIONS INFORMATION
PC Board Laꢀout Considerations
should be routed away from noisy components and
traces, such as the SW line (Pin 4), and their trace
length should be minimized.
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3606B. These items are also illustrated graphically
in the layout diagrams of Figures 3a and 3b. Check the
following in your layout:
4. Keep sensitive components away from the SW pin, if
possible.TheinputcapacitorC ,C andtheresistors
IN LIM
R1, R2, and R should be routed away from the SW
LIM
traces and the inductors.
1. Does the capacitor C connect to the power V (Pin 5)
IN
IN
5. A ground plane is preferred, but if not available, keep
the signal and power grounds segregated with small
signal components returning to the GND pin at a single
point. These ground traces should not share the high
and GND (Pin 9) as closely as possible? This capacitor
provides the AC current of the internal power MOSFETs
and their drivers.
2. Are the respective C
and L closely connected? The
OUT
current path of C or C
.
IN
OUT
(–) plate of C
returns current to GND and the (–)
OUT
6. Flood all unused areas on all layers with copper.
Flooding with copper will reduce the temperature rise
of power components. These copper areas should be
plate of C .
IN
3. The resistor divider, R1 and R2, must be connected
between the (+) plate of C and a ground sense line
OUT
connected to V or GND.
IN
terminated near GND (Pin 9). The feedback signal V
FB
L1
V
IN
V
V
IN
SW
OUT
2.5V TO 5.5V
C
F
LTC3606B
C
IN
R
PGD
RUN
PGOOD
V
FB
R2
RLIM
C
R1
OUT
GND
C
R
LIM
LIM
3606B F03a
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 3a. LTC3606B Laꢀout Diagram (See Board Laꢀout Checklist)
3606bfa
14
LTC3606B
APPLICATIONS INFORMATION
VIA TO
V SENSE
GND
OUT
V
IN
V
GND
RLIM
GND
SW
FB
RUN
PGOOD
V
IN
SW
GND
V
OUT
Figure 3b. LTC3606B Suggested Laꢀout
3606bfa
15
LTC3606B
APPLICATIONS INFORMATION
Design Example
of C = 10ꢀF should suffice, if the source impedance is
IN
very low.
As a design example, consider using the LTC3606B in a
The feedback resistors program the output voltage. To
maintain high efficiency at light loads, the current in these
resistors should be kept small. Choosing 10ꢀA with the
0.6V feedback voltage makes R1~60k. A close standard
1% resistor is 59k. Using Equation (2).
USB-GSM application, with V = 5V, I
= 500mA,
IN
INMAX
with the output charging a SuperCap of 4.4mF. The load
requires800mAinactivemodeand1mAinstandbymode.
The output voltage V
= 3.4V.
OUT
First, calculate the inductor value for about 40% ripple
VOUT
0.6
current (320mA in this example) at maximum V . Using
IN
R2=
1 •R1=276k, 280k for 1%
a derivation of Equation (1):
3.4V
3.4V
5V
A feedforward capacitor is not used since the 4.4mF
SuperCap will inhibit any fast output voltage transients.
Figure 4 shows the complete schematic for this example,
along with the efficiency curve and transient response.
L1=
• 1
=1.51μH
2.25MHz •(320mA)
For the inductor, use the closest standard value of 1.5ꢀH.
The 4.4mF supercaps are used to deliver the required
2A pulses to power the RF power amplifiers, while the
LTC3606B recharges the supercap after the pulse ends,
see Figure 4c. As for the input capacitor, a typical value
Input current limit is set at 475mA average current, R
LIM
= 116k, C
= 2200pF. See Programming Input Current
LIM
Limit section for selecting R
and Selection of C
LIM
LIM
Capacitance section for C
.
LIM
L1
1.5ꢀH
V
IN
V
OUT
USB INPUT 5V
SW
V
3.4V AT
800mA
IN
LTC3606B
RUN
R
PGD
C
IN
499k
+
C
OUT
10ꢀF
R2
280k
PGOOD
RLIM
2.2mF
s2
V
FB
SuperCap
PGOOD
GND
R1
59k
C
R
LIM
2200pF
LIM
116k
I
= 475mA
LIM
L1: COILCRAFT LPS4012-152ML
C
OUT
: AVX 08056D106KAT2A
IN
C
: VISHAY 592D228X96R3X2T20H
3606B F04
Figure 4a. Design Example Circuit
3606bfa
16
LTC3606B
APPLICATIONS INFORMATION
100
10
V
= 3.4V
OUT
90
80
70
60
50
40
30
20
10
0
1
0.1
0.01
0.001
V
V
V
= 3.6V
= 4.2V
= 5V
IN
IN
IN
0.0001
0.001
0.01
0.1
1
OUTPUT CURRENT (A)
3606B F04b
Figure 4b. Efficiencꢀ vs Output Current
V
OUT
200mV/DIV
V
IN
1V/DIV
AC-COUPLED
I
OUT
500mA/DIV
I
IN
500mA/DIV
1ms/DIV
V
= 5V, 500mA COMPLIANT
IN
R
= 116kꢁ, C
= 2200pF
LIM
LIM
I
I
= 0A TO 2A, C
= 4.4mF, V
= 3.4V
LOAD
LIM
OUT
OUT
= 475mA
3606B F04c
Figure 4c. Transient Response
3606bfa
17
LTC3606B
PACKAGE DESCRIPTION
DD Package
8-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698)
0.70 p0.05
3.5 p0.05
2.10 p0.05 (2 SIDES)
1.65 p0.05
PACKAGE
OUTLINE
0.25 p 0.05
0.50
BSC
2.38 p0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
R = 0.125
0.40 p 0.10
TYP
5
8
3.00 p0.10
(4 SIDES)
1.65 p 0.10
(2 SIDES)
PIN 1
TOP MARK
(NOTE 6)
(DD8) DFN 0509 REV C
4
1
0.25 p 0.05
0.75 p0.05
0.200 REF
0.50 BSC
2.38 p0.10
BOTTOM VIEW—EXPOSED PAD
0.00 – 0.05
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON TOP AND BOTTOM OF PACKAGE
3606bfa
18
LTC3606B
REVISION HISTORY
REV
DATE
DESCRIPTION
PAGE NUMBER
A
3/10
Changes to Electrical Characteristics
3
3606bfa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC3606B
TYPICAL APPLICATIONS
800mA Buck Converter, ILIM = ±00mA
L1
1.5ꢀH
V
V
OUT
IN
V
3.4V AT
800mA
USB INPUT 5V
IN
SW
LTC3606B
R
PGD
499k
RUN
C
C
+
IN
OUT
R2
1210k
10ꢀF
2.2mF
s2
PGOOD
RLIM
V
FB
SuperCap
PGOOD
GND
R1
255k
C
R
LIM
LIM
110k
1000pF
L1: COILCRAFT LPS4012-152ML
C
C
: AVX 08056D106KAT2A
OUT
IN
3606B TA02
: VISHAY 592D228X96R3X2T20H
800mA Buck Converter, ILIM = 47±mA or Disabled
L1
1.5ꢀH
V
V
OUT
IN
V
3.4V AT
800mA
USB INPUT 5V
IN
SW
R
LTC3606B
PGD
499k
RUN
C
IN
C
+
OUT
R2
1210k
10ꢀF
2.2mF
×2
PGOOD
RLIM
V
FB
SuperCap
PGOOD
GND
R1
255k
I
LIM
R
LIM
116k
C
LIM
2200pF
DISABLE
L1: COILCRAFT LPS4012-152ML
C
C
: AVX 08056D106KAT2A
OUT
IN
: VISHAY 592D228X96R3X2T20H
3606B TA03
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC3619/LTC3619B
Dual 400mA and 800mA I , 2.25MHz,
95% Efficiency, V
= 2.5V, V
= 5.5V, V
OUT(MIN)
= 0.6V,
= 5.25V,
= 5.25V,
= 0.8V,
= 0.6V,
= 0.6V,
= 0.6V,
OUT
IN(MIN)
IN(MAX)
Synchronous Step-Down DC/DC Converter
I = 50ꢀA, I < 1ꢀA, MS10E, 3mm × 3mm DFN-10
Q SD
LTC3127
LTC3125
1.2A I , 1.6MHz, Synchronous Buck-Boost DC/DC 94% Efficiency, V
= 1.8V, V
= 5.5V, V
IN(MAX) OUT(MAX)
OUT
IN(MIN)
Converter with Adjustable Input Current Limit
I = 18ꢀA, I < 1ꢀA, 3mm × 3mm DFN-MSOP10E
Q SD
1.2A I , 1.6MHz, Synchronous Boost DC/DC
94% Efficiency, V
= 1.8V, V
= 5.5V, V
IN(MAX)
OUT
IN(MIN)
OUT(MAX)
Converter with Adjustable Input Current Limit
I = 15ꢀA, I < 1ꢀA, 2mm × 3mm DFN-8
Q SD
LTC3417A/
LTC3417A-2
LTC3407A/
LTC3407A-2
Dual 1.5A/1A, 4MHz, Synchronous Step-Down
DC/DC Converter
Dual 600mA/600mA, 1.5MHz, Synchronous
Step-Down DC/DC Converter
95% Efficiency, V
= 2.3V, V
= 5.5V, V
IN(MAX)
IN(MIN)
OUT(MIN)
I = 125ꢀA, I = <1ꢀA, TSSOP-16E, 3mm × 5mm DFN-16
Q SD
95% Efficiency, V
= 2.5V, V
= 5.5V, V
IN(MAX) OUT(MIN)
IN(MIN)
I = 40ꢀA, I = <1ꢀA, MS10E, 3mm × 3mm DFN-10
Q SD
LTC3548/LTC3548-1/ Dual 400mA and 800mA I , 2.25MHz,
LTC3548-2
LTC3546
95% Efficiency, V
= 2.5V, V
= 5.5V, V
IN(MAX) OUT(MIN)
OUT
IN(MIN)
Synchronous Step-Down DC/DC Converter
I = 40ꢀA, I = <1ꢀA, MS10E, 3mm × 3mm DFN-10
Q SD
Dual 3A/1A, 4MHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, V
= 2.3V, V
= 5.5V, V
IN(MAX) OUT(MIN)
IN(MIN)
I = 160ꢀA, I = <1ꢀA, 4mm × 5mm QFN-28
Q
SD
3606bfa
LT 0310 REV A • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
20
●
●
© LINEAR TECHNOLOGY CORPORATION 2009
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
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