MAX1953EUB [MAXIM]
Low-Cost, High-Frequency, Current-Mode PWM Buck Controller; 低成本，高频率，电流模式PWM降压控制器
| 型号： | MAX1953EUB |
| 厂家： | 
MAXIM INTEGRATED PRODUCTS |
| 描述： | Low-Cost, High-Frequency, Current-Mode PWM Buck Controller |
| 文件： | 总22页 (文件大小：642K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
19-2373; Rev 0; 4/02
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
General Description
Features
The MAX1953/MAX1954/MAX1957 is a family of versa-
tile, economical, synchronous current-mode, pulse-width
modulation (PWM) buck controllers. These step-down
controllers are targeted for applications where cost and
size are critical.
o Low-Cost Current-Mode Controllers
o Fixed-Frequency PWM
o MAX1953
1MHz Switching Frequency
Small Component Size, Low Cost
Adjustable Current Limit
The MAX1953 operates at a fixed 1MHz switching fre-
quency, thus significantly reducing external component
size and cost. Additionally, excellent transient response
is obtained using less output capacitance. The MAX1953
operates from low 3V to 5.5V input voltage and can sup-
ply up to 10A of output current. Selectable current limit is
provided to tailor to the external MOSFETs’ on-resistance
for optimum cost and performance. The output voltage is
o MAX1954
3V to 13.2V Input Voltage
25A Output Current Capability
93% Efficiency
300kHz Switching Frequency
adjustable from 0.8V to 0.86V .
IN
o MAX1957
With the MAX1954, the drain-voltage range on the high-
side FET is 3V to 13.2V and is independent of the supply
voltage. It operates at a fixed 300kHz switching frequen-
cy and can be used to provide up to 25A of output cur-
rent with high efficiency. The output voltage is adjustable
Tracking 0.4V to 0.86V Output Voltage Range
IN
Sinking and Sourcing Capability of 3A
o Shutdown Feature
o All N-Channel MOSFET Design for Low Cost
o No Current-Sense Resistor Needed
from 0.8V to 0.86V
.
HSD
The MAX1957 features a tracking output voltage range of
0.4V to 0.86V and is capable of sourcing or sinking
IN
†
o Internal Digital Soft-Start
current for applications such as DDR bus termination
and PowerPC™/ASIC/DSP core supplies. The MAX1957
operates from a 3V to 5.5V input voltage and at a fixed
300kHz switching frequency.
o Thermal Overload Protection
o Small 10-Pin µMAX Package
Ordering Information
The MAX1953/MAX1954/MAX1957 provide a COMP pin
that can be pulled low to shut down the converter in
addition to providing compensation to the error amplifier.
An input undervoltage lockout (ULVO) is provided to
ensure proper operation under power-sag conditions to
prevent the external power MOSFETs from overheating.
Internal digital soft-start is included to reduce inrush cur-
rent. The MAX1953/MAX1954/MAX1957 are available in
tiny 10-pin µMAX packages.
PART
TEMP RANGE
-40°C to +85°C
-40°C to +85°C
-40°C to +85°C
PIN-PACKAGE
10 µMAX
MAX1953EUB
MAX1954EUB
MAX1957EUB
10 µMAX
10 µMAX
Pin Configurations
TOP VIEW
Applications
Printers and Scanners
ILIM
1
2
3
4
5
10 BST
Graphic Cards and Video Cards
PCs and Servers
COMP
FB
9
8
7
6
LX
MAX1953EUB
DH
Microprocessor Core Supply
Low-Voltage Distributed Power
Telecommunications and Networking
GND
IN
PGND
DL
µMAX
†Patent Pending
PowerPC is a trademark of Motorola, Inc.
Pin Configurations continued at end of data sheet.
________________________________________________________________ Maxim Integrated Products
1
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
ABSOLUTE MAXIMUM RATINGS
IN, FB to GND...........................................................-0.3V to +6V
LX to BST..................................................................-6V to +0.3V
BST to GND............................................................-0.3V to +20V
DH to LX....................................................-0.3V to (V
DL, COMP to GND.......................................-0.3V to (V + 0.3V)
Continuous Power Dissipation (T = +70°C)
A
(derate 5.6ꢀW/°C above +70°C)..................................444ꢀW
Operating Teꢀperature Range ...........................-40°C to +85°C
Junction Teꢀperature......................................................+150°C
Storage Teꢀperature Range.............................-65°C to +150°C
Lead Teꢀperature (soldering, 10s) .................................+300°C
+ 0.3V)
BST
IN
HSD, ILIM, REFIN to GND ........................................-0.3V to 14V
PGND to GND .......................................................-0.3V to +0.3V
I , I ................................................................ 100ꢀA (RMS)
DH DL
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V = 5V, V
IN
- V = 5V, T = -40°C to +85°C, unless otherwise noted. Typical values are at T = +25°C.) (Note 1)
LX A A
BST
PARAMETER
CONDITIONS
MIN
3.0
TYP
MAX
5.5
UNITS
V
Operating Input Voltage Range
HSD Voltage Range
MAX1954 only (Note 2)
3.0
13.2
2
V
Quiescent Supply Current
V
V
V
= 1.5V, no switching
1
ꢀA
µA
FB
IN
IN
Standby Supply Current (MAX1953/ MAX1957)
= V
= 5.5V, COMP = GND
220
350
BST
= V
= 5.5V, V
= 13.2V,
BST
HSD
Standby Supply Current (MAX1954)
Undervoltage Lockout Trip Level
220
350
µA
V
COMP = GND
Rising and falling V , 3% hysteresis
2.50
0.8
2.78
2.95
IN
0.86 x
Output Voltage Adjust Range (V
)
V
OUT
V
IN
ERROR AMPLIFIER
T
T
= 0°C to +85°C (MAX1953/MAX1954)
= -40°C to +85°C (MAX1953/MAX1954)
0.788
0.776
0.8
0.8
0.812
0.812
A
A
FB Regulation Voltage
V
V
V
REFIN
+ 8ꢀV
REFIN
- 8ꢀV
MAX1957 only
V
REFIN
Transconductance
70
110
5
160
500
500
1.5
µS
nA
nA
V
FB Input Leakage Current
REFIN Input Bias Current
FB Input Coꢀꢀon-Mode Range
V
V
= 0.9V
FB
= 0.8V, MAX1957 only
5
REFIN
-0.1
-0.1
5.67
REFIN Input Coꢀꢀon-Mode Range
MAX1957 only
1.5
V
Current-Sense Aꢀplifier Voltage Gain Low
ILIM = GND (MAX1953 only)
6.3
3.5
6.93
V/V
V
= V or ILIM = open (MAX1953 only)
IN
ILIM
Current-Sense Aꢀplifier Voltage Gain
3.15
3.85
V/V
MAX1954/MAX1957
2
_______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
ELECTRICAL CHARACTERISTICS (continued)
(V = 5V, V
IN
- V = 5V, T = -40°C to +85°C, unless otherwise noted. Typical values are at T = +25°C.) (Note 1)
LX A A
BST
PARAMETER
CONDITIONS
MIN
50
TYP
125
105
210
320
210
MAX
200
125
235
350
235
UNITS
ILIM Input Iꢀpedance
Current-Liꢀit Threshold
MAX1953 only
kΩ
V
V
V
V
- V , ILIM = GND (MAX1953 only)
85
PGND
PGND
PGND
PGND
LX
- V , ILIM = open (MAX1953 only)
190
290
190
LX
ꢀV
- V , ILIM = IN (MAX1953 only)
LX
– V (MAX1954/MAX1957 only)
LX
OSCILLATOR
MAX1953
0.8
240
86
1
1.2
360
96
MHz
kHz
%
Switching Frequency
Maxiꢀuꢀ Duty Cycle
Miniꢀuꢀ Duty Cycle
SOFT-START
MAX1954/MAX1957
300
89
Measured at DH
MAX1953, ꢀeasured at DH
MAX1954/MAX1957, ꢀeasured at DH
15
18
%
4.5
5.5
MAX1953
4
Soft-Start Period
ꢀs
MAX1954/MAX1957
3.4
FET DRIVERS
DH On-Resistance, High State
DH On-Resistance, Low State
DL On-Resistance, High State
DL On-Resistance, Low State
2
3
3
3
2
Ω
Ω
Ω
Ω
1.5
2
0.8
V
= 10.5V, V = V = 5.5V,
LX IN
BST
LX, BST Leakage Current
20
30
µA
µA
MAX1953/MAX1957
V
V
= 18.7V, V = 13.2V, V = 5.5V
LX IN
= 13.2V (MAX1954 only)
BST
LX, BST, HSD Leakage Current
HSD
THERMAL PROTECTION
Therꢀal Shutdown
Rising teꢀperature
160
15
°C
°C
Therꢀal Shutdown Hysteresis
SHUTDOWN CONTROL
COMP Logic Level Low
COMP Logic Level High
COMP Pullup Current
3V < V < 5.5V
0.25
100
V
V
IN
3V < V < 5.5V
0.8
IN
µA
Note 1: Specifications to -40°C are guaranteed by design and not production tested.
Note 2: HSD and IN are externally connected for applications where V < 5.5V.
HSD
_______________________________________________________________________________________
3
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Typical Operating Characteristics
(T = +25°C, unless otherwise noted.)
A
MAX1954
EFFICIENCY vs. LOAD CURRENT
MAX1957
EFFICIENCY vs. LOAD CURRENT
MAX1953
EFFICIENCY vs. LOAD CURRENT
100
100
90
80
70
60
50
40
100
90
80
70
60
50
40
V
= 3.3V
IN
V
= 2.5V
OUT
95
90
85
80
75
70
65
60
55
50
V
= 1.7V
OUT
V
= 1.25V
OUT
V
= 5V
IN
V
= 5V
V
= 2.5V
V = 5V
IN
CIRCUIT OF FIGURE 3
IN
OUT
CIRCUIT OF FIGURE 2
CIRCUIT OF FIGURE 1
0.1
1
10
0.1
1
LOAD CURRENT (A)
10
0.1
1
LOAD CURRENT (A)
10
LOAD CURRENT (A)
MAX1953
OUTPUT VOLTAGE vs. LOAD CURRENT
MAX1954
EFFICIENCY vs. LOAD CURRENT
2.60
2.55
2.50
2.45
2.40
100
95
90
85
80
75
70
65
60
55
50
V
= 1.8V
OUT
V
= 5V
IN
V
= 3.3V
IN
V
= 12V
IN
CIRCUIT OF FIGURE 1
1.5 2.0 2.5 3.0
CIRCUIT OF FIGURE 4
0
0.5
1.0
0
5
10
15 20 25
LOAD CURRENT (A)
LOAD CURRENT (A)
MAX1954
OUTPUT VOLTAGE vs. LOAD CURRENT
MAX1954
OUTPUT VOLTAGE vs. LOAD CURRENT
2.60
2.55
2.50
2.45
2.40
2.35
1.80
1.75
1.70
1.65
1.60
1.55
V
= V = 5V
IN
V
= V = 5V
IN
HSD
HSD
CIRCUIT OF FIGURE 2
CIRCUIT OF FIGURE 2
0
1
2
3
4
5
6
0
1
2
3
4
5
6
LOAD CURRENT (A)
LOAD CURRENT (A)
4
_______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Typical Operating Characteristics (continued)
(T = +25°C, unless otherwise noted.)
A
MAX1953
OUTPUT VOLTAGE vs. INPUT VOLTAGE
MAX1954
OUTPUT VOLTAGE vs. INPUT VOLTAGE
MAX1957
OUTPUT VOLTAGE vs. LOAD CURRENT
1.35
2.60
2.55
2.50
2.45
2.40
1.76
1.74
1.72
1.70
1.68
1.66
1.64
1.30
1.25
I
= 3A
LOAD
I
= 0
LOAD
I
= 0
LOAD
V
= 5V
I
= 5A
4.0
IN
LOAD
1.20
1.15
CIRCUIT OF FIGURE 2
4.5 5.0 5.5
CIRCUIT OF FIGURE 1
4.5 5.0
INPUT VOLTAGE (V)
CIRCUIT OF FIGURE 3
3.0
3.5
4.0
5.5
3.0
3.5
-3
-2
-1
0
1
2
3
INPUT VOLTAGE (V)
LOAD CURRENT (A)
MAX1954
OUTPUT VOLTAGE vs. INPUT VOLTAGE
MAX1957
OUTPUT VOLTAGE vs. INPUT VOLTAGE
2.52
2.51
2.50
2.49
2.48
2.47
2.46
1.29
1.27
1.25
1.23
1.21
1.19
I
= 0
LOAD
I
= 0
LOAD
I
= 5A
LOAD
I
= 3A
LOAD
CIRCUIT OF FIGURE 3
4.5 5.0 5.5
CIRCUIT OF FIGURE 2
4.5 5.0
3.0
3.5
4.0
5.5
3.0
3.5
4.0
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
MAX1953
FREQUENCY vs. INPUT VOLTAGE
MAX1954/MAX1957
FREQUENCY vs. INPUT VOLTAGE
320
315
310
305
300
295
290
285
280
275
270
V
= 1.25V
OUT
V
= 2.5V
OUT
1.06
1.04
1.02
1.00
0.98
0.96
T
= -40°C
A
T
= -40°C
A
T
= +25°C
A
T
= +85°C
A
T
= +85°C
A
T
= +25°C
A
3.0
3.5
4.0
4.5
5.0
5.5
3.0
3.5
4.0
4.5
5.0
5.5
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
_______________________________________________________________________________________
5
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Typical Operating Characteristics (continued)
(T = +25°C, unless otherwise noted.)
A
MAX1954
LOAD TRANSIENT
MAX1953
LOAD TRANSIENT
MAX1953 toc16
MAX1953 toc15
V
OUT
AC-COUPLED
100mV/div
V
OUT
AC-COUPLED
100mV/div
3A
5A
2.5A
I
I
1.5A
LOAD
LOAD
CIRCUIT OF FIGURE 1
400µs/div
400µs/div
MAX1953
NO-LOAD SWITCHING WAVEFORMS
MAX1957
LOAD TRANSIENT
MAX1953 toc18
MAX1953 toc17
I
2A/div
LX
V
OUT
AC-COUPLED
50mV/div
3A
LX
DL
DH
5V/div
5V/div
5V/div
I
LOAD
-3A
2µs/div
400µs/div
6
_______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Typical Operating Characteristics (continued)
(T = +25°C, unless otherwise noted.)
A
MAX1953
MAX1954/MAX1957
NO-LOAD SWITCHING WAVEFORMS
MAX1953
FULL-LOAD SWITCHING WAVEFORMS
SHORT-CIRCUIT SWITCHING WAVEFORMS
MAX1953 toc20
MAX1953 toc21
MAX1953 toc19
I
5A/div
5V/div
I
2A/div
LX
LX
I
2A/div
LX
LX
DL
LX
DL
5V/div
5V/div
LX
DL
10V/div
5V/div
5V/div
5V/div
DH
DH
5V/div
DH
10V/div
2µs/div
4µs/div
2µs/div
MAX1954/MAX1957
FULL-LOAD SWITCHING WAVEFORMS
MAX1954/MAX1957
SHORT-CIRCUIT SWITCHING WAVEFORMS
MAX1953 toc22
MAX1953 toc23
I
2A/div
LX
I
5A/div
LX
LX
DL
DH
10V/div
LX
DL
DH
10V/div
5V/div
5V/div
10V/div
10V/div
4µs/div
4µs/div
_______________________________________________________________________________________
7
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Pin Description
PIN
NAME
FUNCTION
MAX1953
MAX1954
MAX1957
ILIM Sets the Current-Liꢀit Threshold for the Low-Side N-Channel
MOSFET, as well as the Current-Sense Aꢀplifier Gain. Connect to IN
for 320ꢀV, leave floating for 210ꢀV, or connect to GND for 105ꢀV
current-liꢀit threshold.
1
—
—
ILIM
HSD Senses the Voltage at the Drain of the High-Side N-Channel
MOSFET. Connect to the high-side MOSFET drain using a Kelvin
connection.
—
1
—
HSD
REFIN Sets the FB Regulation Voltage. Drive REFIN with the desired
FB regulation voltage using an external resistor-divider. Bypass to
GND with a 0.1µF capacitor.
—
—
1
2
REFIN
COMP
Coꢀpensation and Shutdown Control Pin. Connect an RC network to
coꢀpensate control loop. Drive to GND to shut down the IC.
2
2
Feedback Input. Regulates at V = 0.8V (MAX1953/MAX1954) or
FB
REFIN (MAX1957). Connect FB to a resistor-divider to set the output
voltage (MAX1953/MAX1954). Connect to output through a decoupling
resistor (MAX1957).
3
4
3
4
3
4
FB
GND
Ground
Input Voltage (3V to 5.5V). Provides power for the IC. For the
MAX1953/MAX1957, IN serves as the current-sense input for the high-
side MOSFET. Connect to the drain of the high-side MOSFET
(MAX1953/MAX1957). Bypass IN to GND close to the IC with a
0.22µF (MAX1954) capacitor. Bypass IN to GND close to the IC with
10µF and 4.7µF in parallel (MAX1953/MAX1957) capacitors. Use
ceraꢀic capacitors.
5
5
5
IN
Low-Side Gate-Drive Output. Drives the synchronous-rectifier MOSFET.
6
7
8
6
7
8
6
7
8
DL
PGND
DH
Swings froꢀ PGND to V .
IN
Power Ground. Connect to source of the synchronous rectifier close to
the IC.
High-Side Gate-Drive Output. Drives the high-side MOSFET. DH is a
floating driver output that swings froꢀ V to V
.
LX
BST
Master Controller Current-Sense Input. Connect LX to the junction of
the MOSFETs and inductor. LX is the reference point for the current
liꢀit.
9
9
9
LX
Boost Capacitor Connection for High-Side Gate Driver. Connect a
0.1µF ceraꢀic capacitor froꢀ BST to LX and a Schottky diode to IN.
10
10
10
BST
8
_______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Functional Diagram
IN
MAX1953
MAX1954
MAX1957
THERMAL
LIMIT
UVLO
HSD
(MAX1954
ONLY)
SLOPE
COMPENSATION
SHUTDOWN
COMPARATOR
0.5V
BST
DH
COMP
FB
ERROR
AMPLIFIER
PWM
CONTROL
CIRCUITRY
CURRENT-
SENSE
CIRCUITRY
GND
LX
DL
IN
REFERENCE
AND
SOFT-START
DAC
REFIN
(MAX1957
ONLY)
PGND
CURRENT-LIMIT
COMPARATOR
SHORT-CIRCUIT
CURRENT-LIMIT
CIRCUITRY
ILIM
(MAX1953
ONLY)
CLOCK
Typical Operating Circuit
INPUT
3V TO 5.5V
IN
ILIM
BST
DH
OUTPUT
0.8V TO 0.86V
IN
LX
DL
MAX1953
COMP
PGND
GND
FB
_______________________________________________________________________________________
9
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
DC-DC Converter Control Architecture
Detailed Description
The MAX1953/MAX1954/MAX1957 step-down convert-
The MAX1953/MAX1954/MAX1957 are single-output,
ers use a PWM, current-ꢀode control scheꢀe. An inter-
fixed-frequency, current-ꢀode, step-down, PWM, DC-
nal transconductance aꢀplifier establishes an integrated
DC converter controllers. The MAX1953 switches at
error voltage. The heart of the PWM controller is an open-
1MHz, allowing the use of sꢀall external coꢀponents for
loop coꢀparator that coꢀpares the integrated voltage-
sꢀall applications. Table 1 lists suggested coꢀponents.
feedback signal against the aꢀplified current-sense
The MAX1954 switches at 300kHz for higher efficiency
and operates froꢀ a wider range of input voltages.
Figure 1 is the MAX1953 typical application circuit. The
MAX1953/MAX1954/MAX1957 are designed to drive a
pair of external N-channel power MOSFETs in a syn-
chronous buck topology to iꢀprove efficiency and cost
coꢀpared with a P-channel power MOSFET topology.
signal plus the slope coꢀpensation raꢀp, which are
suꢀꢀed into the ꢀain PWM coꢀparator to preserve
inner-loop stability and eliꢀinate inductor staircasing. At
each rising edge of the internal clock, the high-side
MOSFET turns on until the PWM coꢀparator trips or the
ꢀaxiꢀuꢀ duty cycle is reached. During this on-tiꢀe, cur-
rent raꢀps up through the inductor, storing energy in a
ꢀagnetic field and sourcing current to the output. The
current-ꢀode feedback systeꢀ regulates the peak
inductor current as a function of the output voltage error
signal. The circuit acts as a switch-ꢀode transconduc-
tance aꢀplifier and pushes the output LC filter pole nor-
ꢀally found in a voltage-ꢀode PWM to a higher
frequency.
The on-resistance of the low-side MOSFET is used for
short-circuit current-liꢀit sensing, while the high-side
MOSFET on-resistance is used for current-ꢀode feed-
back and current-liꢀit sensing, thus eliꢀinating the
need for current-sense resistors. The MAX1953 has
three selectable short-circuit current-liꢀit thresholds:
105ꢀV, 210ꢀV, and 320ꢀV. The MAX1954 and
MAX1957 have 210ꢀV fixed short-circuit current-liꢀit
thresholds. The MAX1953/MAX1954/MAX1957 accept
input voltages froꢀ 3V to 5.5V. The MAX1954 is config-
ured with a high-side drain input (HSD) allowing an
extended input voltage range of 3V to 13.2V that is
independent of the input supply (Figure 2). The
MAX1957 is tailored for tracking output voltage applica-
tions such as DDR bus terꢀination supplies, referred to
During the second half of the cycle, the high-side MOS-
FET turns off and the low-side MOSFET turns on. The
inductor releases the stored energy as the current raꢀps
down, providing current to the output. The output capaci-
tor stores charge when the inductor current exceeds the
required load current and discharges when the inductor
current is lower, sꢀoothing the voltage across the load.
Under overload conditions, when the inductor current
exceeds the selected current-liꢀit (see the Current Limit
Circuit section), the high-side MOSFET is not turned on
at the rising clock edge and the low-side MOSFET
reꢀains on to let the inductor current raꢀp down.
as V . It utilizes a resistor-divider network connected
TT
to REFIN to keep the 1/2 ratio tracking between V
TT
and V
(Figure 3). The MAX1957 can source and
DDQ
sink up to 3A. Figure 4 shows the MAX1954 20A circuit.
The MAX1953/MAX1954/MAX1957 operate in a forced-
PWM ꢀode. As a result, the controller ꢀaintains a con-
stant switching frequency, regardless of load, to allow for
easier postfiltering of the switching noise.
Table 1. Suggested Components
DESIGNATION
MAX1953
MAX1954
MAX1957
20A CIRCUIT
10µF, 6.3V X5R CER
Taiyo Yuden
0.22µF, 10V X7R CER
Keꢀet
3 x 22µF, 6.3V X5R CER
Taiyo Yuden
0.22µF, 10V X7R CER
Keꢀet
C1
JMK212BJ106MG
C0603C224M8RAC
JMK316BJ226ML
C0603C224M8RAC
0.1µF, 50V X7R CER
Taiyo Yuden
UMK107BJ104KA
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
0.1µF, 50V X7R CER
Taiyo Yuden
UMK107BJ104KA
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
C2
C3
10µF, 6.3V X5R CER
Taiyo Yuden
0.1µF, 50V X7R CER
Taiyo Yuden
270µF, 2V SP Polyꢀer
Panasonic
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
UMK107BJ104KA
EEFUEOD271R
JMK212BJ106MG
10 ______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Table 1. Suggested Components (continued)
DESIGNATION
MAX1953
MAX1954
MAX1957
20A CIRCUIT
10µF, 6.3V X5R CER
Taiyo Yuden
180µF, 4V SP Polyꢀer
Panasonic
270µF, 2V SP Polyꢀer
Panasonic
10µF, 6.3V X5R CER
Taiyo Yuden
C4
JMK212BJ106MG
EEFUEOG181R
EEFUEOD271R
JMK212BJ106MG
4.7µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ475MG
270µF, 2V SP Polyꢀer
Panasonic
EEFUEOD271R
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
C5
C6
—
—
—
—
—
—
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
10µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ106MG
4.7µF, 6.3V X5R CER
Taiyo Yuden
JMK212BJ475MG
0.1µF, 50V X7R CER
Taiyo Yuden
UMK107BJ104KA
C7
—
—
—
—
0.1µF, 50V X7R CER
Taiyo Yuden
UMK107BJ104KA
270µF, 2V SP polyꢀer
Panasonic
EEFUEOD271R
C8
270µF, 2V SP polyꢀer
Panasonic
EEFUEOD271R
C9-C13
C14
—
1500pF, 50V X7R CER
Murata
—
GRM39X7R152K50
270pF, 10V X7R CER
Keꢀet
1000pF, 10V X7R CER
Keꢀet
470pF, 50V X7R CER
Murata
560pF, 10V X7R CER
Keꢀet
C
C
C0402C271M8RAC
C0402C102M8RAC
GRM39X7R471K50
C0402C561M8RAC
47pF, 10V C0G CER
Keꢀet
68pF, 50V COG CER
Murata
15pF, 10V C0G CER
Keꢀet
C
f
—
C0402C470K8GAC
GRM39COG680J50
C0402C150K8GAC
Schottky diode
Central Seꢀiconductor
CMPSH1-4
Schottky diode
Central Seꢀiconductor
CMPSH1-4
Schottky diode
Central Seꢀiconductor
CMPSH1-4
Schottky diode
Central Seꢀiconductor
CMPSH1-4
D1
L1
2.7µH 6.6A
Coilcraft
DO3316-272HC
2.7µH 6.6A
Coilcraft
DO3316-272HC
0.8µH 27.5A
Suꢀida
CEP125U-0R8
1µH 3.6A
Toko 817FY-1R0M
Dual MOSFET 20V 5A
Fairchild
FDS6898A
Dual MOSFET 20V
Fairchild
FDS6890A
Dual MOSFET 20V
Fairchild
FDS6898A
N-channel 30V
International Rectifier
IRF7811W
N1-N2
N3-N4
N-channel 30V
Siliconix Si4842DY
—
—
—
R1
R2
R3
16.9kΩ 1%
8.06kΩ 1%
9.09kΩ 1%
8.06kΩ 1%
2kΩ 1%
10kΩ 1%
2kΩ 1%
8.06kΩ 1%
10kΩ 5%
51.1kΩ 5%
R
33kΩ 5%
62kΩ 5%
270kΩ 5%
C
______________________________________________________________________________________ 11
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
V
IN
3V TO 5.5V
C6
10µF
C1
10µF
C5
4.7µF
D1
IN
N1
ILIM
BST
DH
L1
1µH
C2
0.1µF
V
OUT
2.5V AT 3A
R
C
LX
MAX1953
33kΩ
COMP
C3
10µF
C4
10µF
C
R1
16.9kΩ
C
DL
270pF
PGND
R2
8.06Ω
GND
FB
Figure 1. Typical Application Circuit for the MAX1953
V
IN
C2
10µF
3V TO 5.5V
C1
0.22µF
D1
IN
BST
DH
N1
V
HSD
5.5V TO 13.2V
HSD
L1
2.7µH
C3
V
OUT
1.7V AT 3A
0.1µF
R
C
LX
MAX1954
62kΩ
COMP
C4
180µF
R1
9.09kΩ
DL
C
C
1000pF
C
f
47pF
PGND
R2
8.06kΩ
GND
FB
Figure 2. Typical Application Circuit for the MAX1954
12 ______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
V
IN
3V TO 5.5V
C6
10µF
C1
3 ✕ 22µF
C7
4.7µF
D1
R
C
51.1kΩ
IN
N1
COMP
BST
DH
V
DDQ
C
C
f
68pF
C
470pF
L1
C2
0.1µF
R1
2.7µH
V
= 1/2 V
DDQ
TT
2kΩ
LX
MAX1957
C14
1500pF
REFIN
C3
270µF
C4
270µF
C5
270µF
R3
10kΩ
DL
C8
0.1µF
R2
2kΩ
PGND
GND
FB
Figure 3. Typical Application Circuit for the MAX1957
V
HSD
10.8V TO 13.2V
V
IN
3V TO 5.5V
C2
10µF
C3
10µF
C4
10µF
C5
10µF
C6
10µF
D1
HSD
BST
DH
C1
0.22µF
N1
N3
N2
N4
IN
L1
0.8µH
C7
0.1µF
V
OUT
1.8V AT 20A
R
C
LX
MAX1954
270kΩ
COMP
C8
270µF
C9
270µF
C10
270µF
C11
270µF
C12
C13
270µF
R1
10kΩ
DL
270µF
C
C
560pF
C
f
15pF
PGND
R2
8.06kΩ
GND
FB
Figure 4. 20A Circuit
______________________________________________________________________________________ 13
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Current-Sense Amplifier
Synchronous Rectifier Driver (DL)
Synchronous rectification reduces conduction losses in
the rectifier by replacing the norꢀal Schottky catch
diode with a low-resistance MOSFET switch. The
MAX1953/MAX1954/MAX1957 use the synchronous
rectifier to ensure proper startup of the boost gate-
driver circuit and to provide the current-liꢀit signal. The
DL low-side waveforꢀ is always the coꢀpleꢀent of the
DH high-side drive waveforꢀ. A dead-tiꢀe circuit ꢀoni-
tors the DL output and prevents the high-side MOSFET
froꢀ turning on until DL is fully off, thus preventing
cross-conduction or shoot-through. In order for the
dead-tiꢀe circuit to work properly, there ꢀust be a low-
resistance, low-inductance path froꢀ the DL driver to
the MOSFET gate. Otherwise, the sense circuitry in the
MAX1953/MAX1954/MAX1957 can interpret the MOS-
FET gate as OFF when gate charge actually reꢀains.
The dead tiꢀe at the other edge (DH turning off) is
deterꢀined through gate sensing as well.
The MAX1953/MAX1954/MAX1957s’ current-sense cir-
cuit aꢀplifies (A = 3.5 typ) the current-sense voltage
V
(the high-side MOSFET’s on-resistance (R
) ꢀulti-
DS(ON)
plied by the inductor current). This aꢀplified current-
sense signal and the internal-slope coꢀpensation
signal are suꢀꢀed (V
) together and fed into the
SUM
PWM coꢀparator’s inverting input. The PWM coꢀpara-
tor shuts off the high-side MOSFET when V
SUM
exceeds the integrated feedback voltage (V
).
COMP
Current-Limit Circuit
The current-liꢀit circuit eꢀploys a lossless current-liꢀit-
ing algorithꢀ that uses the low-side and high-side
MOSFETs’ on-resistances as the sensing eleꢀents. The
voltage across the high-side MOSFET is ꢀonitored for
current-ꢀode feedback, as well as current liꢀit. This
signal is aꢀplified by the current-sense aꢀplifier and is
coꢀpared with a current-sense voltage. If the current-
sense signal is larger than the set current-liꢀit voltage,
the high-side MOSFET turns off. Once the high-side
MOSFET turns off, the low-side MOSFET is ꢀonitored
for current liꢀit. If the voltage across the low-side MOS-
✕
High-Side Gate-Drive Supply (BST)
Gate-drive voltage for the high-side switch is generated
by a flying capacitor boost circuit (Figure 5). The
capacitor between BST and LX is charged froꢀ the V
IN
supply up to V , ꢀinus the diode drop while the low-
IN
side MOSFET is on. When the low-side MOSFET is
switched off, the stored voltage of the capacitor is
stacked above LX to provide the necessary turn-on
voltage (V ) for the high-side MOSFET. The controller
GS
then closes an internal switch between BST and DH to
turn the high-side MOSFET on.
FET (R
I
) does not exceed the short-
DS(ON)
INDUCTOR
circuit current liꢀit, the high-side MOSFET turns on
norꢀally. In this condition, the output drops sꢀoothly
out of regulation. If the voltage across the low-side
MOSFET exceeds the short-circuit current-liꢀit thresh-
old at the beginning of each new oscillator cycle, the
MAX1953/MAX1954/MAX1957 do not turn on the high-
side MOSFET.
In the case where the output is shorted, the low-side
MOSFET is ꢀonitored for current liꢀit. The low-side
MOSFET is held on to let the current in the inductor
raꢀp down. Once the voltage across the low-side
MOSFET drops below the short-circuit current-liꢀit
threshold, the high-side MOSFET is pulsed. Under this
condition, the frequency of the MAX1953/MAX1954/
MAX1957 appears to decrease because the on-tiꢀe of
the low-side MOSFET extends beyond a clock cycle.
Undervoltage Lockout
If the supply voltage at IN drops below 2.75V, the
MAX1953/MAX1954/MAX1957 assuꢀe that the supply
voltage is too low to ꢀake valid decisions, so the UVLO
circuitry inhibits switching and forces the DL and DH
gate drivers low. After the voltage at IN rises above
2.8V, the controller goes into the startup sequence and
resuꢀes norꢀal operation.
Startup
The MAX1953/MAX1954/MAX1957 start switching when
the voltage at IN rises above the UVLO threshold.
However, the controller is not enabled unless all four of
the following conditions are ꢀet:
The actual peak output current is greater than the
short-circuit current-liꢀit threshold by an aꢀount equal
to the inductor ripple current. Therefore, the exact cur-
rent-liꢀit characteristic and ꢀaxiꢀuꢀ load capability
are a function of the low-side MOSFET on-resistance,
inductor value, input voltage, and output voltage.
• V exceeds the 2.8V UVLO threshold.
IN
The short-circuit current-liꢀit threshold is preset for the
MAX1954/MAX1957 at 210ꢀV. The MAX1953, however,
has three options for the current-liꢀit threshold: con-
nect ILIM to IN for a 320ꢀV threshold, connect ILIM to
GND for 105ꢀV, or leave floating for 210ꢀV.
• The internal reference voltage exceeds 92% of its
noꢀinal value (V
> 1 V).
REF
• The internal bias circuitry powers up.
• The therꢀal overload liꢀit is not exceeded.
14 ______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Design Procedures
Setting the Output Voltage
To set the output voltage for the MAX1953/MAX1954,
IN
connect FB to the center of an external resistor-divider
BST
connected between the output to GND (Figures 1 and
DH
LX
DL
2). Select R2 between 8kΩ and 24kΩ, and then calcu-
late R1 by:
MAX1953
MAX1954
MAX1957
V
V
OUT
R1 = R2 ×
−1
FB
where V
= 0.8V. R1 and R2 should be placed as
FB
close to the IC as possible.
For the MAX1957, connect FB directly to the output
through a decoupling resistor of 10kΩ to 21kΩ (Figure
3). The output voltage is then equal to the voltage at
REFIN. Again, this resistor should be placed as close to
the IC as possible.
Figure 5. DH Boost Circuit
Once these conditions are ꢀet, the step-down controller
enables soft-start and starts switching. The soft-start cir-
cuitry gradually raꢀps up to the feedback-regulation
voltage in order to control the rate-of-rise of the output
voltage and reduce input surge currents during startup.
The soft-start period is 1024 clock cycles (1024/f ,
MAX1954/MAX1957) or 4096 clock cycles (4096/f ,
MAX1953) and the internal soft-start DAC raꢀps the
voltage up in 64 steps. The output reaches regulation
when soft-start is coꢀpleted, regardless of output
capacitance and load.
Determining the Inductor Value
There are several paraꢀeters that ꢀust be exaꢀined
when deterꢀining which inductor is to be used. Input
voltage, output voltage, load current, switching frequen-
cy, and LIR. LIR is the ratio of inductor current ripple to
DC load current. A higher LIR value allows for a sꢀaller
inductor, but results in higher losses and higher output
ripple. A good coꢀproꢀise between size, efficiency,
and cost is an LIR of 30%. Once all of the paraꢀeters
are chosen, the inductor value is deterꢀined as follows:
S
S
Shutdown
The MAX1953/MAX1954/MAX1957 feature a low-power
shutdown ꢀode. Use an open-collector transistor to
pull COMP low to shut down the IC. During shutdown,
the output is high iꢀpedance. Shutdown reduces the
V
× V − V
OUT
(
IN
OUT
)
L =
V
× f × I
× LIR
IN
S
LOAD MAX
quiescent current (I ) to approxiꢀately 220µA.
Q
where f is the switching frequency. Choose a standard
S
Thermal Overload Protection
Therꢀal overload protection liꢀits total power dissipation
in the MAX1953/MAX1954/MAX1957. When the junction
value close to the calculated value. The exact inductor
value is not critical and can be adjusted in order to
ꢀake trade-offs aꢀong size, cost, and efficiency. Lower
inductor values ꢀiniꢀize size and cost, but they also
increase the output ripple and reduce the efficiency due
to higher peak currents. By contrast, higher inductor val-
ues increase efficiency, but eventually resistive losses
due to extra turns of wire exceed the benefit gained
froꢀ lower AC current levels.
teꢀperature exceeds T = +160°C, an internal therꢀal
J
sensor shuts down the device, allowing the IC to cool.
The therꢀal sensor turns the IC on again after the junc-
tion teꢀperature cools by 15°C, resulting in a pulsed out-
put during continuous therꢀal overload conditions.
For any area-restricted applications, find a low-core
loss inductor having the lowest possible DC resistance.
Ferrite cores are often the best choice, although pow-
dered iron is inexpensive and can work well at 300kHz.
______________________________________________________________________________________ 15
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
The chosen inductor’s saturation current rating ꢀust
exceed the expected peak inductor current (IPEAK).
Deterꢀine IPEAK as:
FET. A good general rule is to allow 0.5% additional
resistance for each °C of MOSFET junction teꢀperature
rise. The calculated V
ꢀust be less than V
.
CS
VALLEY
For the MAX1953, connect ILIM to GND for a short-
circuit current-liꢀit voltage of 105ꢀV, to V for 320ꢀV
IN
LIR
2
I
= I
+
× I
LOAD MAX
PEAK LOAD MAX
(
)
(
)
or leave ILIM floating for 210ꢀV.
MOSFET Selection
The MAX1953/MAX1954/MAX1957 drive two external,
logic-level, N-channel MOSFETs as the circuit switch
eleꢀents. The key selection paraꢀeters are:
Setting the Current Limit
The MAX1953/MAX1954/MAX1957 use a lossless cur-
rent-sense ꢀethod for current liꢀiting. The voltage
drops across the MOSFETs created by their on-resis-
tances are used to sense the inductor current.
Calculate the current-liꢀit threshold as follows:
• On-Resistance (R
): The lower, the better.
DS(ON)
• Maximum Drain-to-Source Voltage (V
): Should
DSS
be at least 20% higher than the input supply rail at
0.8V
the high side MOSFET’s drain.
V
=
CS
A
• Gate Charges (Q , Q , Q ): The lower, the better.
CS
g
gd
gs
For a 3.3V input application, choose a MOSFET with a
rated R at V = 2.5V. For a 5V input application,
where A
CS
is the gain of the current-sense aꢀplifier.
CS
DS(ON)
GS
A
is 6.3 for the MAX1953 when ILIM is connected to
choose the MOSFETs with rated R
at V
≤ 4.5V.
DS(ON)
GS
GND and 3.5 for the MAX1954/MAX1957, and for the
MAX1953 when ILIM is connected to IN or floating. The
0.8V is the usable dynaꢀic range of COMP (V
For a good coꢀproꢀise between efficiency and cost,
choose the high-side MOSFET (N1) that has conduction
losses equal to switching loss at the noꢀinal input volt-
age and output current. The selected low-side and high-
side MOSFETs (N2 and N1, respectively) ꢀust have
).
COMP
Initially, the high-side MOSFET is ꢀonitored. Once the
voltage drop across the high-side MOSFET exceeds V
,
CS
the high-side MOSFET is turned off and the low-side
MOSFET is turned on. The voltage across the low-side
MOSFET is then ꢀonitored. If the voltage across the low-
side MOSFET exceeds the short-circuit current liꢀit, a
short-circuit condition is deterꢀined and the low-side
MOSFET is held on. Once the ꢀonitored voltage falls
below the short-circuit current-liꢀit threshold, the
MAX1953/MAX1954/MAX1957 switch norꢀally. The short-
circuit current-liꢀit threshold is fixed at 210ꢀV for the
MAX1954/ MAX1957 and is selectable for the MAX1953.
R
that satisfies the current-liꢀit setting condition
DS(ON)
above. For N2, ꢀake sure that it does not spuriously turn
on due to dV/dt caused by N1 turning on, as this would
result in shoot-through current degrading the efficiency.
MOSFETs with a lower Q /Q ratio have higher iꢀꢀu-
gd gs
nity to dV/dt.
For proper therꢀal ꢀanageꢀent design, the power dis-
sipation ꢀust be calculated at the desired ꢀaxiꢀuꢀ
operating junction teꢀperature, T
. N1 and N2
J(MAX)
have different loss coꢀponents due to the circuit oper-
ation. N2 operates as a zero-voltage switch; therefore,
When selecting the high-side MOSFET, use the follow-
ing ꢀethod to verify that the MOSFET’s R
is suffi-
DS(ON)
ꢀajor losses are the channel conduction loss (P
)
N2CC
ciently low at the operating junction teꢀperature (T ):
J
and the body diode conduction loss (P
):
N2DC
0.8V
× I
PEAK
USER
AT T
J(MAX)
R
≤
DS(ON)
DS(ON)N1
A
CS
V
2
OUT
P
= (1−
) × I
× R
LOAD
N2CC
DS(ON)
The voltage drop across the low-side MOSFET at the
V
IN
valley point and at I
is:
LOAD(MAX)
P
= 2 × I
× V × t
× f
DT S
N2DC
LOAD
F
LIR
2
where V is the body diode forward-voltage drop, t is
F
dt
V
=R
× (I
−
× I
)
VALLEY
DS(ON)
LOAD(MAX)
LOAD MAX
(
)
the dead tiꢀe between N1 and N2 switching transi-
tions, and f is the switching frequency.
S
where R
is the ꢀaxiꢀuꢀ value at the desired
ꢀaxiꢀuꢀ operating junction teꢀperature of the MOS-
DS(ON)
16 ______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
N1 operates as a duty-cycle control switch and has the
following ꢀajor losses: the channel conduction loss
ꢀended due to their low ESR and ESL at high frequency,
with relatively low cost. Choose a capacitor that exhibits
less than 10°C teꢀperature rise at the ꢀaxiꢀuꢀ operat-
ing RMS current for optiꢀuꢀ long-terꢀ reliability.
(P
), the voltage and current overlapping switching
N1CC
loss (P
), and the drive loss (P
).
N1SW
N1DR
Output Capacitor
The key selection paraꢀeters for the output capacitor
are the actual capacitance value, the equivalent series
resistance (ESR), the equivalent series inductance
(ESL), and the voltage-rating requireꢀents. These para-
ꢀeters affect the overall stability, output voltage ripple,
and transient response. The output ripple has three
coꢀponents: variations in the charge stored in the out-
put capacitor, the voltage drop across the capacitor’s
ESR, and the voltage drop across the ESL caused by
the current into and out of the capacitor:
V
2
OUT
P
=
× I
× R
USE R
AT T
DS(ON) J(MAX)
LOAD
N1CC
DS(ON)
(
)
V
IN
Q
+ Q
GD
GS
I
P
= V × I
IN LOAD
×
× f
S
N2SW
GATE
where I
is the average DH driver output current
GATE
capability deterꢀined by:
1
2
V
IN
I
≅
×
GATE
R
+ R
DH GATE
V
= V
+ V
+ V
RIPPLE
RIPPLE
(ESR)
RIPPLE(C) RIPPLE(ESL)
where R
is the high-side MOSFET driver’s on-resis-
DH
tance (3Ω ꢀax) and R
is the internal gate resis-
GATE
The output voltage ripple as a consequence of the ESR,
ESL, and output capacitance is:
tance of the MOSFET (~ 2Ω):
V
= I
× ESR
RIPPLE
(ESR)
P−P
R
GATE
P
= Q × V × f
×
N1DR
G
GS
S
I
P−P
R
+ R
DH
GATE
V
RIPPLE(C)
8×C
× f
OUT
S
where V
~ V . In addition to the losses above, allow
IN
GS
V
about 20% ꢀore for additional losses due to MOSFET
output capacitances and N2 body diode reverse recov-
ery charge dissipated in N1 that exists, but is not well
defined in the MOSFET data sheet. Refer to the MOS-
FET data sheet for the therꢀal-resistance specification
to calculate the PC board area needed to ꢀaintain the
desired ꢀaxiꢀuꢀ operating junction teꢀperature with
the above calculated power dissipations.
IN
V
ESL
RIPPLE(ESL) =
L
V
− V
OUT
V
IN
OUT
I
=
×
P−P
f
× L
V
IN
S
where I
is the peak-to-peak inductor current (see the
P-P
Determining the Inductor Value section). These equa-
tions are suitable for initial capacitor selection, but final
values should be chosen based on a prototype or eval-
uation circuit.
The ꢀiniꢀuꢀ load current ꢀust exceed the high-side
MOSFET’s ꢀaxiꢀuꢀ leakage current over teꢀperature
if fault conditions are expected.
As a general rule, a sꢀaller current ripple results in less
output voltage ripple. Since the inductor ripple current
is a factor of the inductor value and input voltage, the
output voltage ripple decreases with larger inductance,
and increases with higher input voltages. Ceraꢀic
capacitors are recoꢀꢀended for the MAX1953 due to
its 1MHz switching frequency. For the MAX1954/
MAX1957, using polyꢀer, tantaluꢀ, or aluꢀinuꢀ elec-
trolytic capacitors is recoꢀꢀended. The aluꢀinuꢀ
electrolytic capacitor is the least expensive; however, it
has higher ESR. To coꢀpensate for this, use a ceraꢀic
capacitor in parallel to reduce the switching ripple and
noise. For reliable and safe operation, ensure that the
capacitor’s voltage and ripple-current ratings exceed
the calculated values.
Input Capacitor
The input filter capacitor reduces peak currents drawn
froꢀ the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitor ꢀust ꢀeet the ripple current
requireꢀent (I
) iꢀposed by the switching currents
RMS
defined by the following equation:
I
×
V
× V −V
(
)
LOAD
OUT
IN
OUT
I
=
RMS
V
IN
I
has a ꢀaxiꢀuꢀ value when the input voltage
RMS
equals twice the output voltage (V = 2 x V
), where
IN
OUT
I
= I
/2. Ceraꢀic capacitors are recoꢀ-
RMS(MAX)
LOAD
______________________________________________________________________________________ 17
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
The MAX1953/MAX1954/MAX1957s’ response to a load
transient depends on the selected output capacitors. In
general, ꢀore low-ESR output capacitance results in
better transient response. After a load transient, the
Below are equations that define the power ꢀodulator:
R
R
× (f × L)
LOAD
S
G
= g
×
ꢀc
MOD
+ f × L
LOAD
S
output voltage instantly changes by ESR ✕ ∆I
.
LOAD
Before the controller can respond, the output voltage
deviates further, depending on the inductor and output
capacitor values. After a short period of tiꢀe (see the
Typical Operating Characteristics), the controller
responds by regulating the output voltage back to its
noꢀinal state. The controller response tiꢀe depends on
its closed-loop bandwidth. With a higher bandwidth,
the response tiꢀe is faster, preventing the output volt-
age froꢀ further deviation froꢀ its regulating value.
where R
DS(ON)
= V
/I
, and g
= 1/(A
✕
LOAD
), where A
aꢀplifier and R
OUT OUT(MAX)
ꢀc
CS
R
is the gain of the current-sense
is the on-resistance of the high-
side power MOSFET. A is 6.3 for the MAX1953 when
CS
DS(ON)
CS
ILIM is connected to GND, and 3.5 for the MAX1954/
MAX1957 and for the MAX1953 when ILIM is connect-
ed to V or floating. The frequencies at which the pole
IN
and zero due to the power ꢀodulator occur are deter-
ꢀined as follows:
1
Compensation Design
The MAX1953/MAX1954/MAX1957 use an internal
transconductance error aꢀplifier whose output coꢀ-
pensates the control loop. The external inductor, high-
side MOSFET, output capacitor, coꢀpensation resistor,
and coꢀpensation capacitors deterꢀine the loop sta-
bility. The inductor and output capacitors are chosen
based on perforꢀance, size, and cost. Additionally, the
coꢀpensation resistor and capacitors are selected to
optiꢀize control-loop stability. The coꢀponent values
shown in the Typical Application Circuits (Figures 1
through 4) yield stable operation over the given range
of input-to-output voltages and load currents.
f
=
=
pMOD
R
× f × L + R
LOAD
R
(
)
ESR
S
2π × C
×
OUT
+ f × L
LOAD
(
)
S
1
f
zMOD
2π × C
× R
OUT
ESR
The feedback voltage-divider used has a gain of G
=
FB
V
/V
, where V
is equal to 0.8V. The transcon-
FB OUT
FB
ductance error aꢀplifier has DC gain, G
= gꢀ ✕
EA(DC)
R . R is typically 10MΩ. A doꢀinant pole is set by the
O
O
coꢀpensation capacitor (C ), the aꢀplifier output
C
resistance (R ), and the coꢀpensation resistor (R ). A
O
C
The controller uses a current-ꢀode control scheꢀe that
regulates the output voltage by forcing the required
current through the external inductor. The MAX1953/
MAX1954/MAX1957 use the voltage across the high-
zero is set by the coꢀpensation resistor (R ) and the
C
coꢀpensation capacitor (C ).
C
There is an optional pole set by C and R to cancel the
output capacitor ESR zero if it occurs before crossover
f
C
side MOSFET’s on-resistance (R
) to sense the
DS(ON)
frequency (f ):
C
inductor current. Current-ꢀode control eliꢀinates the
double pole in the feedback loop caused by the induc-
tor and output capacitor, resulting in a sꢀaller phase
shift and requiring less elaborate error-aꢀplifier coꢀ-
1
f
=
pdEA
2π × C × (R + R )
C
O
C
pensation. A siꢀple single-series R and C is all that
1
C
C
fzEA =
fpEA =
is needed to have a stable high bandwidth loop in
applications where ceraꢀic capacitors are used for
output filtering. For other types of capacitors, due to the
higher capacitance and ESR, the frequency of the zero
created by the capacitance and ESR is lower than the
desired close loop crossover frequency. Another coꢀ-
pensation capacitor should be added to cancel this
ESR zero.
2π × C × R
C
1
C
2π × C × R
f
C
The crossover frequency (f ) should be ꢀuch higher
than the power ꢀodulator pole f
crossover frequency should be less than 1/5 the
switching frequency:
C
. Also, the
pMOD
The basic regulator loop ꢀay be thought of as a power
ꢀodulator, output feedback divider, and an error aꢀpli-
f
S
f
<< f <
C
pMOD
fier. The power ꢀodulator has DC gain set by g
x
ꢀc
, the out-
), and its equivalent series resis-
5
R
, with a pole and zero pair set by R
LOAD
put capacitor (C
LOAD
OUT
tance (R
).
ESR
18 ______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Table 2. Suggested Manufacturers
MANUFACTURER
Central Seꢀiconductor
Coilcraft
COMPONENT
Diode
PHONE
WEBSITE
www.centralseꢀi.coꢀ
www.coilcraft.coꢀ
www.fairchildseꢀi.coꢀ
www.keꢀet.coꢀ
631-435-1110
800-322-2645
800-341-0392
864-963-6300
714-373-7366
408-573-4150
800-745-8656
Inductors
MOSFETs
Capacitors
Capacitors
Capacitors
Inductors
Fairchild
Keꢀet
Panasonic
www.panasonic.coꢀ
www.t-yuden.coꢀ
www.toko.coꢀ
Taiyo Yuden
Toko
so the loop-gain equation at the crossover frequency is:
Applications Information
See Table 2 for suggested ꢀanufacturers of the coꢀ-
ponents used with the MAX1953/MAX1954/MAX1957.
V
FB
G
× G
×
= 1
EA(fC )
MOD(fC )
V
OUT
PC Board Layout Guidelines
Careful PC board layout is critical to achieve low
switching losses and clean, stable operation. The
switching power stage requires particular attention.
Follow these guidelines for good PC board layout:
For the case where f
G
is greater than f :
c
zESR
= g
× R
C
EA(fC )
ꢀEA
and
1) Place decoupling capacitors as close to IC pins as
possible. Keep separate power ground plane (con-
nected to pin 7) and signal ground plane (connect-
ed to pin 4).
f
R
× (f × L)
pMOD
LOAD
s
G
= g
×
ꢀc
×
MOD(fC )
R
+ (f × L)
f
C
LOAD
s
then R is calculated as:
C
2) Input and output capacitors are connected to the
power ground plane; all other capacitors are con-
nected to the signal ground plane.
V
OUT
R
=
C
g
× V
× G
ꢀEA
FB MOD(fC )
3) Keep the high current paths as short as possible.
where g
= 110µS.
ꢀEA
4) Connect the drain leads of the power MOSFET to a
large copper area to help cool the device. Refer to
the power MOSFET data sheet for recoꢀꢀended
copper area.
The error aꢀplifier coꢀpensation zero forꢀed by R
C
.
and C should be set at the ꢀodulator pole f
C
pMOD
C
is calculated by:
C
V
OUT
× (f × L)
5) Ensure all feedback connections are short and
direct. Place the feedback resistors as close to the
IC as possible.
S
I
C
OUT(MAX)
OUT
C
=
×
C
V
R
C
OUT
+ (f × L)
S
6) Route high-speed switching nodes away froꢀ sensi-
tive analog areas (FB, COMP).
I
OUT(MAX)
As the load current decreases, the ꢀodulator pole also
decreases. However, the ꢀodulator gain increases
accordingly, and the crossover frequency reꢀains the
7) Place the high-side MOSFET as close as possible to
the controller and connect IN (MAX1953/MAX1957)
or HSD (MAX1954) and LX to the MOSFET.
saꢀe. For the case where f
is less than f , add
zESR
C
8) Use very short, wide traces (50ꢀils to 100ꢀils wide
if the MOSFET is 1in froꢀ the device).
another coꢀpensation capacitor C froꢀ COMP to GND
f
to cancel the ESR zero at f
. C is calculated by:
zESR
f
Chip Information
TRANSISTOR COUNT: 2930
1
C =
f
2π × R × f
C
zESR
PROCESS: BiCMOS
Figure 6 illustrates a nuꢀerical exaꢀple that calculates
and C values for the typical application circuit of
R
C
C
Figure 1 (MAX1953).
______________________________________________________________________________________ 19
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
V
= 2.5V
OUT
I
= 3A
OUT(MAX)
C
= 20µF
OUT
L=1µH
R
= 0.0025Ω
=110µS
= 6.3A
ESR
g
ꢀEA
A
VCS
R
= 0.013Ω
DS(ON)
1
g
=
=12.21S
ꢀc
A
× R
DS(ON)
VCS
f =1MHz
S
V
2.5V
OUT
R
=
=
= 0.833Ω
LOAD
I
3A
OUT(MAX)
1
1
f
=
=
=17.42kHz
pMOD
R
× f × L
LOAD
(
)
S
0.833Ω × 1MHz × 1µH
0.833Ω + 1MHz × 1µH
2π × C
×
+R
)
2π × 20µF ×
+ 0.0025Ω
OUT
1
ESR
R
f
× L
LOAD
(
)
(
)
S
1
f
=
=
= 3.2MHz
zESR
2π × C
× R
2π× 20µF × .0025Ω
OUT
ESR
Pick the crossover frequency (f ) at <1/5 the switching frequency (f ). We choose100kHz < f
,so C
F
C
S
zESR
is not needed. The power ꢀodulator gain at f is:
C
f
R
× (f × L)
0.833Ω × (1MHz × 1µH)
0.833Ω +(1MHz × 1µH)
17.42kHz
100kHz
pMOD
LOAD
S
G
= g
×
×
=12.21S ×
×
= 0.967
MOD(f )
ꢀc
C
R
(f × L)
f
C
LOAD S
then:
V
2.5V
OUT
R
=
=
≈ 33kΩ
C
g
× V × G
110µS × 0.8V × .937
ꢀEA
FB
MOD(f )
C
And:
V
OUT
2.5V
× (f × L)
S
× (1MHz × 1µH)
I
C
20µF
33kΩ
OUT(MAX)
OUT
3A
2.5V
C
=
×
=
×
≈ 270pF
C
V
R
OUT
C
+(f × L)
+(1MHz × 1µH)
S
I
3A
OUT(MAX)
Figure 6. Numerical Example to Calculate R and C Values of the Typical Operating Circuit of Figure 1 (MAX1953)
C
C
20 ______________________________________________________________________________________
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Pin Configurations (continued)
TOP VIEW
HSD
COMP
FB
1
2
3
4
5
10 BST
REFIN
COMP
FB
1
2
3
4
5
10 BST
9
8
7
6
LX
9
8
7
6
LX
MAX1954EUB
MAX1957EUB
DH
DH
GND
IN
PGND
DL
GND
IN
PGND
DL
µMAX
µMAX
______________________________________________________________________________________ 21
Low-Cost, High-Frequency, Current-Mode PWM
Buck Controller
Package Information
(The package drawing(s) in this data sheet ꢀay not reflect the ꢀost current specifications. For the latest package outline inforꢀation,
go to www.maxim-ic.com/packages.)
e
4X S
10
10
INCHES
MAX
MILLIMETERS
MAX
1.10
0.15
0.95
3.05
3.00
3.05
3.00
5.05
0.70
DIM MIN
MIN
-
A
-
0.043
0.006
0.037
0.120
0.118
0.120
0.118
0.199
A1
A2
D1
D2
E1
E2
H
0.002
0.030
0.116
0.114
0.116
0.114
0.187
0.05
0.75
2.95
2.89
2.95
2.89
4.75
0.40
H
ÿ 0.50±0.1
0.6±0.1
L
0.0157 0.0275
0.037 REF
L1
b
0.940 REF
0.007
0.0106
0.177
0.270
0.200
1
1
e
0.0197 BSC
0.500 BSC
0.6±0.1
c
0.0035 0.0078
0.0196 REF
0.090
BOTTOM VIEW
0.498 REF
S
α
TOP VIEW
0∞
6∞
0∞
6∞
D2
E2
GAGE PLANE
A2
c
A
E1
b
L
α
A1
D1
L1
FRONT VIEW
SIDE VIEW
PROPRIETARY INFORMATION
TITLE:
PACKAGE OUTLINE, 10L uMAX/uSOP
APPROVAL
DOCUMENT CONTROL NO.
REV.
1
21-0061
I
1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
22 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2002 Maxiꢀ Integrated Products
Printed USA
is a registered tradeꢀark of Maxiꢀ Integrated Products.
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