MIC26601 [MICREL]
28V, 6A Hyper Speed Control⢠Synchronous DC/DC Buck Regulator; 28V ，6A的Hyper Speed Controlâ ?? ¢同步DC / DC降压稳压器
| 型号： | MIC26601 |
| 厂家： | 
MICREL SEMICONDUCTOR |
| 描述： | 28V, 6A Hyper Speed Control⢠Synchronous DC/DC Buck Regulator |
| 文件： | 总30页 (文件大小：1094K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC26601
28V, 6A Hyper Speed Control™
Synchronous DC/DC Buck Regulator
SuperSwitcher II™
General Description
Features
The Micrel MIC26601 is a constant-frequency, synchronous
buck regulator featuring a unique adaptive on-time control
architecture. The MIC26601 operates over an input supply
range of 4.5V to 28V and provides a regulated output of up to
6A of output current. The output voltage is adjustable down to
0.8V with a guaranteed accuracy of ±1%, and the device
operates at a switching frequency of 600kHz.
•
Hyper Speed Control™ architecture enables
- High Delta V operation (VIN = 28V and VOUT = 0.8V)
- Small output capacitance
•
•
•
•
•
•
•
•
•
4.5V to 28V voltage input
6A output current capability, up to 95% efficiency
Adjustable output from 0.8V to 5.5V
±1% feedback accuracy
Micrel’s Hyper Speed Control™ architecture allows for ultra-
fast transient response while reducing the output capacitance
and also makes (High VIN)/(Low VOUT) operation possible.
This adaptive tON ripple control architecture combines the
advantages of fixed-frequency operation and fast transient
response in a single device.
Any Capacitor™ stable - zero-to-high ESR
600kHz switching frequency
No external compensation
Power Good (PG) output
Foldback current-limit and “hiccup mode” short-circuit
protection
Supports safe startup into a pre-biased load
–40°C to +125°C junction temperature range
28-pin 5mm × 6mm MLF® package
The MIC26601 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, foldback current limit, “hiccup mode” short-
circuit protection and thermal shutdown. An open-drain
Power Good (PG) pin is provided.
•
•
•
Applications
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
•
•
•
•
Distributed power systems
Communications/networking infrastructure
Set-top box, gateways, and routers
Printers, scanners, graphic cards, and video cards
_________________________________________________________________________________________________________________________
Typical Application
Efficiency (VIN = 12V)
vs. Output Current
100
5.0V
3.3V
95
90
85
80
75
70
65
60
55
50
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
0
1
2
3
4
5
6
7
8
OUTPUT CURRENT (A)
Hyper Speed Control, SuperSwitcher II, and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
M9999-071311-A
July 2011
Micrel, Inc.
MIC26601
Ordering Information
Junction Temperature
Range
Lead
Finish
Part Number
Voltage
Switching Frequency
Package
28-Pin 5mm × 6mm MLF®
MIC26601YJL
Adjustable
600kHz
–40°C to +125°C
Pb-Free
Pin Configuration
28-Pin 5mm × 6mm MLF® (YJL)
Pin Description
Pin Number
Pin Name Pin Function
5V Internal Linear Regulator (Output): PVDD supply is the power MOSFET gate drive supply voltage
and created by internal LDO from VIN. When VIN < +5.5V, PVDD should be tied to PVIN pins. A 2.2µF
ceramic capacitor from the PVDD pin to PGND (Pin 2) must be place next to the IC.
1
3
PVDD
NC
No Connect.
Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET
drain. Due to the high-speed switching on this pin, the SW pin should be routed away from sensitive
nodes.
4, 9, 10,
11, 12
SW
Power Ground. PGND is the ground path for the MIC26601 buck converter power stage. The PGND
pins connect to the low-side N-Channel internal MOSFET gate drive supply ground, the sources of
the MOSFETs, the negative terminals of input capacitors, and the negative terminals of output
capacitors. The loop for the power ground should be as small as possible and separate from the
Signal ground (SGND) loop.
2, 5, 6, 7, 8,
21
PGND
High-Side N-internal MOSFET Drain Connection (Input): The PVIN operating voltage range is from
4.5V to 28V. Input capacitors between the PVIN pins and the Power Ground (PGND) are required and
keep the connection short.
13,14,15,
PVIN
BST
16,17,18,19
Boost (Output): Bootstrapped voltage to the high-side N-channel MOSFET driver. A Schottky diode is
connected between the PVDD pin and the BST pin. A boost capacitor of 0.1μF is connected between
the BST pin and the SW pin. Adding a small resistor at the BST pin can slow down the turn-on time of
high-side N-Channel MOSFETs.
20
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Micrel, Inc.
MIC26601
Pin Description (Continued)
Pin Number
Pin Name Pin Function
Current Sense (Input): The CS pin senses current by monitoring the voltage across the low-side
MOSFET during the OFF-time. The current sensing is necessary for short circuit protection. In order
to sense the current accurately, connect the low-side MOSFET drain to SW using a Kelvin
connection. The CS pin is also the high-side MOSFET’s output driver return.
22
CS
Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin to
the PGND Pad on the top layer (see PCB Layout Guidelines for details).
23
24
25
SGND
FB
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated
to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
Power Good (Output): Open Drain Output. The PG pin is externally tied with a resistor to VDD. A high
output is asserted when VOUT > 92% of nominal.
PG
Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high =
enable, logic low = shutdown. In the off state, supply current of the device is greatly reduced (typically
5µA). The EN pin should not be left open.
26
27
EN
VIN
Power Supply Voltage (Input): Requires bypass capacitor to SGND.
5V Internal Linear Regulator (Output): VDD supply is the power MOSFET gate drive supply voltage
and the supply bus for the IC. VDD is created by internal LDO from VIN. When VIN < +5.5V, VDD
should be tied to PVIN pins. A 1.0µF ceramic capacitor from the VDD pin to PGND pins must be place
next to the IC.
28
VDD
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July 2011
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Micrel, Inc.
MIC26601
Absolute Maximum Ratings(1, 2)
Operating Ratings(3)
Supply Voltage (PVIN, VIN) .................................4.5V to 28V
PVDD, VDD Supply Voltage (PVDD, VDD)..........4.5V to 5.5V
Enable Input (VEN).................................................. 0V to VIN
Junction Temperature (TJ)........................ −40°C to +125°C
Maximum Power Dissipation .....................................Note 4
Package Thermal Resistance(4)
PVIN to PGND................................................ −0.3V to +29V
VIN to PGND ....................................................−0.3V to PVIN
PVDD, VDD to PGND......................................... −0.3V to +6V
VSW, VCS to PGND..............................−0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND.................................................. −0.3V to 35V
VFB, VPG to PGND...............................−0.3V to (VDD + 0.3V)
VEN to PGND ........................................ −0.3V to (VIN +0.3V)
PGND to SGND ........................................... −0.3V to +0.3V
Junction Temperature ..............................................+150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
5mm x 6mm MLF®(θJA) .....................................28°C/W
Electrical Characteristics(5)
PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Power Supply Input
Input Voltage Range (VIN, PVIN)
Quiescent Supply Current
Shutdown Supply Current
VDD Supply Voltage
4.5
28
1500
10
V
VFB = 1.5V (non-switching)
VEN = 0V
730
5
µA
µA
VDD Output Voltage
VIN = 7V to 28V, IDD = 40mA
VDD Rising
4.8
3.7
5
5.4
4.5
V
V
VDD UVLO Threshold
4.2
400
380
VDD UVLO Hysteresis
mV
mV
Dropout Voltage (VIN – VDD
)
IDD = 25mA
600
5.5
DC/DC Controller
Output-Voltage Adjust Range (VOUT
)
0.8
V
Reference
0.792
0.788
0.8
0.8
0.808
0.812
0°C ≤ TJ ≤ 85°C (±1.0%)
−40°C ≤ TJ ≤ 125°C (±1.5%)
IOUT = 0A to 6A (continuous mode)
VIN = 4.5V to 28V
V
Load Regulation
Line Regulation
0.25
0.25
50
%
%
FB Bias Current
VFB = 0.8V
nA
Enable Control
EN Logic Level High
EN Logic Level Low
EN Bias Current
1.8
V
V
0.6
30
VEN = 12V
6
µA
Oscillator
Switching Frequency(6)
Maximum Duty Cycle(7)
Minimum Duty Cycle
Minimum Off-Time
VFB = 0V
450
600
82
750
kHz
%
VFB = 1.0V
0
%
300
ns
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Micrel, Inc.
MIC26601
Electrical Characteristics(5) (Continued)
PVIN = VIN = VEN = 12V, VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Soft-Start
Soft-Start Time
5
ms
Short-Circuit Protection
Current-Limit Threshold
Current-Limit Threshold
Short-Circuit Current
Internal FETs
7.5
6.6
13
13
17
17
A
A
A
V
V
FB = 0.8V, TJ = 25°C
FB = 0.8V, TJ = 125°C
VFB = 0V
2.7
Top-MOSFET RDS (ON)
Bottom-MOSFET RDS (ON)
SW Leakage Current
VIN Leakage Current
Power Good (PG)
ISW = 1A
ISW = 1A
VEN = 0V
VEN = 0V
42
mΩ
mΩ
µA
12.5
60
25
µA
PG Threshold Voltage
PG Hysteresis
Sweep VFB from Low to High
Sweep VFB from High to Low
Sweep VFB from Low to High
Sweep VFB < 0.9 × VNOM, IPG = 1mA
85
92
5.5
100
70
95
%VOUT
%VOUT
µs
PG Delay Time
PG Low Voltage
200
mV
Thermal Protection
Over-Temperature Shutdown
TJ Rising
160
15
°C
°C
Over-Temperature Shutdown
Hysteresis
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. A 5 square inch 4 layer, 0.62”, FR-4 PCB with 2oz finish copper weight
per layer is used for the θJA.
5. Specification for packaged product only.
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 300ns.
M9999-071311-A
July 2011
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Micrel, Inc.
MIC26601
Typical Characteristics
VIN Shutdown Current
vs. Input Voltage
VDD Output Voltage
vs. Input Voltage
VIN Operating Supply Current
vs. Input Voltage
20
10
8
60
45
30
15
0
VEN = 0V
VOUT = 1.8V
REN = OPEN
16
IOUT = 0A
SWITCHING
12
8
6
4
VFB = 0.9V
IDD = 10mA
4
2
0
0
4
10
16
22
28
4
10
16
22
28
4
10
16
22
28
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Feedback Voltage
vs. Input Voltage
Current Limit
vs. Input Voltage
Total Regulation
vs. Input Voltage
0.808
0.804
0.800
0.796
0.792
1.0%
0.5%
0.0%
-0.5%
-1.0%
20
15
10
5
VOUT = 1.8V
IOUT = 0A to 6A
VOUT = 1.8V
OUT = 0A
VOUT = 1.8V
I
0
4
10
16
22
28
4
10
16
22
28
4
10
16
22
28
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Switching Frequency
vs. Input Voltage
Enable Input Current
vs. Input Voltage
PG/VREF Ratio
vs. Input Voltage
700
650
600
550
500
16
12
8
100%
95%
90%
85%
80%
VEN = VIN
VOUT = 1.8V
OUT = 0A
I
4
VREF = 0.7V
0
4
4
10
16
22
28
10
16
22
28
4.0
10.0
16.0
22.0
28.0
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
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July 2011
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Micrel, Inc.
MIC26601
Typical Characteristics (Continued)
VIN Operating Supply Current
vs. Temperature
VIN Shutdown Current
vs. Temperature
VDD UVLO Threshold
vs. Temperature
20
16
12
8
20
15
10
5
5
4
3
2
1
0
Rising
VIN = 12V
IOUT = 0A
VEN = 0V
Falling
VIN = 12V
OUT = 1.8V
V
IOUT = 0A
4
SWITCHING
Hyst
0
0
-50
-25
0
25
50
75
100
125
125
125
-50
-25
0
25
50
75
100
100
100
125
125
125
-50
-25
0
25
50
75
100
125
125
125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
Load Regulation
vs. Temperature
Line Regulation
vs. Temperature
Feedback Voltage
vs. Temperature
0.808
0.804
0.800
0.796
0.792
0.4%
0.2%
0.0%
-0.2%
-0.4%
0.2%
0.1%
0.0%
-0.1%
-0.2%
VIN = 12V
VOUT = 1.8V
I
OUT = 0A
VIN = 4.5V to 28V
V
OUT = 1.8V
IOUT = 0A
VIN = 12V
V
OUT = 1.8V
IOUT =0A to 6A
-50
-25
0
25
50
75
-50
-25
0
25
50
75
100
-50
-25
0
25
50
75
100
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
Current Limit
vs. Temperature
Switching Frequency
vs. Temperature
VDD
vs. Temperature
700
650
600
550
500
6
5
4
3
25
20
15
10
5
VIN = 12V
OUT = 1.8V
OUT = 0A
V
I
VIN = 12V
VOUT = 1.8V
VIN = 12V
IOUT = 0A
2
0
-50
-25
0
25
50
75
100
-50
-25
0
25
50
75
-50
-25
0
25
50
75
100
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
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July 2011
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Micrel, Inc.
MIC26601
Typical Characteristics (Continued)
Feedback Voltage
vs. Output Current
Efficiency
vs. Output Current
Output Voltage
vs. Output Current
100
90
80
70
60
50
0.808
0.804
0.800
0.796
0.792
1.819
1.814
1.810
1.805
1.800
1.796
1.791
1.787
1.782
12VIN
VIN = 12V
VOUT = 1.8V
24VIN
VIN = 12V
OUT = 1.8V
V
VOUT = 1.8V
0
1
2
3
4
5
6
6
8
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Switching Frequency
vs. Output Current
Output Voltage (VIN = 5V)
vs. Output Current
Line Regulation
vs. Output Current
1.0%
0.5%
0.0%
-0.5%
-1.0%
700
650
600
550
500
5
4.6
4.2
3.8
3.4
VIN = 5V
VIN = 4.5V to 28V
OUT = 1.8V
VFB < 0.8V
V
TA
VIN = 12V
25ºC
85ºC
125ºC
V
OUT = 1.8V
3
0
1
2
3
4
5
6
7
8
0
1
2
3
4
5
0
1
2
3
4
5
6
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Die Temperature* (VIN = 5V)
vs. Output Current
Efficiency (VIN = 5V)
vs. Output Current
IC Power Dissipation (VIN = 5V)
vs. Output Current
100
95
90
85
80
75
70
65
60
55
60
50
40
30
20
10
2.5
2.0
1.5
1.0
0.5
0.0
V
IN = 5V
V
OUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
VIN = 5V
3.3V
V
OUT = 1.8V
0.8V
5
0
0
50
0
0
1
2
3
4
6
1
2
3
4
5
6
1
2
3
4
5
6
7
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
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July 2011
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Micrel, Inc.
MIC26601
Typical Characteristics (Continued)
Efficiency (VIN = 12V)
vs. Output Current
Die Temperature* (VIN = 12V)
vs. Output Current
IC Power Dissipation (VIN = 12V)
vs. Output Current
100
95
90
85
80
75
70
65
60
55
50
60
50
40
30
20
10
0
2.5
2.0
1.5
1.0
0.5
0.0
VIN = 12V
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
VOUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V, 5.0V
5.0V
VIN = 12V
OUT = 1.8V
0.8V
V
0
1
2
3
4
5
6
0
1
2
3
4
5
6
0
1
2
3
4
5
6
7
8
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
IC Power Dissipation (VIN = 24V)
vs. Output Current
Die Temperature* (VIN = 24V)
vs. Output Current
Efficiency (VIN = 24V)
vs. Output Current
2.5
2.0
1.5
1.0
0.5
0.0
VIN = 24V
95
90
85
80
75
70
65
60
55
50
60
50
40
30
20
10
0
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
VOUT = 0.8,V, 1.0V, 1.2V, 1.5V, 1.8V,2.5V, 3.3V, 5.0V
5.0V
0.8V
VIN = 24V
VOUT = 1.8V
0
1
2
3
4
5
6
0
1
2
3
4
5
6
0
1
2
3
4
5
6
7
8
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Thermal Derating*
Thermal Derating*
Thermal Derating*
vs. Ambient Temperature
vs. Ambient Temperature
vs. Ambient Temperature
12
10
8
12
12
10
8
10
8
0.8V
1.8V
0.8V
1.8V
6
1.5V
6
6
3.3V
4
4
4
VIN = 12V
OUT = 0.8, 1.2, 1.8V
VIN = 5V
OUT = 0.8, 1.2, 1.5V
VIN = 5V
V
V
VOUT = 1.8, 2.5, 3.3V
2
2
2
0
0
0
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
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Micrel, Inc.
MIC26601
Typical Characteristics (Continued)
Thermal Derating*
Thermal Derating*
vs. Ambient Temperature
vs. Ambient Temperature
12
10
8
12
10
8
2.5V
0.8V
5V
6
6
2.5V
4
4
VIN = 24V
VIN = 12V
OUT = 2.5, 3.3, 5V
V
OUT = 0.8, 1.2, 2.5V
V
2
2
0
0
-50
-25
0
25
50
75
100
125
-50
-25
0
25
50
75
100
125
AMBIENT TEMPERATURE (°C)
AMBIENT TEMPERATURE (°C)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26601 case mounted on a 5 square inch 4 layer, 0.62”,
FR-4 PCB with 2oz finish copper weight per layer, see Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient
temperature and proximity to other heat emitting components.
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July 2011
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Micrel, Inc.
MIC26601
Functional Characteristics
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MIC26601
Functional Characteristics (Continued)
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Micrel, Inc.
MIC26601
Functional Characteristics (Continued)
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Micrel, Inc.
MIC26601
Functional Diagram
Figure 1. MIC26601 Block Diagram
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Micrel, Inc.
MIC26601
The maximum duty cycle is obtained from the 300ns
Functional Description
tOFF(min)
:
The MIC26601 is an adaptive ON-time synchronous
step-down DC-DC regulator with an internal 5V linear
regulator and a Power Good (PG) output. It is designed
to operate over a wide input voltage range from 4.5V to
28V and provides a regulated output voltage at up to 7A
of output current. An adaptive ON-time control scheme is
employed in to obtain a constant switching frequency
and to simplify the control compensation. Over-current
protection is implemented without the use of an external
sense resistor. The device includes an internal soft-start
function which reduces the power supply input surge
current at start-up by controlling the output voltage rise
time.
tS − tOFF(MIN)
300ns
tS
DMAX
=
= 1−
Eq. 2
tS
where tS = 1/600kHz = 1.66μs.
It is not recommended to use MIC26601 with a OFF-time
close to tOFF(min) during steady-state operation. Also, as
VOUT increases, the internal ripple injection will increase
and reduce the line regulation performance. Therefore,
the maximum output voltage of the MIC26601 should be
limited to 5.5V and the maximum external ripple injection
should be limited to 200mV. Please refer to “Setting
Output Voltage” subsection in Application Information for
more details.
Theory of Operation
The MIC26601 operates in a continuous mode as shown
in Figure 1.
The actual ON-time and resulting switching frequency
will vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 24V to 1.0V. The minimum tON
measured on the MIC26601 evaluation board is about
100ns. During load transients, the switching frequency is
changed due to the varying OFF-time.
Continuous Mode
In continuous mode, the output voltage is sensed by the
MIC26601 feedback pin FB via the voltage divider R1
and R2, and compared to a 0.8V reference voltage VREF
at the error comparator through
a
low gain
transconductance (gm) amplifier. If the feedback voltage
decreases and the output of the gm amplifier is below
0.8V, then the error comparator will trigger the control
logic and generate an ON-time period. The ON-time
period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios.
Figure 2 shows the MIC26601 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ON-
time is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
VOUT
tON(ESTIMATED)
=
Eq. 1
VIN × 600kHz
where VOUT is the output voltage and VIN is the power
stage input voltage.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
300ns, the MIC26601 control logic will apply the tOFF(min)
instead. tOFF(min) is required to maintain enough energy in
the boost capacitor (CBST) to drive the high-side
MOSFET.
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Micrel, Inc.
MIC26601
In order to meet the stability requirements, the
MIC26601 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV. If a low-ESR output capacitor is selected,
then the feedback voltage ripple may be too small to be
sensed by the gm amplifier and the error comparator.
Also, the output voltage ripple and the feedback voltage
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. In these cases, ripple injection is required to ensure
proper operation. Please refer to “Ripple Injection”
subsection in Application Information for more details
about the ripple injection technique.
Figure 2. MIC26601 Control Loop Timing
Figure 3 shows the operation of the MIC26601 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC26601 converter.
VDD Regulator
The MIC26601 provides a 5V regulated output for input
voltage VIN ranging from 5.5V to 28V. When VIN < 5.5V,
VDD should be tied to PVIN pins to bypass the internal
linear regulator.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC26601 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 6ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a stair-
case VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
Current Limit
The MIC26601 uses the RDS(ON) of the internal low-side
power MOSFET to sense over-current conditions. This
method will avoid adding cost, board space and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the
high-side MOSFET.
Figure 3. MIC26601 Load Transient Response
Unlike true current-mode control, the MIC26601 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC26601 control loop has the advantage
of eliminating the need for slope compensation.
In each switching cycle of the MIC26601 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the peak inductor current
is greater than 13A, then the MIC26601 turns off the
high-side MOSFET and a soft-start sequence is
triggered. This mode of operation is called “hiccup
mode” and its purpose is to protect the downstream load
in case of a hard short. The load current-limit threshold
has a fold-back characteristic related to the feedback
voltage as shown in Figure 4.
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Micrel, Inc.
MIC26601
Current Limit Threshold
vs. Feedback Voltage
20
16
12
8
4
0
0.0
0.2
0.4
0.6
0.8
1.0
FEEDBACK VOLTAGE (V)
Figure 4. MIC26601 Current-Limit
Foldback Characteristic
Power Good (PG)
The Power Good (PG) pin is an open drain output which
indicates logic high when the output is nominally 92% of
its steady state voltage. A pull-up resistor of more than
10kꢀ should be connected from PG to VDD.
MOSFET Gate Drive
The block diagram (Figure 1) shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
CBST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged, while the low-side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the high-side MOSFET turns on, the voltage on
the SW pin increases to approximately VIN. Diode D1 is
reverse biased and CBST floats high while continuing to
keep the high-side MOSFET on. The bias current of the
high-side driver is less than 10mA so a 0.1μF to 1μF is
sufficient to hold the gate voltage with minimal droop for
the power stroke (high-side switching) cycle, i.e. ΔBST =
10mA x 1.67μs/0.1μF = 167mV. When the low-side
MOSFET is turned back on, CBST is recharged through
D1. A small resistor RG, which is in series with CBST, can
be used to slow down the turn-on time of the high-side
N-channel MOSFET.
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
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Micrel, Inc.
MIC26601
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high-frequency operation of the MIC26601 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by Equation 7:
Application Information
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated in Equation 3.
2
VOUT × (VIN(MAX) − VOUT
)
P
= IL(RMS × RWINDING
Eq. 7
INDUCTOR(CU)
)
L =
Eq. 3
VIN(MAX) × fSW × 20%×IOUT(MAX)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
where:
fSW = switching frequency, 600kHz
20% = ratio of AC ripple current to DC output current
IN(MAX) = maximum power stage input voltage
PWINDING(Ht) = RWINDING(20°C) × (1+ 0.0042 × (TH − T20°C ))
V
Eq. 8
The peak-to-peak inductor current ripple is:
where:
TH = temperature of wire under full load
T20°C = ambient temperature
VOUT × (VIN(MAX) − VOUT
VIN(MAX) × fSW × L
)
ΔIL(PP)
=
Eq. 4
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
Output Capacitor Selection
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are ceramic, low-ESR aluminum electrolytic, OS-
CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view.
IL(PK) = IOUT(MAX) + 0.5 × ΔIL(PP)
Eq. 5
The RMS inductor current is used to calculate the I2R
losses in the inductor.
2
ΔIL(PP)
2
IL(RMS) = IOUT(MAX)
+
Eq. 6
12
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Micrel, Inc.
MIC26601
The power dissipated in the output capacitor is:
PDISS(C ) = IC × ESRC
The maximum value of ESR is calculated:
Eq. 12
ΔVOUT(PP)
OUT (RMS)
OUT
OUT
ESRC
≤
Eq. 9
OUT
ΔIL(PP)
Input Capacitor Selection
where:
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
ΔVOUT(pp) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated in
Equation 10:
2
ΔIL(PP)
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
2
ΔVOUT(PP)
=
+
(
ΔIL(PP) × ESRC
)
OUT
COUT ×fSW × 8
Eq. 10
ΔVIN = IL(PK) × ESRC
Eq. 13
IN
where:
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
D = duty cycle
COUT = output capacitance value
fSW = switching frequency
As described in the “Theory of Operation” subsection in
Functional Description, the MIC26601 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly.
Also, the output voltage ripple should be in phase with
the inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
IC (RMS) ≈ IOUT(MAX) × D × (1− D)
Eq. 14
IN
The power dissipated in the input capacitor is:
PDISS(C = IC
× ESRC
IN
Eq. 15
)
IN(RMS)
IN
Ripple Injection
The VFB ripple required for proper operation of the
MIC26601 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
then the MIC26601 will lose control and the output
voltage is not regulated. In order to have some amount
of VFB ripple, a ripple injection method is applied for low
output voltage ripple applications.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 11:
ΔIL(PP)
IC
=
Eq. 11
OUT (RMS)
12
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Micrel, Inc.
MIC26601
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1. Enough ripple at the feedback voltage due to the
large ESR of the output capacitors.
As shown in Figure 5a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
Figure 5c. Invisible Ripple at FB
R2
ΔVFB(PP)
=
× ESRC
× ΔIL(PP)
Eq. 16
OUT
R1+ R2
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 5c. The injected ripple
is:
where ΔIL(pp) is the peak-to-peak value of the inductor
current ripple.
2. Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
1
ΔVFB(PP) = VIN × KDIV × D × (1− D)×
Eq. 18
Eq. 19
fSW × τ
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 5b. The typical Cff value is between 1nF and
100nF. With the feedforward capacitor, the feedback
voltage ripple is very close to the output voltage ripple:
R1/R2
KDIV
=
RINJ + R1//R2
where:
ΔVFB(PP) ≈ ESR × ΔIL(PP)
Eq. 17
VIN = Power stage input voltage
D = Duty cycle
fSW = Switching frequency
τ = (R1//R2//RINJ) × Cff
3. Virtually no ripple at the FB pin voltage due to the
very-low ESR of the output capacitors.
In Equations 20 and 21, it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
=
<< 1
Eq. 20
fsw × τ
τ
If the voltage divider resistors R1 and R2 are in the kꢀ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
Figure 5a. Enough Ripple at FB
Figure 5b. Inadequate Ripple at FB
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Micrel, Inc.
MIC26601
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kꢀ range.
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 23.
ΔVFB(PP)
fSW × τ
KDIV
=
×
Eq. 21
Figure 6. Voltage-Divider Configuration
VIN
D × (1− D)
In addition to the external ripple injection added at the
FB pin, internal ripple injection is added at the inverting
input of the comparator inside the MIC26601, as shown
in Figure 7. The inverting input voltage VINJ is clamped to
1.2V. As VOUT is increased, the swing of VINJ will be
clamped. The clamped VINJ reduces the line regulation
because it is reflected as a DC error on the FB terminal.
Therefore, the maximum output voltage of the MIC26601
should be limited to 5.5V to avoid this problem.
Then the value of Rinj is obtained as:
⎛
⎞
1
⎜
⎜
⎟
− 1
RINJ = (R1//R2)×
Eq. 22
⎟
KDIV
⎝
⎠
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
Setting Output Voltage
The MIC26601 requires two resistors to set the output
voltage as shown in Figure 6.
The output voltage is determined by Equation 23:
R1
R2
⎛
⎞
⎟
VOUT = VFB × 1+
Eq. 23
⎜
⎝
⎠
where VFB = 0.8V. A typical value of R1 can be between
3kꢀ and 10kꢀ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small, it will decrease the efficiency of the power supply,
especially at light loads. Once R1 is selected, R2 can be
calculated using:
Figure 7. Internal Ripple Injection
Thermal Measurements
VFB × R1
Measuring the IC’s case temperature is recommended to
insure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
R2 =
Eq. 24
VOUT − VFB
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Micrel, Inc.
MIC26601
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer.
If a thermal couple wire is used, it must be constructed
of 36 gauge wire or higher then (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36)
is adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, an IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
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Micrel, Inc.
MIC26601
PCB Layout Guidelines
Inductor
•
•
•
•
Keep the inductor connection to the switch node
(SW) short.
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
Do not route any digital lines underneath or close to
the inductor.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
Keep the switch node (SW) away from the feedback
(FB) pin.
The following guidelines should be followed to insure
proper operation of the MIC26601 regulator.
The CS pin should be connected directly to the SW
pin to accurate sense the voltage across the low-
side MOSFET.
IC
•
•
To minimize noise, place a ground plane underneath
the inductor.
•
A 2.2µF ceramic capacitor, which is connected to
the PVDD pin, must be located right at the IC. The
PVDD pin is very noise sensitive and placement of
the capacitor is very critical. Use wide traces to
connect to the PVDD and PGND pins.
The inductor can be placed on the opposite side of
the PCB with respect to the IC. It does not matter
whether the IC or inductor is on the top or bottom as
long as there is enough air flow to keep the power
components within their temperature limits. The
input and output capacitors must be placed on the
same side of the board as the IC.
•
A 1.0uF ceramic capacitor must be placed right
between VDD and the signal ground SGND. The
SGND must be connected directly to the ground
planes. Do not route the SGND pin to the PGND
Pad on the top layer.
Output Capacitor
•
•
Place the IC close to the point-of-load (POL).
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
Use fat traces to route the input and output power
lines.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
•
Signal and power grounds should be kept separate
and connected at only one location.
Input Capacitor
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current
load trace can degrade the DC load regulation.
•
•
Place the input capacitor next.
Place the input capacitors on the same side of the
board and as close to the IC as possible.
•
•
•
•
Keep both the PVIN pin and PGND connections
short.
Optional RC Snubber
Place several vias to the ground plane close to the
input capacitor ground terminal.
•
Place the RC snubber on either side of the board
and as close to the SW pin as possible.
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
•
•
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
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Micrel, Inc.
MIC26601
Evaluation Board Schematic
Figure 8. Schematic of MIC26601 Evaluation Board
(J11, R13, R15 are for testing purposes)
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Micrel, Inc.
MIC26601
Bill of Materials
Item
Part Number
Manufacturer
Description
Qty.
C1
Open
12105C475KAZ2A
GRM32ER71H475KA88L
C3225X7R1H475K
Open
AVX(1)
Murata(2)
TDK(3)
C2, C3
4.7µF Ceramic Capacitor, X7R, Size 1210, 50V
2
C4, C13, C15
C5
12106D107MAT2A
GRM32ER60J107ME20L
C3225X5R0J107M
06035C104KAT2A
GRM188R71H104KA93D
C1608X7R1H104K
0603ZC105KAT2A
GRM188R71A105KA61D
C1608X7R1A105K
0603ZD225KAT2A
GRM188R61A225KE34D
C1608X5R1A225K
06035C472KAZ2A
GRM188R71H472K
C1608X7R1H472K
B41851F7227M
AVX(1)
Murata(2)
TDK(3)
AVX(1)
Murata(2)
TDK(3)
AVX(1)
Murata(2)
TDK(3)
AVX(1)
Murata(2)
TDK(3)
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
1.0µF Ceramic Capacitor, X7R, Size 0603, 10V
2.2µF Ceramic Capacitor, X5R, Size 0603, 10V
1
3
1
1
C6, C7, C10
C8
C9
AVX(1)
Murata(2)
TDK(3)
C12
4.7nF Ceramic Capacitor, X7R, Size 0603, 50V
220µF Aluminum Capacitor, 35V
1
1
C14
EPCOS(4)
C11, C16
Open
SD103AWS
MCC(5)
Diodes Inc(6)
Vishay(7)
D1
40V, 350mA, Schottky Diode, SOD323
1
SD103AWS-7
SD103AWS
Cooper Bussmann(8) 2.2µH Inductor, 15A Saturation Current
1
1
1
1
1
1
3
1
1
1
1
1
1
L1
HCF1305-2R2-R
R1
CRCW06032R21FKEA
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
2.21ꢀ Resistor, Size 0603, 1%
2.00ꢀ Resistor, Size 0603, 1%
19.6kꢀ Resistor, Size 0603, 1%
2.49kꢀ Resistor, Size 0603, 1%
20.0kꢀ Resistor, Size 0603, 1%
10.0kꢀ Resistor, Size 0603, 1%
4.99kꢀ Resistor, Size 0603, 1%
2.87kꢀ Resistor, Size 0603, 1%
2.00kꢀ Resistor, Size 0603, 1%
1.18kꢀ Resistor, Size 0603, 1%
806ꢀ Resistor, Size 0603, 1%
475ꢀ Resistor, Size 0603, 1%
R2
CRCW06032R00FKEA
CRCW060319K6FKEA
CRCW06032K49FKEA
CRCW060320K0FKEA
CRCW060310K0FKEA
CRCW06034K99FKEA
CRCW06032K87FKEA
CRCW06032K006FKEA
CRCW06031K18FKEA
CRCW0603806RFKEA
CRCW0603475RFKEA
R3
R4
R5
R6, R14, R17
R7
R8
R9
R10
R11
R12
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Micrel, Inc.
MIC26601
Bill of Materials (Continued)
Item
Part Number
Manufacturer
Vishay Dale(7)
Vishay Dale(7)
Vishay Dale(7)
Description
Qty.
R13
CRCW06030000FKEA
CRCW060349R9FKEA
CRCW06031R21FKEA
Open
0ꢀ Resistor, Size 0603, 5%
49.9ꢀ Resistor, Size 0603, 1%
1.21ꢀ Resistor, Size 0603, 1%
1
1
2
R15
R16, R18
R20
28V, 6A Hyper Speed Control™ Synchronous
DC/DC Buck Regulator
U1
MIC26601YJL
Micrel. Inc.(9)
1
Notes:
1. AVX: www.avx.com.
2. Murata: www.murata.com.
3. TDK: www.tdk.com.
4. EPCOS: www.epcos.com.
5. SANYO: www.sanyo.com.
6. Diode Inc.: www.diodes.com.
7. Vishay: www.vishay.com.
8. Cooper Bussmann: www.cooperbussmann.com.
9. Micrel, Inc.: www.micrel.com.
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Micrel, Inc.
MIC26601
PCB Layout Recommendations
Figure 9. MIC26601 Evaluation Board Top Layer
Figure 10. MIC26601 Evaluation Board Mid-Layer 1 (Ground Plane)
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Micrel, Inc.
MIC26601
PCB Layout Recommendations (Continued)
Figure 11. MIC26601 Evaluation Board Mid-Layer 2
Figure 12. MIC26601 Evaluation Board Bottom Layer
M9999-071311-A
July 2011
28
Micrel, Inc.
MIC26601
Recommended Land and Solder Stencil Pattern
M9999-071311-A
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Micrel, Inc.
MIC26601
Package Information
28-Pin 5mm x 6mm MLF® (YJL)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2011 Micrel, Incorporated.
M9999-071311-A
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