MIC2171WU-TR [MICROCHIP]
5.5A SWITCHING REGULATOR, 115kHz SWITCHING FREQ-MAX, PSSO5;
| 型号： | MIC2171WU-TR |
| 厂家： | 
MICROCHIP |
| 描述： | 5.5A SWITCHING REGULATOR, 115kHz SWITCHING FREQ-MAX, PSSO5 开关 |
| 文件： | 总12页 (文件大小：187K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC2171
100kHz 2.5A Switching Regulator
General Description
Features
The MIC2171 is a complete 100kHz SMPS current-mode
controller with an internal 65V 2.5A power switch.
• 2.5A, 65V internal switch rating
• 3V to 40V input voltage range
Although primarily intended for voltage step-up applica-
tions, the floating switch architecture of the MIC2171
makes it practical for step-down, inverting, and Cuk config-
urations as well as isolated topologies.
• Current-mode operation, 2.5A peak
• Internal cycle-by-cycle current limit
• Thermal shutdown
• Twice the frequency of the LM2577
• Low external parts count
• Operates in most switching topologies
• 7mA quiescent current (operating)
• Fits LT1171/LM2577 TO-220 and TO-263 sockets
Operating from 3V to 40V, the MIC2171 draws only 7mA
of quiescent current, making it attractive for battery
operated supplies.
The MIC2171 is available in a 5-pin TO-220 or TO-263 for
–40°C to +85°C operation.
Data sheets and support documentation can be found on
Micrel’s web site at: www.micrel.com.
Applications
• Laptop/palmtop computers
• Battery operated equipment
• Hand-held instruments
• Off-line converter up to 50W(requires external power
switch)
• Pre-driver for higher power capability
___________________________________________________________________________________________________________
Typical Application
+5V
VOUT
5V, 0.5A
VIN
4V to 6V
(4.75V min.)
T1
C1*
L1
15µH
D2
1N5818
C1
R4*
C3*
47µF
47µF
R1
VOUT
+12V, 0.25A
C4
470µF
3.74k
1%
D1
IN
D1*
SW
IN
R1
10.7k
1%
1N5822
SW
MIC2171
COMP
1:1.25
LPRI= 12µH
MIC2171
FB
R2
R3
1k
C2
470µF
GND
COMP
GND
FB
1.24k
R2
1%
R3
1k
C3
1µF
1.24k
1%
C2
1µF
* Locate near MIC2171 when supply leads > 2”
* Optional voltage clipper (may be req’d if T1 leakage inductance too high)
Figure 1. MIC2171 5V to 12V Boost Converter
Figure 2. MIC2171 5VFlyback Converter
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
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MIC2171
Ordering Information
Part Number
Temperature Range
–40° to +85°C
Package
Standard
MIC2171BT
MIC2171BU
RoHS Compliant*
MIC2171WT
MIC2171WU
5-Pin TO-220
5-Pin TO-263
–40° to +85°C
*RoHS compliant with "high-melting solder" exemption.
Pin Configuration
5 IN
5 IN
4 SW
4 SW
3 GND
2 FB
3 GND
2 FB
1 COMP
1 COMP
Tab GND
Tab GND
5-Pin TO-220 (T)
5-Pin TO-263 (U)
Pin Description
Pin Number
Pin Name
Pin Function
1
COMP
Frequency Compensation: Output of transconductance-type error amplifier.
Primary function is for loop stabilization. Can also be used for output voltage
soft-start and current limit tailoring.
2
3
4
5
FB
GND
SW
IN
Feedback: Inverting input of error amplifier. Connect to external resistive divider
to set power supply output voltage.
Ground: Connect directly to the input filter capacitor for proper operation (see
applications info).
Power Switch Collector: Collector of NPN switch. Connect to external inductor
or input voltage depending on circuit topology.
Supply Voltage: 3.0V to 40V
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Absolute Maximum Ratings
Operating Ratings
Supply Voltage (VIN).......................................................40V
Switch Voltage (VSW)......................................................65V
Feedback Voltage (transient, 1ms) (VFB) .....................±15V
Lead Temperature (soldering, 10 sec.)...................... 300°C
Storage Temperature (Ts) .........................–65°C to +150°C
ESD Rating(1)
Operating Temperature Range...................–40°C to +85°C
Junction Temperature (TJ) ........................–55°C to +150°C
Thermal Resistance
TO-220-5 (θJA) (2) ...............................................45°C/W
TO-263-5 (θJA) (3)................................................45°C/W
Electrical Characteristics
VIN = 5V; TA = 25°C, bold values indicate –40°C< TA < +85°C, unless noted.
Parameter
Condition
Min
Typ
Max
Units
Reference Section
Feedback Voltage (VFB)
VCOMP = 1.24V
1.220
1.214
1.240
0.6
1.264
1.274
V
V
Feedback Voltage
Line Regulation
3V ≤ VIN ≤ 40V
VCOMP = 1.24V
%/V
Feedback Bias Current (IFB)
VFB = 1.24V
310
750
1100
nA
nA
Error Amplifier Section
Transconductance (gm)
∆ICOMP = ±25µA
3.0
2.4
3.9
6.0
7.0
µA/mV
µA/mV
Voltage Gain (AV)
Output Current
0.9V ≤ VCOMP ≤ 1.4V
400
800
175
2000
V/V
VCOMP = 1.5V
125
100
350
400
µA
µA
Output Swing
High Clamp, VFB = 1V
Low Clamp, VFB = 1.5V
1.8
0.25
2.1
0.35
2.3
0.52
V
V
Compensation Pin Threshold
Duty Cycle = 0
0.8
0.6
0.9
1.08
1.25
V
V
Output Switch Section
ON Resistance
ISW = 2A, VFB = 0.8V
0.37
0.50
0.55
Ω
Ω
Current Limit
Duty Cycle = 50%, TJ ≥ 25°C
Duty Cycle = 50%, TJ < 25°C
Duty Cycle = 80%, Note 4
2.5
2.5
2.5
3.6
4.0
3.0
5.0
5.5
5.0
A
A
A
Breakdown Voltage (BV)
3V ≤ VIN ≤ 40V
65
75
V
ISW = 5mA
Oscillator Section
Frequency (fO)
88
85
100
90
112
115
kHz
kHz
80
95
%
Duty Cycle [δ(max)]
Input Supply Voltage Section
Minimum Operating Voltage
Quiescent Current (IQ)
2.7
7
3.0
9
V
3V ≤ VIN ≤ 40V, VCOMP = 0.6V, ISW = 0
mA
mA
Supply Current Increase (∆IIN)
∆ISW = 2A, VCOMP = 1.5V, during switch on-time
9
20
Notes:
1. Devices are ESD sensitive. Handling precautions recommended.
2. Mounted vertically, no external heat sink, 1/4 inch leads soldered to PC board containing approximately 4 inch squared copper area surrounding
leads.
3. All ground leads soldered to approximately 2 inches squared of horizontal PC board copper area.
4. For duty cycles (δ) between 50% and 95%, minimum guaranteed switch current is ICL = 1.66 (2-δ) Amp (Pk).
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Typical Characteristics
Feedback Voltage
Line Regulation
Minimum
Operating Voltage
Feedback Bias Current
5
4
2.9
800
700
600
500
400
300
200
100
0
2.8
2.7
T
= 125°C
3
J
2
1
0
2.6
T
= 25°C
J
Switch Current = 2A
2.5
-1
-2
-3
-4
-5
2.4
2.3
T
= -40°C
20
J
0
10
30
40
-100 -50
0
50
100 150
-100 -50
0
50
100 150
VIN Operating (V)
Temperature (°C)
Temperature (°C)
Supply Current
Supply Current
Supply Current
15
14
13
12
11
10
9
50
40
30
20
10
0
10
9
8
7
6
5
4
3
2
1
0
I
= 0
VCOMP = 0.6V
S W
D.C.= 90%
δ = 90%
D.C.= 50%
D.C.= 0%
8
δ = 50%
7
6
5
0
10
20
30
40
-100 -50
0
50
100 150
0
1
2
3
4
Temperature(°C)
VIN Operating Voltage (V)
Switch Current (A)
Switch On-Voltage
Oscillator Frequency
Current Limit
1.6
8
6
120
110
100
90
1.4
1.2
1.0
0.8
0.6
0.4
0.2
0
T
= 25°C
J
T
= –40°C
J
25°C
–40°C
4
2
0
80
125°C
T
= 125°C
J
70
60
-50
0
50
100
150
0
20
40
60
80
100
0
1
2
3
Temperature(°C)
Duty Cycle (%)
Switch Current (A)
Error Amplifier Gain
Error Amplifier Gain
Error Amplifier Phase
5.0
4.5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
7000
6000
5000
4000
3000
2000
1000
0
-30
0
30
60
90
120
150
180
210
-100 -50
0
50
100 150
1
10
100
1000 10000
1
10
100
1000 10000
Temperature(°C)
Frequency (kHz)
Frequency (kHz)
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MIC2171
Functional Diagram
D1
IN
2.3V
SW
Reg.
Anti-Sat.
Driver
100kHz
Osc.
Logic
Q1
Com-
parator
FB
Current
Amp.
Error
Amp.
1.24V
Ref.
COMP
GND
reliability and provides automatic output current limiting.
Finally, current-mode operation provides automatic input
voltage feed forward which prevents instantaneous input
voltage changes from disturbing the output voltage
setting.
Functional Description
Refer to “Block Diagram MIC2171”.
Internal Power
The MIC2171 operates when VIN is ≥ 2.6V. An internal
2.3V regulator supplies biasing to all internal circuitry
including a precision 1.24V band gap reference.
Anti-Saturation
The anti-saturation diode (D1) increases the usable duty
cycle range of the MIC2171 by eliminating the base to
collector stored charge which would delay Q1’s turnoff.
PWM Operation
The 100kHz oscillator generates a signal with a duty
cycle of approximately 90%. The current-mode
comparator output is used to reduce the duty cycle when
the current amplifier output voltage exceeds the error
amplifier output voltage. The resulting PWM signal
controls a driver which supplies base current to output
transistor Q1.
Compensation
Loop stability compensation of the MIC2171 can be
accomplished by connecting an appropriate network
from either COMP to circuit ground (see “Typical
Applications”) or COMP to FB.
The error amplifier output (COMP) is also useful for soft
start and current limiting. Because the error amplifier
output is a transconductance type, the output impedance
is relatively high which means the output voltage can be
easily clamped or adjusted externally.
Current-Mode Advantages
The MIC2171 operates in current mode rather than
voltage mode. There are three distinct advantages to
this technique. Feedback loop compensation is greatly
simplified because inductor current sensing removes a
pole from the closed loop response. Inherent cycle-by-
cycle current limiting greatly improves the power switch
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MIC2171
the losses of the power switch driver circuitry. The dc
losses are calculated from the supply voltage (VIN) and
device supply current (IQ).The MIC2171 supply current is
almost constant regardless of the supply voltage (see
“Electrical Characteristics”). The driver section losses
(not including the switch) are a function of supply
voltage, power switch current, and duty cycle.
Application Information
Soft Start
A diode-coupled capacitor from COMP to circuit ground
slows the output voltage rise at turn on (Figure 3).
VIN
IN
P(bias+driver) = (VIN IQ) + (VIN(min) x ISW x ∆IIN)
where:
MIC2171
P
(bias+driver) = device operating losses
IN(min) = supply voltage = VIN – VSW
COMP
V
IQ = typical quiescent supply current
CL = power switch current limit
∆IIN = typical supply current increase
D1
D2
C1
R1
C2
I
As a practical example refer to Figure 1.
Figure 3. Soft Start
VIN = 5.0V
IQ = 0.007A
The additional time it takes for the error amplifier to
charge the capacitor corresponds to the time it takes the
output to reach regulation. Diode D1 discharges C1
when VIN is removed.
ICL = 2.21A
δ = 66.2% (0.662)
then:
Current Limit
VIN(min) = 5.0V – (2.21 x 0.37) = 4.18V
P(bias+driver) = (5 x 0.007) + (4.18 x 2.21 x 0.009)
P(bias+driver) = 0.1W
VIN
IN
SW
MIC2171
Power switch dissipation calculations are greatly
simplified by making two assumptions which are usually
fairly accurate. First, the majority of losses in the power
switch are due to on-losses. To find these losses, assign
a resistance value to the collector/emitter terminals of
the device using the saturation voltage versus collector
current curves (see Typical Performance Character-
istics). Power switch losses are calculated by modeling
the switch as a resistor with the switch duty cycle
modifying the average power dissipation.
VOUT
FB
COMP
GND
R1
C1
R2
ICL 0.6V/R2
R3
Q1
Note: Input and output
returns not common
C2
Figure 4. Current Limit
P
SW = (ISW)2 RSW
δ
The maximum current limit of the MIC2171 can be
reduced by adding a voltage clamp to the COMP output
(Figure 4). This feature can be useful in applications
requiring either a complete shutdown of Q1’s switching
action or a form of current fold-back limiting. This use of
the COMP output does not disable the oscillator,
amplifiers or other circuitry, therefore, the supply current
is never less than approximately 5mA.
where:
δ = duty cycle
VOUT + VF − VIN(min)
VOUT + VF
δ =
VSW = ICL (RSW
)
VOUT = output voltage
Thermal Management
VF = D1 forward voltage drop at IOUT
Although the MIC2171 family contains thermal protection
circuitry, for best reliability, avoid prolonged operation
with junction temperatures near the rated maximum.
From the Typical performance Characteristics:
RSW = 0.37ꢀ
then:
The junction temperature is determined by first
calculating the power dissipation of the device. For the
MIC2171, the total power dissipation is the sum of the
device operating losses and power switch losses.
PSW = (2.21)2 × 0.37 × 0.662
P
SW = 1.2W
P(total) = 1.2 + 0.1
P(total) = 1.3W
The device operating losses are the dc losses
associated with biasing all of the internal functions plus
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MIC2171
The junction temperature for any semiconductor is
calculated using the following:
Discontinuous mode is preferred because the feedback
control of the converter is simpler.
TJ = TA + P(total) θJA
where:
When L1 discharges its current completely during the
MIC2171 off-time, it is operating in discontinuous mode.
L1 is operating in continuous mode if it does not
discharge completely before the MIC2171 power switch
is turned on again.
TJ = junction temperature
TA = ambient temperature (maximum)
P(total) = total power dissipation
θJA = junction to ambient thermal resistance
For the practical example:
TA = 70°C
Discontinuous Mode Design
Given the maximum output current, solve equation (1) to
determine whether the device can operate in
discontinuous mode without initiating the internal device
current limit.
θJA = 45°C/W (TO-220)
then:
I
⎛
⎞
CL
⎜
⎜
⎟
⎟
VIN(min)
δ
TJ = 70 + (1.24 × 45)
TJ = 126°C
2
⎝
⎠
(1)
IOUT ≤
VOUT
This junction temperature is below the rated maximum of
150°C.
VOUT + VF − VIN(min)
VOUT + VF
(1a)
δ =
Grounding
where:
Refer to Figure 5. Heavy lines indicate high current
paths.
ICL = internal switch current limit
ICL = 2.5A when δ < 50%
VIN
ICL = 1.67 (2 – δ) when δ ≥ 50%
IN
SW
(Refer to Electrical Characteristics.)
IOUT = maximum output current
VIN(min) = minimum input voltage = VIN – VSW
δ = duty cycle
MIC2171
FB
VC
GND
VOUT = required output voltage
VF = D1 forward voltage drop
For the example in Figure 1.
IOUT = 0.25A
Single point ground
Figure 5. Single Point Ground
ICL = 1.67 (2–0.662) = 2.24A
A single point ground is strongly recommended for
proper operation.
VIN(min) = 4.18V
δ = 0.662
The signal ground, compensation network ground, and
feed-back network connections are sensitive to minor
voltage variations. The input and output capacitor
grounds and power ground conductors will exhibit
voltage drop when carrying large currents. Keep the
sensitive circuit ground traces separate from the power
ground traces. Small voltage variations applied to the
sensitive circuits can prevent the MIC2171 or any
switching regulator from functioning properly.
V
OUT = 12.0V
VF = 0.36V (@ .26A, 70°C)
then:
2.235
2
⎛
⎜
⎞
⎟
× 4.178 × 0.662
⎝
⎠
IOUT
≤
12
IOUT ≤ 0.258A
This value is greater than the 0.25A output current
requirement, so we can proceed to find the minimum
inductance value of L1 for discontinuous operation at
Boost Conversion
Refer to Figure 1 for a typical boost conversion
application where a +5V logic supply is available but
+12V at 0.25A is required.
POUT
.
2
(VINδ )
The first step in designing a boost converter is
determining whether inductor L1 will cause the converter
to operate in either continuous or discontinuous mode.
(2)
L1 ≥
2POUT SW
f
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where:
MIC2171
solutions to be connected to circuit ground, although a
more conventional technique of connecting the
components from the error amplifier output to its
inverting input is also possible.
POUT = 12 × 0.25 = 3W
SW = 1×105Hz (100kHz)
For our practical example:
4.178 × 0.662
f
Voltage Clipper
2
(
)
L1 ≥
Care must be taken to minimize T1’s leakage
inductance, otherwise it may be necessary to
incorporate the voltage clipper consisting of D1, R4, and
C3 to avoid second breakdown (failure) of the
MIC2171’s internal power switch.
2× 3.0 ×1×105
L1 ≥ 12.4µH (use 15µH)
Equation (3) solves for L1’s maximum current value.
VINTON
(3)
IL1(peak)
=
Discontinuous Mode Design
L1
When designing a discontinuous flyback converter, first
determine whether the device can safely handle the
peak primary current demand placed on it by the output
power. Equation (8) finds the maximum duty cycle
required for a given input voltage and output power. If
the duty cycle is greater than 0.8, discontinuous
operation cannot be used.
where:
T
ON = δ / fSW = 6.62×10-6 sec
4.178 × 6.62 ×10−6
15 ×10−6
IL1(peak)
=
IL1(peak) = 1.84A
Use a 15µH inductor with a peak current rating of at
least 2A.
2POUT
(8)
δ ≥
ICL
(
VIN(min) − VSW
)
Flyback Conversion
For a practical example let: (see Figure 2)
POUT = 5.0V × 0.5A = 2.5W
VIN = 4.0V to 6.0V
Flyback converter topology may be used in low power
applications where voltage isolation is required or
whenever the input voltage can be less than or greater
than the output voltage. As with the step-up converter
the inductor (transformer primary) current can be
continuous or discontinuous. Discontinuous operation is
recommended.
ICL = 2.5A when δ < 50%
1.67 (2 – δ) when δ ≥ 50%
then:
VIN(min) = VIN – (ICL × RSW
V IN(min) = 4 – 0.78V
V IN(min) = 3.22V
Figure 2 shows a practical flyback converter design
using the MIC2171.
Switch Operation
δ ≥ 0.74 (74%), less than 0.8 so discontinuous is
During Q1’s on time (Q1 is the internal NPN transistor—
see block diagrams), energy is stored in T1’s primary
inductance. During Q1’s off time, stored energy is
partially discharged into C4 (output filter capacitor).
Careful selection of a low ESR capacitor for C4 may
provide satisfactory output ripple voltage making
additional filter stages unnecessary.
permitted.
A few iterations of equation (8) may be required if the
duty cycle is found to be greater than 50%.
Calculate the maximum transformer turns ratio a, or
NPRI/NSEC, that will guarantee safe operation of the
MIC2171 power switch.
C1 (input capacitor) may be reduced or eliminated if the
MIC2171 is located near a low impedance voltage
source.
VCEFCE − VIN(max)
(9)
a ≤
VSEC
where:
Output Diode
a = transformer maximum turns ratio
The output diode allows T1 to store energy in its primary
inductance (D2 non-conducting) and release energy into
C4 (D2 conducting). The low forward voltage drop of a
Schottky diode minimizes power loss in D2.
VCE = power switch collector to emitter maximum
voltage
FCE = safety derating factor (0.8 for most
commercial and industrial applications)
Frequency Compensation
VIN(max) = maximum input voltage
A simple frequency compensation network consisting of
R3 and C2 prevents output oscillations.
VSEC = transformer secondary voltage (VOUT
VF)
+
High impedance output stages (transconductance type)
in the MIC2171 often permit simplified loop-stability
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MIC2171
For the practical example:
then:
V
CE = 65V max. for the MIC2171
11.4
7.9
a ≤
= 1.20
FCE = 0.8
VSEC = 5.6V
This ratio is less than the ratio calculated in equation (9).
When specifying the transformer it is necessary to know
the primary peak current which must be withstood
without saturating the transformer core.
then:
65 × 0.8 − 6.0
a ≤
5.6
VIN(min)TON
a ≤ 8.2 (NPRI/NSEC
)
(13)
so:
IPEAK(pri)
=
LPRI
Next, calculate the maximum primary inductance
required to store the needed output energy with a power
switch duty cycle of 55%.
6
3.22 × 7.6 ×10−
2
2
IPEAK(pri)
=
0.5fSW VIN(min) TON
LPRI
(10)
LPRI ≥
POUT
IPEAK(pri) = 2.1A
where:
Now find the minimum reverse voltage requirement for
the output rectifier. This rectifier must have an average
current rating greater than the maximum output current
of 0.5A.
LPRI = maximum primary inductance
fSW = device switching frequency (100kHz)
VIN(min) = minimum input voltage
TON = power switch on time
VIN(max) + (VOUTa)
(14)
VBR ≥
FBR
a
then:
where:
0.5 ×1×105 × (3.22)2 × (7.4 ×10−6 )2
LPRI
≥
VBR = output rectifier maximum peak reverse
voltage rating
2.5
L
PRI ≥ 11.4µH
a = transformer turns ratio (1.2)
Use a 12µH primary inductance to overcome circuit
inefficiencies.
FBR = reverse voltage safety derating factor (0.8)
then:
To complete the design the inductance value of the
secondary is found which will guarantee that the energy
stored in the transformer during the power switch on
time will be completed discharged into the output during
the off-time. This is necessary when operating in
discontinuous-mode.
6.0 + (5.0 ×1.2)
0.8 ×1.2
VBR ≥ 12.5V
VBR
≥
A 1N5817 will safely handle voltage and current require-
ments in this example.
2
2
0.5fSW VSEC TOFF
(11)
LSEC ≤
Forward Converters
POUT
Micrel’s MIC2171 can be used in several circuit
configurations to generate an output voltage which is
less than the input voltage (buck or step-down topology).
Figure 6 shows the MIC2171 in a voltage step-down
application. Because of the internal architecture of these
devices, more external components are required to
implement a step-down regulator than with other devices
offered by Micrel (refer to the LM257x or MIC457x family
of buck switchers). However, for step-down conversion
requiring a transformer (forward), the MIC2171 is a good
choice.
where:
LSEC = maximum secondary inductance
TOFF = power switch off time
then:
0.5 ×1×105 × (5.41)2 × (2.6 ×10−
)
6
2
LSEC
≤
2.5
LSEC ≤ 7.9µH
Finally, recalculate the transformer turns ratio to insure
that it is less than the value earlier found in equation (9).
A 12V to 5V step-down converter using transformer
isolation (forward) is shown in Figure 6. Unlike the
isolated flyback converter which stores energy in the
primary inductance during the controller’s on-time and
releases it to the load during the off-time, the forward
converter transfers energy to the output during the on-
LPRI
(12)
a ≤
LSEC
M9999-051107
May 2007
9
Micrel, Inc.
MIC2171
time, using the off-time to reset the transformer core. In
the application shown, the transformer core is reset by
the tertiary winding discharging T1’s peak magnetizing
current through D2.
off-time would require the voltage across the power
switch to be ten times the input voltage. This would limit
the input voltage to 6V or less for forward converter
applications.
For most forward converters the duty cycle is limited to
50%, allowing the transformer flux to reset with only two
times the input voltage appearing across the power
switch. Although during normal operation this circuit’s
duty cycle is well below 50%, the MIC2172 has a
maximum duty cycle capability of 90%. If 90% was
required during operation (start-up and high load
currents), a complete reset of the transformer during the
To prevent core saturation, the application given here
uses a duty cycle limiter consisting of Q1, C4 and R3.
Whenever the MIC2171 exceeds a duty cycle of 50%,
T1’s reset winding current turns Q1 on. This action
reduces the duty cycle of the MIC2171 until T1 is able to
reset during each cycle.
T1
1:1:1
D3
1N5819
L1 100µH
VOUT
VIN
12V
5V, 1A
R4
D4
1N5819
C5
470µF
3.74k
1%
R1*
C2*
D1*
IN
SW
FB
MIC2171
C1
22µF
D2
1N5819
COMP
GND
R5
1.24k
1%
R2
1k
†
†
Q1
R3
C3
1µF
†
C4
* Voltage clipper
†
Duty cycle limiter
Figure 6. MIC2171 Forward Converter
M9999-051107
May 2007
10
Micrel, Inc.
MIC2171
Package Information
5-Pin TO-220 (T)
5-Pin TO-263 (U)
M9999-051107
May 2007
11
Micrel, Inc.
MIC2171
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2005 Micrel, Incorporated.
M9999-051107
May 2007
12
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