LM2731 [NSC]
0.6/1.6 MHz Boost Converters With 22V Internal FET Switch in SOT-23; 0.6 / 1.6 MHz的升压转换器， 22V内部FET开关采用SOT -23
| 型号： | LM2731 |
| 厂家： | 
National Semiconductor |
| 描述： | 0.6/1.6 MHz Boost Converters With 22V Internal FET Switch in SOT-23 |
| 文件： | 总16页 (文件大小：525K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
July 2003
LM2731
0.6/1.6 MHz Boost Converters With 22V Internal FET
Switch in SOT-23
General Description
The LM2731 switching regulators are current-mode boost
converters operating at fixed frequencies of 1.6 MHz (“X”
option) and 600 kHz (“Y” option).
Features
n 22V DMOS FET switch
n 1.6 MHz (“X”), 0.6 MHz (“Y”) switching frequency
n Low RDS(ON) DMOS FET
The use of SOT-23 package, made possible by the minimal
power loss of the internal 1.8A switch, and use of small
inductors and capacitors result in the industry’s highest
power density. The 22V internal switch makes these solu-
tions perfect for boosting to voltages up to 20V.
n Switch current up to 1.8A
n Wide input voltage range (2.7V–14V)
<
n Low shutdown current ( 1 µA)
n 5-Lead SOT-23 package
n Uses tiny capacitors and inductors
n Cycle-by-cycle current limiting
n Internally compensated
These parts have a logic-level shutdown pin that can be
used to reduce quiescent current and extend battery life.
Protection is provided through cycle-by-cycle current limiting
and thermal shutdown. Internal compensation simplifies de-
sign and reduces component count.
Applications
n White LED Current Source
n PDA’s and Palm-Top Computers
n Digital Cameras
Switch Frequency
X
Y
n Portable Phones and Games
n Local Boost Regulator
1.6 MHz
0.6 MHz
Typical Application Circuit
20059110
20059130
© 2003 National Semiconductor Corporation
DS200591
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Typical Application Circuit (Continued)
20059153
20059156
20059155
White LED Flash Application
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2
Connection Diagram
Top View
20059111
5-Lead SOT-23 Package
See NS Package Number MF05A
Ordering Information
Order Number Package Type Package Drawing
Supplied As
Package ID
S51A
LM2731XMF
1K Tape and Reel
3K Tape and Reel
1K Tape and Reel
3K Tape and Reel
LM2731XMFX
S51A
SOT23-5
MF05A
LM2731YMF
S51B
LM2731YMFX
S51B
Pin Description
Pin
1
Name
SW
Function
Drain of the internal FET switch.
Analog and power ground.
2
GND
FB
3
Feedback point that connects to external resistive divider.
Shutdown control input. Connect to Vin if the feature is not used.
Analog and power input.
4
SHDN
VIN
5
3
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Block Diagram
20059112
the Gm amplifier is derived from the feedback (which
samples the voltage at the output), the action of the PWM
comparator constantly sets the correct peak current through
the FET to keep the output voltage in regulation.
Theory of Operation
The LM2731 is a switching converter IC that operates at a
fixed frequency (0.6 or 1.6 MHz) for fast transient response
over a wide input voltage range and incorporates pulse-by-
pulse current limiting protection. Because this is current
mode control, a 33 mΩ sense resistor in series with the
switch FET is used to provide a voltage (which is propor-
tional to the FET current) to both the input of the pulse width
modulation (PWM) comparator and the current limit ampli-
fier.
Q1 and Q2 along with R3 - R6 form a bandgap voltage
reference used by the IC to hold the output in regulation. The
currents flowing through Q1 and Q2 will be equal, and the
feedback loop will adjust the regulated output to maintain
this. Because of this, the regulated output is always main-
tained at a voltage level equal to the voltage at the FB node
"multiplied up" by the ratio of the output resistive divider.
At the beginning of each cycle, the S-R latch turns on the
FET. As the current through the FET increases, a voltage
(proportional to this current) is summed with the ramp com-
ing from the ramp generator and then fed into the input of the
PWM comparator. When this voltage exceeds the voltage on
the other input (coming from the Gm amplifier), the latch
resets and turns the FET off. Since the signal coming from
The current limit comparator feeds directly into the flip-flop
that drives the switch FET. If the FET current reaches the
limit threshold, the FET is turned off and the cycle terminated
until the next clock pulse. The current limit input terminates
the pulse regardless of the status of the output of the PWM
comparator.
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4
Absolute Maximum Ratings (Note 1)
FB Pin Voltage
−0.4V to +6V
−0.4V to +22V
−0.4V to +14.5V
SW Pin Voltage
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Input Supply Voltage
Shutdown Input Voltage
(Survival)
Storage Temperature Range
Operating Junction
−65˚C to +150˚C
−0.4V to +14.5V
265˚C/W
θJ-A (SOT23-5)
Temperature Range
−40˚C to +125˚C
300˚C
ESD Rating (Note 3)
Human Body Model
Lead Temp. (Soldering, 5 sec.)
Power Dissipation (Note 2)
2 kV
Internally Limited
Electrical Characteristics
Limits in standard typeface are for TJ = 25˚C, and limits in boldface type apply over the full operating temperature range
(−40˚C ≤ TJ ≤ +125˚C). Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL = 0A.
Min
(Note 4)
2.7
5.4
8
Typical
(Note 5)
Max
(Note 4)
14
Symbol
Parameter
Conditions
Units
VIN
Input Voltage
Minimum Output Voltage RL = 43Ω
Under Load
V
V
VOUT (MIN)
VIN = 2.7V
VIN = 3.3V
VIN = 5V
7
10
17
10
12
16
5
X Option
(Note 8)
13
RL = 43Ω
Y Option
(Note 8)
VIN = 2.7V
VIN = 3.3V
VIN = 5V
8.25
10.5
14
RL = 15Ω
X Option
(Note 8)
VIN = 2.7V
VIN = 3.3V
VIN = 5V
3.75
5
6.5
11
6
8.75
5
RL = 15Ω
Y Option
(Note 8)
VIN = 2.7V
VIN = 3.3V
VIN = 5V
5.5
9
7.5
11
2
ISW
Switch Current Limit
(Note 6)
1.8
1.4
A
RDS(ON)
Switch ON Resistance
ISW = 100 mA
Vin = 5V
260
300
400
500
450
550
mΩ
ISW = 100 mA
Vin = 3.3V
Device ON
Device OFF
VSHDN = 0
VSHDN = 5V
SHDNTH
ISHDN
VFB
Shutdown Threshold
1.5
V
0.50
Shutdown Pin Bias
Current
0
0
µA
2
Feedback Pin Reference VIN = 3V
Voltage
1.205
1.230
60
1.255
500
V
IFB
Feedback Pin Bias
Current
VFB = 1.23V
nA
IQ
Quiescent Current
VSHDN = 5V, Switching "X"
2
3.0
2
mA
µA
VSHDN = 5V, Switching "Y"
VSHDN = 5V, Not Switching
VSHDN = 0
1.0
400
0.024
500
1
FB Voltage Line
Regulation
2.7V ≤ VIN ≤ 14V
0.02
%/V
FSW
Switching Frequency
(Note 7)
“X” Option
“Y” Option
“X” Option
“Y” Option
1
0.40
86
1.6
0.60
93
1.85
0.8
MHz
%
DMAX
Maximum Duty Cycle
(Note 7)
92
96
5
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Electrical Characteristics (Continued)
Limits in standard typeface are for TJ = 25˚C, and limits in boldface type apply over the full operating temperature range
(−40˚C ≤ TJ ≤ +125˚C). Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL = 0A.
Min
(Note 4)
Typical
(Note 5)
Max
(Note 4)
1
Symbol
Parameter
Conditions
Units
IL
Switch Leakage
Not Switching VSW = 5V
µA
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the
device outside of the limits set forth under the operating ratings which specify the intended range of operating conditions.
Note 2: The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature, T (MAX) = 125˚C,
J
the junction-to-ambient thermal resistance for the SOT-23 package, θ
= 265˚C/W, and the ambient temperature, T . The maximum allowable power dissipation
J-A
A
at any ambient temperature for designs using this device can be calculated using the formula:
If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing the output voltage as required
to maintain a safe junction temperature.
Note 3: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
Note 4: Limits are guaranteed by testing, statistical correlation, or design.
Note 5: Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most likely expected value
of the parameter at room temperature.
Note 6: Switch current limit is dependent on duty cycle (see Typical Performance Characteristics).
Note 7: Guaranteed limits are the same for Vin = 3.3V input.
Note 8: L = 10 µH, C
= 4.7 µF, duty cycle = maximum
OUT
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Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN
.
Iq Vin (Active) vs Temperature - "X"
Iq Vin (Active) vs Temperature - "Y"
20059104
20059102
Oscillator Frequency vs Temperature - "X"
Oscillator Frequency vs Temperature - "Y"
20059105
20059101
Max. Duty Cycle vs Temperature - "X"
Max. Duty Cycle vs Temperature - "Y"
20059106
20059107
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Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin tied to
VIN. (Continued)
Iq Vin (Idle) vs Temperature
Feedback Bias Current vs Temperature
20059125
20059126
Feedback Voltage vs Temperature
RDS(ON) vs Temperature
20059128
20059127
Current Limit vs Temperature
RDS(ON) vs VIN
20059129
20059152
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Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin tied to
VIN. (Continued)
Efficiency vs Load Current - "X"
VIN = 2.7V, VOUT = 5V
Efficiency vs Load Current - "X"
VIN = 3.3V, VOUT = 5V
20059135
20059136
Efficiency vs Load Current - "X"
VIN = 4.2V, VOUT = 5V
Efficiency vs Load Current - "X"
VIN = 2.7V, VOUT = 12V
20059137
20059138
Efficiency vs Load Current - "X"
VIN = 3.3V, VOUT = 12V
Efficiency vs Load Current - "X"
VIN = 5V, VOUT = 12V
20059140
20059139
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Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin tied to
VIN. (Continued)
Efficiency vs Load Current - "X"
VIN = 5V, VOUT = 18V
Efficiency vs Load Current - "Y"
VIN = 2.7V, VOUT = 5V
20059142
20059141
Efficiency vs Load Current - "Y"
VIN = 3.3V, VOUT = 5V
Efficiency vs Load Current - "Y"
VIN = 4.2V, VOUT = 5V
20059144
20059143
Efficiency vs Load Current - "Y"
VIN = 2.7V, VOUT = 12V
Efficiency vs Load Current - "Y"
VIN = 3.3V, VOUT = 12V
20059145
20059146
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Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin tied to
VIN. (Continued)
Efficiency vs Load Current - "Y"
VIN = 5V, VOUT = 12V
20059147
11
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LAYOUT HINTS
Application Hints
High frequency switching regulators require very careful lay-
out of components in order to get stable operation and low
noise. All components must be as close as possible to the
LM2731 device. It is recommended that a 4-layer PCB be
used so that internal ground planes are available.
SELECTING THE EXTERNAL CAPACITORS
The best capacitors for use with the LM2731 are multi-layer
ceramic capacitors. They have the lowest ESR (equivalent
series resistance) and highest resonance frequency which
makes them optimum for use with high frequency switching
converters.
As an example, a recommended layout of components is
shown:
When selecting a ceramic capacitor, only X5R and X7R
dielectric types should be used. Other types such as Z5U
and Y5F have such severe loss of capacitance due to effects
of temperature variation and applied voltage, they may pro-
vide as little as 20% of rated capacitance in many typical
applications. Always consult capacitor manufacturer’s data
curves before selecting a capacitor. High-quality ceramic
capacitors can be obtained from Taiyo-Yuden, AVX, and
Murata.
SELECTING THE OUTPUT CAPACITOR
A single ceramic capacitor of value 4.7 µF to 10 µF will
provide sufficient output capacitance for most applications. If
larger amounts of capacitance are desired for improved line
support and transient response, tantalum capacitors can be
used. Aluminum electrolytics with ultra low ESR such as
Sanyo Oscon can be used, but are usually prohibitively
expensive. Typical AI electrolytic capacitors are not suitable
for switching frequencies above 500 kHz due to significant
ringing and temperature rise due to self-heating from ripple
current. An output capacitor with excessive ESR can also
reduce phase margin and cause instability.
20059116
Recommended PCB Component Layout
Some additional guidelines to be observed:
1. Keep the path between L1, D1, and C2 extremely short.
Parasitic trace inductance in series with D1 and C2 will
increase noise and ringing.
2. The feedback components R1, R2 and CF must be kept
close to the FB pin of U1 to prevent noise injection on
the FB pin trace.
In general, if electrolytics are used, it is recommended that
they be paralleled with ceramic capacitors to reduce ringing,
switching losses, and output voltage ripple.
3. If internal ground planes are available (recommended)
use vias to connect directly to ground at pin 2 of U1, as
well as the negative sides of capacitors C1 and C2.
SELECTING THE INPUT CAPACITOR
An input capacitor is required to serve as an energy reservoir
for the current which must flow into the coil each time the
switch turns ON. This capacitor must have extremely low
ESR, so ceramic is the best choice. We recommend a
nominal value of 2.2 µF, but larger values can be used. Since
this capacitor reduces the amount of voltage ripple seen at
the input pin, it also reduces the amount of EMI passed back
along that line to other circuitry.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the external resistors R1 and
R2 (see Basic Application Circuit). A value of approximately
13.3 kΩ is recommended for R2 to establish a divider current
of approximately 92 µA. R1 is calculated using the formula:
R1 = R2 X (VOUT/1.23 − 1)
SWITCHING FREQUENCY
FEED-FORWARD COMPENSATION
The LM2731 is provided with two switching frequencies: the
“X” version is typically 1.6 MHz, while the “Y” version is
typically 600 kHz. The best frequency for a specific applica-
tion must be determined based on the trade-offs involved:
Although internally compensated, the feed-forward capacitor
Cf is required for stability (see Basic Application Circuit).
Adding this capacitor puts a zero in the loop response of the
converter. The recommended frequency for the zero fz
should be approximately 6 kHz. Cf can be calculated using
the formula:
Higher switching frequency means the inductors and capaci-
tors can be made smaller and cheaper for a given output
voltage and current. The down side is that efficiency is
slightly lower because the fixed switching losses occur more
frequently and become a larger percentage of total power
loss. EMI is typically worse at higher switching frequencies
because more EMI energy will be seen in the higher fre-
quency spectrum where most circuits are more sensitive to
such interference.
Cf = 1 / (2 X π X R1 X fz)
SELECTING DIODES
The external diode used in the typical application should be
a Schottky diode. A 20V diode such as the MBR0520 is
recommended.
The MBR05XX series of diodes are designed to handle a
maximum average current of 0.5A. For applications exceed-
ing 0.5A average but less than 1A, a Microsemi UPS5817
can be used.
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12
Application Hints (Continued)
20059117
Basic Application Circuit
DUTY CYCLE
means the ON time of the switch is 0.390 µs. It should be
noted that when the switch is ON, the voltage across the
inductor is approximately 4.5V.
The maximum duty cycle of the switching regulator deter-
mines the maximum boost ratio of output-to-input voltage
that the converter can attain in continuous mode of opera-
tion. The duty cycle for a given boost application is defined
as:
Using the equation:
V = L (di/dt)
We can then calculate the di/dt rate of the inductor which is
found to be 0.45 A/µs during the ON time. Using these facts,
we can then show what the inductor current will look like
during operation:
This applies for continuous mode operation.
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I
make the inductor?” (because they are the largest sized
component and usually the most costly). The answer is not
simple and involves trade-offs in performance. Larger induc-
tors mean less inductor ripple current, which typically means
less output voltage ripple (for a given size of output capaci-
tor). Larger inductors also mean more load power can be
delivered because the energy stored during each switching
cycle is:
20059118
10 µH Inductor Current,
5V–12V Boost (LM2731X)
E = L/2 X (lp)2
During the 0.390 µs ON time, the inductor current ramps up
0.176A and ramps down an equal amount during the OFF
time. This is defined as the inductor “ripple current”. It can
also be seen that if the load current drops to about 33 mA,
the inductor current will begin touching the zero axis which
means it will be in discontinuous mode. A similar analysis
can be performed on any boost converter, to make sure the
ripple current is reasonable and continuous operation will be
maintained at the typical load current values.
Where “lp” is the peak inductor current. An important point to
observe is that the LM2731 will limit its switch current based
on peak current. This means that since lp(max) is fixed,
increasing L will increase the maximum amount of power
available to the load. Conversely, using too little inductance
may limit the amount of load current which can be drawn
from the output.
Best performance is usually obtained when the converter is
operated in “continuous” mode at the load current range of
interest, typically giving better load regulation and less out-
put ripple. Continuous operation is defined as not allowing
the inductor current to drop to zero during the cycle. It should
be noted that all boost converters shift over to discontinuous
operation as the output load is reduced far enough, but a
larger inductor stays “continuous” over a wider load current
range.
MAXIMUM SWITCH CURRENT
The maximum FET switch current available before the cur-
rent limiter cuts in is dependent on duty cycle of the appli-
cation. This is illustrated in the graphs below which show
typical values of switch current for both the "X" and "Y"
versions as a function of effective (actual) duty cycle:
To better understand these trade-offs, a typical application
circuit (5V to 12V boost with a 10 µH inductor) will be
analyzed. We will assume:
VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V
Since the frequency is 1.6 MHz (nominal), the period is
approximately 0.625 µs. The duty cycle will be 62.5%, which
13
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switching losses of the FET and diode. For actual load
current in typical applications, we took bench data for vari-
ous input and output voltages for both the "X" and "Y"
versions of the LM2731 and displayed the maximum load
current available for a typical device in graph form:
Application Hints (Continued)
20059150
Switch Current Limit vs Duty Cycle - "X"
20059148
Max. Load Current (typ) vs VIN - "X"
20059151
Switch Current Limit vs Duty Cycle - "Y"
20059149
Max. Load Current (typ) vs VIN - "Y"
CALCULATING LOAD CURRENT
As shown in the figure which depicts inductor current, the
load current is related to the average inductor current by the
relation:
DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in the
equations) is dependent on load current. A good approxima-
tion can be obtained by multiplying the "ON Resistance" of
the FET times the average inductor current.
ILOAD = IIND(AVG) x (1 - DC)
Where "DC" is the duty cycle of the application. The switch
current can be found by:
FET on resistance increases at VIN values below 5V, since
the internal N-FET has less gate voltage in this input voltage
range (see Typical performance Characteristics curves).
Above VIN = 5V, the FET gate voltage is internally clamped
to 5V.
1
ISW = IIND(AVG) + ⁄
2
(IRIPPLE
)
Inductor ripple current is dependent on inductance, duty
cycle, input voltage and frequency:
IRIPPLE = DC x (VIN-VSW) / (f x L)
combining all terms, we can develop an expression which
allows the maximum available load current to be calculated:
The maximum peak switch current the device can deliver is
dependent on duty cycle. For higher duty cycles, see Typical
performance Characteristics curves.
THERMAL CONSIDERATIONS
At higher duty cycles, the increased ON time of the FET
means the maximum output current will be determined by
power dissipation within the LM2731 FET switch. The switch
power dissipation from ON-state conduction is calculated by:
The equation shown to calculate maximum load current
takes into account the losses in the inductor or turn-OFF
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14
that the continuous current rating is high enough to avoid
saturation at peak currents. A suitable core type must be
used to minimize core (switching) losses, and wire power
losses must be considered when selecting the current rating.
Application Hints (Continued)
P(SW) = DC x IIND(AVE)2 x RDS(ON)
There will be some switching losses as well, so some derat-
ing needs to be applied when calculating IC power dissipa-
tion.
SHUTDOWN PIN OPERATION
The device is turned off by pulling the shutdown pin low. If
this function is not going to be used, the pin should be tied
directly to VIN. If the SHDN function will be needed, a pull-up
resistor must be used to VIN (approximately 50k-100kΩ rec-
ommended). The SHDN pin must not be left unterminated.
INDUCTOR SUPPLIERS
Recommended suppliers of inductors for this product in-
clude, but are not limited to Sumida, Coilcraft, Panasonic,
TDK and Murata. When selecting an inductor, make certain
15
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Physical Dimensions inches (millimeters) unless otherwise noted
5-Lead SOT-23 Package
Order Number LM2731XMF, LM2731XMFX, LM2731YMF or LM2731YMFX
NS Package Number MF05A
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DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
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systems which, (a) are intended for surgical implant
into the body, or (b) support or sustain life, and
whose failure to perform when properly used in
accordance with instructions for use provided in the
labeling, can be reasonably expected to result in a
significant injury to the user.
2. A critical component is any component of a life
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.
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