LMR64010XMF [NSC]
SIMPLE SWITCHER® 40Vout, 1A Step-Up Voltage Regulator in SOT-23; SIMPLE SWITCHER® 40Vout ， 1A升压型稳压器采用SOT -23
| 型号： | LMR64010XMF |
| 厂家： | 
National Semiconductor |
| 描述： | SIMPLE SWITCHER® 40Vout, 1A Step-Up Voltage Regulator in SOT-23 |
| 文件： | 总12页 (文件大小：376K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
September 23, 2011
LMR64010
SIMPLE SWITCHER® 40Vout, 1A Step-Up Voltage
Regulator in SOT-23
Applications
Boost Conversions from 3.3V, 5V, and 12V Rails
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Space Constrained Applications
Embedded Systems
LCD Displays
LED Applications
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System Performance
Features
Input voltage range of 2.7V to 14V
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Output voltage up to 40V
Switch current up to 1A
1.6 MHz switching frequency
Low shutdown Iq, <1 µA
Cycle-by-cycle current limiting
Internally compensated
SOT23-5 packaging (2.92 x 2.84 x 1.08mm)
Fully enabled for WEBENCH® Power Designer
■
Performance Benefits
30167557
Extremely easy to use
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Tiny overall solution reduces system cost
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30167558
© 2011 National Semiconductor Corporation
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Connection Diagram
Top View
30167502
5-Lead SOT-23 Package
See NS Package Number MF05A
Ordering Information
Order Number
LMR64010XMFE
LMR64010XMF
LMR64010XMFX
Package Type
Package Drawing
Supplied As
Package ID
250 Units, Tape and Reel
1000 Units, Tape and Reel
3000 Units, Tape and Reel
SF9B
SOT23-5
MF05A
Pin Descriptions
Pin
1
Name
SW
Function
Drain of the internal FET switch.
Analog and power ground.
2
GND
FB
3
Feedback point that connects to external resistive divider.
Shutdown control input. Connect to VIN if this feature is not used.
Analog and power input.
4
SHDN
VIN
5
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2
SW Pin Voltage
Input Supply Voltage
SHDN Pin Voltage
−0.4V to +40V
−0.4V to +14.5V
−0.4V to VIN + 0.3V
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
θ
J-A (SOT23-5)
265°C/W
ESD Rating (Note 3)
Human Body Model
Machine Model
For soldering specifications: see product folder at
www.national.com and www.national.com/ms/MS/MS-
SOLDERING.pdf
Storage Temperature Range
Operating Junction
Temperature Range
Lead Temp. (Soldering, 5 sec.)
Power Dissipation (Note 2)
FB Pin Voltage
−65°C to +150°C
2 kV
200V
−40°C to +125°C
300°C
Internally Limited
−0.4V to +6V
Electrical Characteristics
Limits in standard typeface are for TJ = 25°C, and limits in boldface type apply over the full operating temperature range
(−40°C ≤ TJ ≤ +125°C). Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL = 0A.
Min
(Note 4)
Typical
(Note 5)
Max
(Note 4)
Symbol
VIN
Parameter
Input Voltage
Conditions
Units
2.7
14
V
A
ISW
Switch Current Limit
Switch ON Resistance
Shutdown Threshold
(Note 6)
1.0
1.5
RDS(ON)
SHDNTH
ISW = 100 mA
Device ON
Device OFF
VSHDN = 0
500
650
0.50
mΩ
1.5
V
µA
V
ISHDN
Shutdown Pin Bias Current
0
0
VSHDN = 5V
VIN = 3V
2
VFB
Feedback Pin Reference
Voltage
1.205
1.230
1.255
IFB
IQ
Feedback Pin Bias Current
Quiescent Current
VFB = 1.23V
60
2.1
nA
VSHDN = 5V, Switching
VSHDN = 5V, Not Switching
VSHDN = 0
3.0
500
1
mA
400
µA
0.024
FB Voltage Line Regulation
2.7V ≤ VIN ≤ 14V
0.02
%/V
FSW
DMAX
IL
Switching Frequency
Maximum Duty Cycle
Switch Leakage
1.15
87
1.6
93
1.85
MHz
%
Not Switching VSW = 5V
1
µA
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the
device outside of the limits set forth under the operating ratings which specify the intended range of operating conditions.
Note 2: The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature, TJ(MAX) = 125°
C, the junction-to-ambient thermal resistance for the SOT-23 package, θJ-A = 265°C/W, and the ambient temperature, TA. The maximum allowable power
dissipation at any ambient temperature for designs using this device can be calculated using the formula:
If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing the output voltage as
required to maintain a safe junction temperature.
Note 3: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. The machine model is a 200 pF capacitor discharged
directly into each pin.
Note 4: Limits are guaranteed by testing, statistical correlation, or design.
Note 5: Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most likely expected value
of the parameter at room temperature.
Note 6: Switch current limit is dependent on duty cycle (see Typical Performance Characteristics). Limits shown are for duty cycles ≤ 50%.
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Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin is tied to VIN.
Iq VIN (Active) vs Temperature
Oscillator Frequency vs Temperature
30167508
30167510
Max. Duty Cycle vs Temperature
Feedback Voltage vs Temperature
30167506
30167555
RDS(ON) vs Temperature
Current Limit vs Temperature
30167509
30167507
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4
RDS(ON) vs VIN
Efficiency vs Load Current (VOUT = 12V)
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Efficiency vs Load Current (VOUT = 15V)
Efficiency vs Load Current (VOUT = 20V)
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Efficiency vs Load Current (VOUT = 25V)
Efficiency vs Load Current (VOUT = 30V)
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Efficiency vs Load Current (VOUT = 35V)
Efficiency vs Load Current (VOUT = 40V)
30167550
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6
Block Diagram
30167503
currents flowing through Q1 and Q2 will be equal, and the
General Description
feedback loop will adjust the regulated output to maintain this.
Because of this, the regulated output is always maintained at
a voltage level equal to the voltage at the FB node "multiplied
up" by the ratio of the output resistive divider.
The LMR64010 switching regulators is a current-mode boost
converter operating at a fixed frequency of 1.6 MHz.
The use of SOT-23 package, made possible by the minimal
power loss of the internal 1A switch, and use of small induc-
tors and capacitors result in the industry's highest power
density. The 40V internal switch makes these solutions per-
fect for boosting to voltages of 16V or greater.
The current limit comparator feeds directly into the flip-flop,
that drives the switch FET. If the FET current reaches the limit
threshold, the FET is turned off and the cycle terminated until
the next clock pulse. The current limit input terminates the
pulse regardless of the status of the output of the PWM com-
parator.
These parts have a logic-level shutdown pin that can be used
to reduce quiescent current and extend battery life.
Protection is provided through cycle-by-cycle current limiting
and thermal shutdown. Internal compensation simplifies de-
sign and reduces component count.
Application Hints
SELECTING THE EXTERNAL CAPACITORS
The best capacitors for use with the LMR64010 are multi-lay-
er ceramic capacitors. They have the lowest ESR (equivalent
series resistance) and highest resonance frequency which
makes them optimum for use with high frequency switching
converters.
Theory of Operation
The LMR64010 is a switching converter IC that operates at a
fixed frequency (1.6 MHz) using current-mode control for fast
transient response over a wide input voltage range and in-
corporates pulse-by-pulse current limiting protection. Be-
cause this is current mode control, a 50 mΩ sense resistor in
series with the switch FET is used to provide a voltage (which
is proportional to the FET current) to both the input of the
pulse width modulation (PWM) comparator and the current
limit amplifier.
When selecting a ceramic capacitor, only X5R and X7R di-
electric types should be used. Other types such as Z5U and
Y5F have such severe loss of capacitance due to effects of
temperature variation and applied voltage, they may provide
as little as 20% of rated capacitance in many typical applica-
tions. Always consult capacitor manufacturer’s data curves
before selecting a capacitor.
At the beginning of each cycle, the S-R latch turns on the FET.
As the current through the FET increases, a voltage (propor-
tional to this current) is summed with the ramp coming from
the ramp generator and then fed into the input of the PWM
comparator. When this voltage exceeds the voltage on the
other input (coming from the Gm amplifier), the latch resets
and turns the FET off. Since the signal coming from the Gm
amplifier is derived from the feedback (which samples the
voltage at the output), the action of the PWM comparator
constantly sets the correct peak current through the FET to
keep the output volatge in regulation.
SELECTING THE OUTPUT CAPACITOR
A single ceramic capacitor of value 4.7 µF to 10 µF will provide
sufficient output capacitance for most applications. For output
voltages below 10V, a 10 µF capacitance is required. If larger
amounts of capacitance are desired for improved line support
and transient response, tantalum capacitors can be used in
parallel with the ceramics. Aluminum electrolytics with ultra
low ESR such as Sanyo Oscon can be used, but are usually
prohibitively expensive. Typical AI electrolytic capacitors are
not suitable for switching frequencies above 500 kHz due to
significant ringing and temperature rise due to self-heating
Q1 and Q2 along with R3 - R6 form a bandgap voltage refer-
ence used by the IC to hold the output in regulation. The
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from ripple current. An output capacitor with excessive ESR
can also reduce phase margin and cause instability.
SELECTING THE INPUT CAPACITOR
An input capacitor is required to serve as an energy reservoir
for the current which must flow into the coil each time the
switch turns ON. This capacitor must have extremely low
ESR, so ceramic is the best choice. We recommend a nomi-
nal value of 2.2 µF, but larger values can be used. Since this
capacitor reduces the amount of voltage ripple seen at the
input pin, it also reduces the amount of EMI passed back
along that line to other circuitry.
FEED-FORWARD COMPENSATION
Although internally compensated, the feed-forward capacitor
Cf is required for stability (see Basic Application Circuit).
Adding this capacitor puts a zero in the loop response of the
converter. Without it, the regulator loop can oscillate. The
recommended frequency for the zero fz should be approxi-
mately 8 kHz. Cf can be calculated using the formula:
30167522
Cf = 1 / (2 X π X R1 X fz)
Recommended PCB Component Layout
SELECTING DIODES
Some additional guidelines to be observed:
The external diode used in the typical application should be
a Schottky diode. If the switch voltage is less than 15V, a 20V
diode such as the MBR0520 is recommended. If the switch
voltage is between 15V and 25V, a 30V diode such as the
MBR0530 is recommended. If the switch voltage exceeds
25V, a 40V diode such as the MBR0540 should be used.
1. Keep the path between L1, D1, and C2 extremely short.
Parasitic trace inductance in series with D1 and C2 will
increase noise and ringing.
2. The feedback components R1, R2 and CF must be kept
close to the FB pin of U1 to prevent noise injection on the
FB pin trace.
The MBR05XX series of diodes are designed to handle a
maximum average current of 0.5A. For applications exceed-
ing 0.5A average but less than 1A, a Toshiba CRS08 can be
used.
3. If internal ground planes are available (recommended)
use vias to connect directly to ground at pin 2 of U1, as
well as the negative sides of capacitors C1 and C2.
LAYOUT HINTS
SETTING THE OUTPUT VOLTAGE
High frequency switching regulators require very careful lay-
out of components in order to get stable operation and low
noise. All components must be as close as possible to the
LMR64010 device. It is recommended that a 4-layer PCB be
used so that internal ground planes are available.
The output voltage is set using the external resistors R1 and
R2 (see Basic Application Circuit). A value of approximately
13.3 kΩ is recommended for R2 to establish a divider current
of approximately 92 µA. R1 is calculated using the formula:
R1 = R2 X (VOUT/1.23 − 1)
As an example, a recommended layout of components is
shown:
30167505
Basic Application Circuit
DUTY CYCLE
This applies for continuous mode operation.
The maximum duty cycle of the switching regulator deter-
mines the maximum boost ratio of output-to-input voltage that
the converter can attain in continuous mode of operation. The
duty cycle for a given boost application is defined as:
The equation shown for calculating duty cycle incorporates
terms for the FET switch voltage and diode forward voltage.
The actual duty cycle measured in operation will also be af-
fected slightly by other power losses in the circuit such as wire
losses in the inductor, switching losses, and capacitor ripple
current losses from self-heating. Therefore, the actual (effec-
tive) duty cycle measured may be slightly higher than calcu-
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8
lated to compensate for these power losses. A good
approximation for effctive duty cycle is :
inductor current will begin touching the zero axis which means
it will be in discontinuous mode. A similar analysis can be
performed on any boost converter, to make sure the ripple
current is reasonable and continuous operation will be main-
tained at the typical load current values.
DC (eff) = (1 - Efficiency x (VIN/VOUT))
Where the efficiency can be approximated from the curves
provided.
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I
make the inductor?” (because they are the largest sized com-
ponent and usually the most costly). The answer is not simple
and involves tradeoffs in performance. Larger inductors mean
less inductor ripple current, which typically means less output
voltage ripple (for a given size of output capacitor). Larger
inductors also mean more load power can be delivered be-
cause the energy stored during each switching cycle is:
E =L/2 X (lp)2
30167524
Typical Application, 5V–12V Boost
Where “lp” is the peak inductor current. An important point to
observe is that the LMR64010 will limit its switch current
based on peak current. This means that since lp(max) is fixed,
increasing L will increase the maximum amount of power
available to the load. Conversely, using too little inductance
may limit the amount of load current which can be drawn from
the output.
MAXIMUM SWITCH CURRENT
The maximum FET swtch current available before the current
limiter cuts in is dependent on duty cycle of the application.
This is illustrated in the graphs below which show both the
typical and guaranteed values of switch current as a function
of effective (actual) duty cycle:
Best performance is usually obtained when the converter is
operated in “continuous” mode at the load current range of
interest, typically giving better load regulation and less output
ripple. Continuous operation is defined as not allowing the in-
ductor current to drop to zero during the cycle. It should be
noted that all boost converters shift over to discontinuous op-
eration as the output load is reduced far enough, but a larger
inductor stays “continuous” over a wider load current range.
To better understand these tradeoffs, a typical application cir-
cuit (5V to 12V boost with a 10 µH inductor) will be analyzed.
We will assume:
VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V
Since the frequency is 1.6 MHz (nominal), the period is ap-
proximately 0.625 µs. The duty cycle will be 62.5%, which
means the ON time of the switch is 0.390 µs. It should be
noted that when the switch is ON, the voltage across the in-
ductor is approximately 4.5V.
Using the equation:
V = L (di/dt)
30167525
Switch Current Limit vs Duty Cycle
We can then calculate the di/dt rate of the inductor which is
found to be 0.45 A/µs during the ON time. Using these facts,
we can then show what the inductor current will look like dur-
ing operation:
CALCULATING LOAD CURRENT
As shown in the figure which depicts inductor current, the load
current is related to the average inductor current by the rela-
tion:
ILOAD = IIND(AVG) x (1 - DC)
Where "DC" is the duty cycle of the application. The switch
current can be found by:
ISW = IIND(AVG) + ½ (IRIPPLE
)
Inductor ripple current is dependent on inductance, duty cy-
cle, input voltage and frequency:
IRIPPLE = DC x (VIN-VSW) / (f x L)
combining all terms, we can develop an expression which al-
lows the maximum available load current to be calculated:
30167512
10 µH Inductor Current,5V–12V Boost
During the 0.390 µs ON time, the inductor current ramps up
0.176A and ramps down an equal amount during the OFF
time. This is defined as the inductor “ripple current”. It can also
be seen that if the load current drops to about 33 mA, the
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The equation shown to calculate maximum load current takes
into account the losses in the inductor or turn-OFF switching
losses of the FET and diode. For actual load current in typical
applications, we took bench data for various input and output
voltages and displayed the maximum load current available
for a typical device in graph form:
possible inductance value for cost and size savings. The con-
verter will operate in discontinuous mode in such a case.
The minimum inductance should be selected such that the
inductor (switch) current peak on each cycle does not reach
the 1A current limit maximum. To understand how to do this,
an example will be presented.
In the example, minimum switching frequency of 1.15 MHz
will be used. This means the maximum cycle period is the
reciprocal of the minimum frequency:
TON(max) = 1/1.15M = 0.870 µs
We will assume the input voltage is 5V, VOUT = 12V, VSW
0.2V, VDIODE = 0.3V. The duty cycle is:
=
Duty Cycle = 60.3%
Therefore, the maximum switch ON time is 0.524 µs. An in-
ductor should be selected with enough inductance to prevent
the switch current from reaching 1A in the 0.524 µs ON time
interval (see below):
30167534
Max. Load Current vs VIN
Discontinuous Design, 5V–12V Boost 30167513
DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in the
equations) is dependent on load current. A good approxima-
tion can be obtained by multiplying the "ON Resistance" of
the FET times the average inductor current.
The voltage across the inductor during ON time is 4.8V. Min-
imum inductance value is found by:
V = L X dl/dt, L = V X (dt/dl) = 4.8 (0.524µ/1) = 2.5 µH
FET on resistance increases at VIN values below 5V, since
the internal N-FET has less gate voltage in this input voltage
range (see Typical performance Characteristics curves).
Above VIN = 5V, the FET gate voltage is internally clamped to
5V.
In this case, a 2.7 µH inductor could be used assuming it pro-
vided at least that much inductance up to the 1A current value.
This same analysis can be used to find the minimum induc-
tance for any boost application.
When selecting an inductor, make certain that the continuous
current rating is high enough to avoid saturation at peak cur-
rents. A suitable core type must be used to minimize core
(switching) losses, and wire power losses must be considered
when selecting the current rating.
The maximum peak switch current the device can deliver is
dependent on duty cycle. The minimum value is guaranteed
to be > 1A at duty cycle below 50%. For higher duty cycles,
see Typical performance Characteristics curves.
THERMAL CONSIDERATIONS
SHUTDOWN PIN OPERATION
At higher duty cycles, the increased ON time of the FET
means the maximum output current will be determined by
power dissipation within the LMR64010 FET switch. The
switch power dissipation from ON-state conduction is calcu-
lated by:
The device is turned off by pulling the shutdown pin low. If this
function is not going to be used, the pin should be tied directly
to VIN. If the SHDN function will be needed, a pull-up resistor
must be used to VIN (approximately 50k-100kΩ recommend-
ed). The SHDN pin must not be left unterminated.
P(SW) = DC x IIND(AVE)2 x RDSON
There will be some switching losses as well, so some derating
needs to be applied when calculating IC power dissipation.
MINIMUM INDUCTANCE
In some applications where the maximum load current is rel-
atively small, it may be advantageous to use the smallest
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10
Physical Dimensions inches (millimeters) unless otherwise noted
5-Lead SOT-23 Package
Order Number LMR64010XMF, LMR64010XMFX
NS Package Number MF05A
11
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相关型号：
LMR64010XMF/NOPB
SIMPLE SWITCHER® 2.7V to 14V, 1A Step-Up Regulator in SOT-23 Package 5-SOT-23 -40 to 125
TI
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