ISL6420AIR-TK [RENESAS]
SWITCHING CONTROLLER, 1400kHz SWITCHING FREQ-MAX, PQCC20, 4 X 4 MM, PLASTIC, MO-220VGGD-1, QFN-20;型号: | ISL6420AIR-TK |
厂家: | RENESAS TECHNOLOGY CORP |
描述: | SWITCHING CONTROLLER, 1400kHz SWITCHING FREQ-MAX, PQCC20, 4 X 4 MM, PLASTIC, MO-220VGGD-1, QFN-20 开关 |
文件: | 总21页 (文件大小:1082K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
DATASHEET
ISL6420A
FN9169
Rev 4.00
December 4, 2009
Advanced Single Synchronous Buck Pulse-Width Modulation (PWM) Controller
The ISL6420A simplifies the implementation of a
complete control and protection scheme for a
Features
• Operates From:
- 4.5V to 5.5V Input
- 5.5V to 28V Input
high-performance DC/DC buck converter. It is designed
to drive N-Channel MOSFETs in a synchronous rectified
buck topology. The ISL6420A integrates control, output
adjustment, monitoring and protection functions into a
single package. Additionally, the IC features an
external reference voltage tracking mode for externally
referenced buck converter applications and DDR
termination supplies, as well as a voltage margining
mode for system testing in networking DC/DC
converter applications.
• 0.6V Internal Reference Voltage
- ±1.0% Reference Accuracy
• Resistor-Selectable Switching Frequency
- 100kHz to 1.4MHz
• Voltage Margining and External Reference Tracking
Modes
The ISL6420A provides simple, single feedback loop,
voltage mode control with fast transient response. The
output voltage of the converter can be precisely
regulated to as low as 0.6V.
• Output can Sink or Source Current
• Lossless, Programmable Overcurrent Protection
- Uses Upper MOSFET’s r
DS(ON)
The operating frequency is fully adjustable from
100kHz to 1.4MHz. High frequency operation offers
cost and space savings.
• Programmable Soft-Start
• Drives N-Channel MOSFETs
• Simple Single-Loop Control Design
- Voltage-Mode PWM Control
The error amplifier features a 15MHz gain-bandwidth
product and 6V/µs slew rate that enables high
converter bandwidth for fast transient response. The
PWM duty cycle ranges from 0% to 100% in transient
conditions. Selecting the capacitor value from the
ENSS pin to ground sets a fully adjustable PWM
soft-start. Pulling the ENSS pin LOW disables the
controller.
• Fast Transient Response
- High-Bandwidth Error Amplifier
- Full 0% to 100% Duty Cycle
• Extensive Circuit Protection Functions
- PGOOD, Overvoltage, Overcurrent, Shutdown
The ISL6420A monitors the output voltage and
generates a PGOOD (power good) signal when
soft-start sequence is complete and the output is
within regulation. A built-in overvoltage protection
circuit prevents the output voltage from going above
typically 115% of the set point. Protection from
overcurrent conditions is provided by monitoring the
• Diode Emulation during Startup for Pre-Biased
Load Applications
• Offered in 20 Ld QFN and QSOP Packages
• QFN (4x4) Package
- QFN compliant to JEDEC PUB95 MO-220
QFN -Quad Flat No Leads - Product Outline
r
of the upper MOSFET to inhibit the PWM
- QFN Near Chip Scale Package Footprint;
Improves PCB Efficiency, Thinner in Profile
DS(ON)
operation appropriately. This approach simplifies the
implementation and improves efficiency by eliminating
the need for a current sensing resistor.
• Pb-Free (RoHS Compliant)
Applications
• Power Supplies for Microprocessors/ASICs
- Embedded Controllers
- DSP and Core Processors
- DDR SDRAM Bus Termination
• Ethernet Routers and Switchers
• High-Power DC/DC Regulators
• Distributed DC/DC Power Architecture
• Personal Computer Peripherals
• Externally Referenced Buck Converters
FN9169 Rev 4.00
December 4, 2009
Page 1 of 21
ISL6420A
Ordering Information
PART NUMBER
(Notes 2, 3)
PART
MARKING
TEMP.
RANGE (°C)
PACKAGE
(Pb-Free)
PKG.
DWG. #
ISL6420AIAZ
6420 AIAZ
-40 to +85
-40 to +85
-40 to +85
-40 to +85
20 Ld QSOP
M20.15
ISL6420AIAZ-TK (Note 1)
ISL6420AIRZ
6420 AIAZ
64 20AIRZ
64 20AIRZ
20 Ld QSOP (Tape and Reel)
20 Ld 4x4 QFN
M20.15
L20.4x4
ISL6420AIRZ-TK (Note 1)
20 Ld 4x4 QFN (Tape and Reel) L20.4x4
NOTES:
1. Please refer to TB347 for details on reel specifications.
2. These Intersil Pb-free plastic packaged products employ special Pb-free material sets, molding compounds/die attach
materials, and 100% matte tin plate plus anneal (e3 termination finish, which is RoHS compliant and compatible with both
SnPb and Pb-free soldering operations). Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that
meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
3. For Moisture Sensitivity Level (MSL), please see device information page for ISL6240A For more information on MSL please
see techbrief TB363.
Pin Configurations
ISL6420A
(20 LD QFN)
TOP VIEW
ISL6420A
(20 LD QSOP)
TOP VIEW
1
PGOOD
ENSS
CDEL
PGND
20
19
2
3
4
5
6
7
8
9
20 19 18 17 16
LGATE
18 COMP
17 FB
PVCC
GPIO2
GPIO1/REFIN
OCSET
1
2
3
4
5
15 PGND
14 CDEL
13 PGOOD
12 ENSS
11 COMP
PHASE
16 RT
UGATE
SGND
VIN
15
14
BOOT
GPIO2
13 VCC5
12
REFOUT
GPIO1/REFIN
VMSET/MODE
VMSET/MODE
OCSET 10
11 REFOUT
6
7
8
9
10
FN9169 Rev 4.00
December 4, 2009
Page 2 of 21
Block Diagram
VIN
VCC5
ENSS
10µA
INTERNAL SERIES
LINEAR
REFOUT
POWER-ON
OCSET
ENSS
RESET (POR)
INTERNAL
0.6V
OTP
SSDONE
100µA
BOOT
GPIO1/REFIN
GPIO2
UGATE
SSDONE
VOLTAGE
MARGINING
CONTROL
FAULT LOGIC
SSDONE
VMSET/MODE
PHASE
GATE
CONTROL
LOGIC
CDEL
PWM
COMP
SS
REF
PVCC
V
FB
EA
LGATE
PGND
COMP
OSCILLATOR
PGOOD
OV/UV
COMP
PGOOD
COMP
SGND
EP (QFN ONLY)
RT
ISL6420A
Typical 5V Input DC/DC Application Schematic
5V ±10%
C
6
C
1
C
C
C
3
5
2
C
4
VCC5
VIN
PVCC
D
1
9
R
1
OCSET
BOOT
MONITOR AND
PROTECTION
ENSS
RT
Q
1
C
UGATE
PHASE
PGOOD
CDEL
OSC
C
7
R
2
L
1
0.1µF
REF
V
OUT
C
8
SGND
FB
Q
LGATE
PGND
2
-
C
10
+
+
-
R
3
COMP
GPIO1/REFIN
GPIO2
C
11
R
6
C
R
12
5
C
13
VOLTAGE MARGINING ENABLED WITH INTERNAL V
SEE PAGE 12 FOR MORE DETAILS
R
REF
4
Typical 5.5V to 28V Input DC/DC Application Schematic
5.5V to 28V
C
6
C
1
C
C
C
5
3
2
C
4
VIN VCC5
PVCC
D
1
R
1
OCSET
BOOT
MONITOR AND
PROTECTION
ENSS
RT
Q
1
UGATE
PHASE
C
9
PGOOD
CDEL
OSC
C
7
R
2
L
1
0.1µF
REF
V
OUT
C
8
SGND
FB
Q
LGATE
PGND
C
2
-
10
+
+
-
R
3
COMP
GPIO1/REFIN
GPIO2
C
R
11
R
6
C
12
5
C
13
R
4
VOLTAGE MARGINING ENABLED WITH INTERNAL V
SEE PAGE 12 FOR MORE DETAILS
REF
FN9169 Rev 4.00
December 4, 2009
Page 4 of 21
ISL6420A
Typical 5V Input DC/DC Application Schematic
5V ±10%
C6
C
1
C
C
3
C
5
2
C
4
D
PVCC
VIN
VCC5
1
R
1
OCSET
BOOT
MONITOR AND
PROTECTION
ENSS
RT
Q
1
C
8
CDEL
UGATE
PHASE
OSC
R
C
2
7
L
PGOOD
1
REF
2.5V/1.25V
SGND
FB
LGATE
PGND
Q
2
-
+
C
9
+
-
COMP
R
3
C
10
GPIO1/REFIN <-- V
REF
= VDDQ/2
1.25V V
REF
TO REFIN OF VTT SUPPLY
C
11
R
5
C
12
R
4
VCC5
CONFIGURATION FOR DDR TERMINATION/EXTERNALLY REFERENCED TRACKING APPLICATIONS
Typical 5.5V to 28V Input DC/DC Application Schematic
5.5V to 28V
C
6
C
1
C
C
3
C
5
2
C
4
D
VIN
VCC5
PVCC
1
R
1
OCSET
BOOT
MONITOR AND
PROTECTION
ENSS
RT
Q
1
C
UGATE
PHASE
8
CDEL
OSC
R
C7
2
L
PGOOD
1
REF
2.5V/1.25V
SGND
FB
Q
LGATE
PGND
2
-
+
C
9
+
-
R
3
COMP
C
C
10
GPIO1/REFIN <-- V
= VDDQ/2
REF
1.25V V
REF
TO REFIN OF VTT SUPPLY
11
R
5
R
4
VCC5
CONFIGURATION FOR DDR TERMINATION/EXTERNALLY REFERENCED TRACKING APPLICATIONS
FN9169 Rev 4.00
December 4, 2009
Page 5 of 21
ISL6420A
Typical 5V Input DC/DC Application Schematic
5V ±10%
C
6
C
1
C
C
C
5
3
2
C
4
VCC5
VIN
PVCC
D
1
R
1
OCSET
BOOT
MONITOR AND
PROTECTION
ENSS
RT
PGOOD
CDEL
Q
1
C
9
UGATE
PHASE
OSC
C7
0.1µF
R
2
L
1
REF
V
OUT
C
8
SGND
FB
Q
LGATE
PGND
C
10
2
-
+
+
-
R
3
COMP
C
11
C
12
R
5
USE INTERNAL V
VOLTAGE MARGINING
SEE PAGE 12 FOR MORE DETAILS
WITH NO
REF
C13
R
4
Typical 5.5V to 28V Input DC/DC Application Schematic
5.5V to 28V
C
6
C
1
C
C
C
5
3
2
C
4
VCC5
VIN
PVCC
D
1
R
OCSET
BOOT
1
MONITOR AND
PROTECTION
ENSS
RT
Q
1
C
9
UGATE
PHASE
PGOOD
CDEL
OSC
C7
0.1µF
R
2
L
1
REF
V
OUT
C
8
SGND
Q
LGATE
PGND
2
C
10
-
+
+
F
B
-
COMP
R
3
C
11
C
12
R
5
USE INTERNAL V
VOLTAGE MARGINING
WITH NO
REF
SEE PAGE 12 FOR MORE DETAILS
C
13
R
4
FN9169 Rev 4.00
December 4, 2009
Page 6 of 21
ISL6420A
Absolute Maximum Ratings (Note 4)
Thermal Information
Bias Voltage, VIN . . . . . . . . . . . . . . . . . . . . . . . . . . . +30V
Thermal Resistance (Typical)
(°C/W) (°C/W)
JC
JA
BOOT and U
Pins . . . . . . . . . . . . . . . . . . . . . . . . . +36V
GATE
QFN Package (Notes 5, 6) . . . . . . .
QSOP Package (Note 5) . . . . . . . .
47
90
8.5
NA
Maximum Junction Temperature (Plastic Package) . . +150°C
Maximum Storage Temperature Range . . . -65°C to +150°C
Ambient Temperature Range . -40°C to +85°C (for “I” suffix)
Junction Temperature Range . . . . . . . . . . -40°C to +125°C
Pb-Free Reflow Profile . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact
product reliability and result in failures not covered by warranty.
NOTES:
4. All voltages are with respect to GND.
5. is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach”
JA
features. See Tech Brief TB379.
6. For , the “case temp” location is the center of the exposed metal pad on the package underside.
JC
Electrical Specifications
Operating Conditions: VIN = 12V, PVCC shorted with VCC5, T = +25°C. Boldface limits
apply over the operating temperature range, -40°C to +85°C.
A
MIN
MAX
PARAMETER
VIN SUPPLY
SYMBOL
TEST CONDITIONS
(Note 12) TYP (Note 12) UNITS
Input Voltage Range
5.6
12
28
V
VIN SUPPLY CURRENT
Shutdown Current (Note 7)
Operating Current (Notes 7, 8)
VCC5 SUPPLY (Notes 8, 9)
Input Voltage Range
ENSS = GND
-
-
1.4
2.0
-
mA
mA
3.0
VIN = VCC5 for 5V configuration
4.5
4.5
50
5.0
5.0
-
5.5
5.5
-
V
V
Output Voltage
VIN = 5.6V to 28V, I = 3mA to 50mA
L
Maximum Output Current
POWER-ON RESET
VIN = 12V
mA
Rising VCC5 Threshold
VIN connected to VCC5, 5V input
operation
4.310
4.400
4.475
V
Falling VCC5 Threshold
UVLO Threshold Hysteresis
PWM CONVERTERS
Maximum Duty Cycle
Minimum Duty Cycle
FB Pin Bias Current
Undervoltage Protection
Overvoltage Protection
OSCILLATOR
4.090
0.16
4.100
-
4.250
-
V
V
f
f
= 300kHz
= 300kHz
90
-
96
-
-
0
%
%
nA
%
%
SW
SW
-
80
-
-
V
Fraction of the set point; ~3µs noise filter
Fraction of the set point; ~1µs noise filter
75
112
85
120
UV
V
-
OVP
Free Running Frequency
Total Variation
RT = VCC5, T = -40°C to +85°C
270
-
300
330
-
kHz
%
A
T = -40°C to +85°C, with frequency set
A
by external resistor at RT
±10%
Frequency Range (Set by RT)
Ramp Amplitude (Note 10)
VIN = 12V
100
-
-
1400
-
kHz
V
1.25
V
P-P
OSC
FN9169 Rev 4.00
December 4, 2009
Page 7 of 21
ISL6420A
Electrical Specifications
Operating Conditions: VIN = 12V, PVCC shorted with VCC5, T = +25°C. Boldface limits
apply over the operating temperature range, -40°C to +85°C. (Continued)
A
MIN
MAX
PARAMETER
SYMBOL
TEST CONDITIONS
(Note 12) TYP (Note 12) UNITS
REFERENCE AND SOFT-START/ENABLE
Internal Reference Voltage
Soft-Start Current
V
0.594
-
10
-
0.606
V
µA
V
REF
I
-
1.0
-
-
-
SS
Soft-Start Threshold
V
SOFT
Enable Low
-
1.0
V
(Converter Disabled)
PWM CONTROLLER GATE DRIVERS
Gate Drive Peak Current
Rise Time
-
-
-
-
0.7
20
20
20
-
-
-
-
A
Co = 1000pF
Co = 1000pF
ns
ns
ns
Fall Time
Dead Time Between Drivers
ERROR AMPLIFIER
DC Gain (Note 10)
-
-
88
15
-
-
dB
Gain-Bandwidth Product
(Note 10)
GBW
MHz
Slew Rate (Note 10)
SR
-
6
-
V/µs
mA
COMP Souce/Sink Current
(Note 10)
±0.4
OVERCURRENT PROTECTION
OCSET Current Source
I
V
= 4.5V
OCSET
80
100
120
µA
OCSET
POWER-GOOD AND CONTROL FUNCTIONS
Power-Good Lower Threshold
Power-Good Higher Threshold
PGOOD Leakage Current
PGOOD Voltage Low
V
Fraction of the set point; ~3µs noise filter
Fraction of the set point; ~3µs noise filter
-14
-10
-
-8
16
1
%
%
µA
V
PG-
V
9
-
-
-
-
-
PG+
I
V
= 5.0V (Note 11)
= 4mA
-
PGLKG
PULLUP
I
-
0.5
-
PGOOD
PGOOD Delay
CDEL = 0.1µF
125
2
ms
µA
V
CDEL Current for PGOOD
CDEL Threshold
CDEL threshold = 2.5V
-
2.5
-
EXTERNAL REFERENCE
Min External Reference Input at
GPIO1/REFIN
V
V
= H, C
= H, C
= 2.2µF
= 2.2µF
-
-
0.600
-
-
V
V
MSET/MODE
MSET/MODE
REFOUT
REFOUT
Max External Reference Input
at GPIO1/REFIN
1.250
REFERENCE BUFFER
Buffered Output Voltage -
Internal Reference
V
V
V
V
I
C
= 1mA,V
= High,
0.583
0.575
1.227
1.219
0.595
0.587
1.246
1.238
0.607
0.599
1.265
1.257
V
V
V
V
REFOUT
REFOUT
REFOUT
REFOUT
REFOUT
REFOUT
MSET/MODE
= 2.2µF, T = -40°C to +85°C
A
Buffered Output Voltage -
Internal Reference
I
C
= 20mA,V
= High,
REFOUT
REFOUT
MSET/MODE
= 2.2µF, T = -40°C to +85°C
A
Buffered Output Voltage -
External Reference
V
V
= 1.25V, I
= 1mA,
REFOUT
REFIN
= High, C
= 2.2µF
MSET2/MODE
REFOUT
Buffered Output Voltage -
External Reference
V
V
= 1.25V, I
= 20mA,
= 2.2µF
REFIN
REFOUT
= High,C
MSET2/MODE
REFOUT
FN9169 Rev 4.00
December 4, 2009
Page 8 of 21
ISL6420A
Electrical Specifications
Operating Conditions: VIN = 12V, PVCC shorted with VCC5, T = +25°C. Boldface limits
apply over the operating temperature range, -40°C to +85°C. (Continued)
A
MIN
MAX
PARAMETER
Current Drive Capability
VOLTAGE MARGINING
SYMBOL
TEST CONDITIONS
(Note 12) TYP (Note 12) UNITS
C
= 2.2µF
20
-
-
mA
REFOUT
Voltage Margining Range
(Note 10)
-10
-
-
+10
-
%
CDEL Current for Voltage
Margining
100
µA
Slew Time
CDEL = 0.1µF, VMSET = 330k
-
-
2.5
-
-
ms
µA
ISET1 on FB Pin
VMSET = 330k,
GPIO1 = L
7.48
GPIO2 = H
ISET2 on FB Pin
VMSET = 330k,
GPIO1 = H
-
7.48
-
µA
GPIO2 = L
THERMAL SHUTDOWN
Shutdown Temperature
(Note 10)
-
-
150
20
-
-
°C
°C
Thermal Shutdown Hysteresis
(Note 10)
NOTES:
7. The operating supply current and shutdown current specifications for 5V input are the same as VIN supply current specifications,
i.e., 5.6V to 28V input conditions. These should also be tested with part configured for 5V input configuration, i.e., VIN = VCC5
= PVCC = 5V.
8. This is the V
CC
current consumed when the device is active but not switching. Does not include gate drive current.
9. When the input voltage is 5.6V to 28V at VIN pin, the VCC5 pin provides a 5V output capable of 50mA (max) total from the
internal LDO. When the input voltage is 5V, VCC5 pin will be used as a 5V input, the internal LDO regulator is disabled and the
VIN must be connected to the VCC5. In both cases the PVCC pin should always be connected to VCC5 pin (refer to “Pin
Descriptions” on page 11 for more details).
10. Limits established by characterization and are not production tested.
11. It is recommended to use VCC5 as the pull-up source.
12. Parameters with MIN and/or MAX limits are 100% tested at +25°C, unless otherwise specified. Temperature limits established
by characterization and are not production tested.
FN9169 Rev 4.00
December 4, 2009
Page 9 of 21
ISL6420A
Typical Performance Curves
0.604
320
310
300
290
280
270
0.602
0.600
0.598
0.596
0.594
-40
-15
10
35
60
85
-40
-15
10
35
60
85
TEMPERATURE (°C)
TEMPERATURE (°C)
FIGURE 1. V
vs TEMPERATURE
FIGURE 2. V
vs TEMPERATURE
REF
SW
94
92
90
88
86
84
82
80
1.15
I
= 5A
OUT
1.05
0.95
0.85
-40
-15
10
35
60
85
0
5
10
15
VIN (V)
20
25
30
TEMPERATURE (°C)
FIGURE 3. I
vs TEMPERATURE
FIGURE 4. EFFICIENCY vs VIN
OCSET
+25°C, VIN = 28V, I = 1.367, I
IN
= 10A
OUT
FIGURE 5.
FIGURE 6.
FN9169 Rev 4.00
December 4, 2009
Page 10 of 21
ISL6420A
Typical Performance Curves (Continued)
98
96
94
VIN = 5V
VIN = 12V
92
90
88
86
84
82
80
0
1
2
3
4
5
6
7
8
9
10
LOAD (A)
FIGURE 7. EFFICIENCY vs LOAD CURRENT (V
= 3.3V)
OUT
1400
1300
1200
1100
Pin Descriptions
VIN
This pin powers the controller and must be decoupled
to ground using a ceramic capacitor as close as
possible to the VIN pin.
1000
900
800
700
600
500
400
300
200
100
0
TABLE 1. INPUT SUPPLY CONFIGURATION
INPUT
PIN CONFIGURATION
5.5V to Connect the input to the VIN pin. The VCC5 pin
28V
will provide a 5V output from the internal LDO.
Connect PVCC to VCC5.
5V ±10% Connect the input to the VCC5 pin. Connect the
PVCC and VIN pins to VCC5.
0
25
50
75
100
125
150
RT (k)
FIGURE 8. OSCILLATOR FREQUENCY vs RT
SGND
This pin provides the signal ground for the IC. Tie this
pin to the ground plane through the lowest impedance
connection.
FB
This pin is connected to the feedback resistor divider
and provides the voltage feedback signal for the
controller. This pin sets the output voltage of the
converter.
LGATE
This pin provides the PWM-controlled gate drive for
the lower MOSFET.
COMP
PHASE
This pin is the error amplifier output pin. It is used as
the compensation point for the PWM error amplifier.
This pin is the junction point of the output filter
inductor, the upper MOSFET source and the lower
MOSFET drain. This pin is used to monitor the voltage
drop across the upper MOSFET for overcurrent
protection. This pin also provides a return path for the
upper gate drive.
PGOOD
This pin provides a power good status. It is an open
collector output used to indicate the status of the
output voltage.
UGATE
RT
This pin provides the PWM-controlled gate drive for
the upper MOSFET.
This is the oscillator frequency selection pin.
Connecting this pin directly to VCC5 will select the
oscillator free running frequency of 300kHz. By placing
a resistor from this pin to GND, the oscillator frequency
can be programmed from 100kHz to 1.4MHz. Figure 8
shows the oscillator frequency vs. the RT resistance.
BOOT
This pin powers the upper MOSFET driver. Connect this
pin to the junction of the bootstrap capacitor and the
cathode of the bootstrap diode. The anode of the
bootstrap diode is connected to the PVCC pin.
FN9169 Rev 4.00
December 4, 2009
Page 11 of 21
ISL6420A
CDEL
GPIO1/REFIN
The PGOOD signal can be delayed by a time
This is a dual function pin. If VMSET/MODE is not
connected to VCC5 then this pin serves as GPIO1.
Refer to Table 3 for GPIO commands interpretation.
proportional to a CDEL current of 2µA and the value of
the capacitor connected between this pin and ground.
A 0.1µF will typically provide 125ms delay. When in the
Voltage Margining mode, the CDEL current is 100µA
typical and provides the delay for the output voltage
slew rate, 2.5ms typical for the 0.1µF capacitor.
To use GPIO1/REFIN as input reference, connect
VMSET/MODE to VCC5 and GPIO2 to SGND. Connect
the desired reference voltage to the GPIO1/REFIN pin
in the range of 0.6V to 1.25V.
Connect GPIO1/REFIN and VMSET/MODE pins to
VCC5, GPIO2 to SGND, the IC operates with the
internal reference and no voltage margining function.
PGND
This pin provides the power ground for the IC. Tie this
pin to the ground plane through the lowest impedance
connection.
REFOUT
It provides buffered reference output for REFIN.
Connect 2.2µF decoupling capacitor to this pin.
PVCC
This pin is the power connection for the gate drivers.
Connect this pin to the VCC5 pin.
VMSET/MODE
This pin is a dual function pin. Tie this pin to VCC5 to
disable voltage margining. When not tied to VCC5, this
pin serves as VMSET. Connect a resistor from this pin
to ground to set delta for voltage margining.
VCC5
This pin is the output of the internal 5V LDO. Connect a
minimum of 4.7µF ceramic decoupling capacitor as
close to the IC as possible at this pin. Refer to Table 1.
If voltage margining and external reference tracking
mode are not needed, VNSET/MODE, GPIO1/REFIN
and GPIO2 all together can be tied directly to ground.
ENSS
This pin provides enable/disable function and soft-start
for the PWM output. The output drivers are turned off
when this pin is held below 1V.
GPIO2
This is general purpose IO pin for voltage margining.
Refer to Table 3.
OCSET
Connect a resistor (R ) and a capacitor from this
pin to the drain of the upper MOSFET. R , an
internal 100µA current source (I
MOSFET on resistance r
OCSET
Exposed Thermal Pad
OCSET
), and the upper
This pad is electrically isolated. Connect this pad to the
signal ground plane using at least five vias for a robust
thermal conduction path.
OCSET
set the converter
DS(ON)
overcurrent (OC) trip point.
TABLE 2. VOLTAGE MARGINING/DDR OR TRACKING SUPPLY PIN CONFIGURATION
PIN CONFIGURATIONS
FUNCTION/MODES
VMSET/MODE
REFOUT
GPIO1/REFIN
GPIO2
Enable Voltage Margining
Pin Connected to
Connect a 2.2µF
Serves as a general
Serves as a general
GND with resistor. It capacitor for bypass of purpose I/O. Refer to purposeI/O. Referto
is used as VMSET.
external reference.
Table 3.
Table 3.
No Voltage Margining. Normal
operation with internal reference.
H
Connect a 2.2µF
capacitor to GND.
H (Note 14)
L
L
Buffered V
= 0.6V.
REFOUT
No Voltage Margining. External
reference. Buffered
H
Connect a 2.2µF
capacitor to GND.
Connect to an
external reference
voltage source (0.6V
to 1.25V)
V
= V
REFOUT
REFIN
NOTES:
13. The GPIO1/REFIN and GPIO2 pins cannot be left floating.
14. Ensure that GPIO1/REFIN is tied high prior to the logic change at VMSET/MODE.
FN9169 Rev 4.00
December 4, 2009
Page 12 of 21
ISL6420A
TABLE 3. VOLTAGE MARGINING CONTROLLED BY
GPIO1 AND GPIO2
V
OUT
GPIO1
GPIO2
VOUT
No Change
+VOUT
-VOUT
L
L
L
H
L
I
OUT
PHASE
H
H
H
Ignored
ENSS
Functional Description
Initialization
FIGURE 10. TYPICAL OVERCURRENT HICCUP MODE
The ISL6420A automatically initializes upon receipt of
power. The Power-On Reset (POR) function monitors
the internal bias voltage generated from LDO output
(VCC5) and the ENSS pin. The POR function initiates
the soft-start operation after the VCC5 exceeds the
POR threshold. The POR function inhibits operation
with the chip disabled (ENSS pin <1V).
Overcurrent Protection
The overcurrent function protects the converter from
a shorted output by using the upper MOSFET’s
ON-resistance, r
method enhances the converter’s efficiency and
to monitor the current. This
DS(ON)
reduces cost by eliminating a current sensing resistor.
The device can operate from an input supply voltage of
5.5V to 28V connected directly to the VIN pin using the
internal 5V linear regulator to bias the chip and supply
the gate drivers. For 5V ±10% applications, connect
VIN to VCC5 to bypass the linear regulator.
The overcurrent function cycles the soft-start function
in a hiccup mode to provide fault protection. A resistor
connected to the drain of the upper FET and the
OCSET pin programs the overcurrent trip level. The
PHASE node voltage will be compared against the
voltage on the OCSET pin, while the upper FET is on.
A current (100µA typically) is pulled from the OCSET
pin to establish the OCSET voltage. If PHASE is lower
than OCSET while the upper FET is on then an
overcurrent condition is detected for that clock cycle.
The upper gate pulse is immediately terminated, and
a counter is incremented. If an overcurrent condition
is detected for 8 consecutive clock cycles, and the
circuit is not in soft-start, the ISL6420A enters into
the soft-start hiccup mode. During hiccup, the
external capacitor on the ENSS pin is discharged.
After the capacitor is discharged, it is released and a
soft-start cycle is initiated. There are three dummy
soft-start delay cycles to allow the MOSFETs to cool
down, to keep the average power dissipation in hiccup
mode at an acceptable level. At the fourth soft-start
cycle, the output starts a normal soft-start cycle, and
the output tries to ramp.
Soft-Start/Enable
The ISL6420A soft-start function uses an internal
current source and an external capacitor to reduce
stresses and surge current during start-up.
When the output of the internal linear regulator
reaches the POR threshold, the POR function initiates
the soft-start sequence. An internal 10µA current
source charges an external capacitor on the ENSS pin
linearly from 0V to 3.3V.
When the ENSS pin voltage reaches 1V typically, the
internal 0.6V reference begins to charge following the
dv/dt of the ENSS voltage. As the soft-start pin
charges from 1V to 1.6V, the reference voltage charges
from 0V to 0.6V. Figure 9 shows a typical soft-start
sequence.
VIN = 28V, V
OUT
= 3.3V, I = 10A
OUT
During soft-start, pulse termination current limiting is
enabled, but the 8-cycle hiccup counter is held in reset
until soft-start is completed. Figure 10 shows the
overcurrent hiccup mode.
The overcurrent function will trip at a peak inductor
current (I ) determined from Equation 1, where
OC
I
is the internal OCSET current source.
OCSET
I
R
OCSET
OCSET
(EQ. 1)
---------------------------------------------------
I
=
OC
r
DSON
The OC trip point varies mainly due to the upper
MOSFETs r variations. To avoid overcurrent
DS(ON)
tripping in the normal operating load range, find the
resistor from Equation 1 with:
R
OCSET
FIGURE 9. TYPICAL SOFT-START WAVEFORM
FN9169 Rev 4.00
December 4, 2009
Page 13 of 21
ISL6420A
1. The maximum r
temperature.
at the highest junction
DS(ON)
VIN = 12V, V
OUT
= 3.3V, NO LOAD
2. Determine I
for I
I
+ I 2 ,
OUTMAX
OC
OC
where I is the output inductor ripple current.
A small ceramic capacitor should be placed in parallel
with R
to smooth the voltage across R
in
OCSET
OCSET
the presence of switching noise on the input voltage.
Voltage Margining
The ISL6420A has a voltage margining mode that can
be used for system testing. The voltage margining
percentage is resistor selectable up to ±10%. The
voltage margining mode can be enabled by connecting
a margining set resistor from VMSET pin to ground and
using the control pins GPIO1/2 to toggle between
positive and negative margining (Refer to Table 2).
With voltage margining enabled, the VMSET resistor to
ground will set a current, which is switched to the FB
pin. The current will be equal to 2.468V divided by the
value of the external resistor tied to the VMSET pin.
Use a resistor in the range of 150k to 400k for
VMSET resistor.
FIGURE 12A.
VIN = 12V, V
= 3.3V, NO LOAD
OUT
2.468V
-----------------------
I
=
(EQ. 2)
VM
R
VMSET
R
FB
-----------------------
V
= 2.468V
VM
R
(EQ. 3)
VMSET
The power supply output increases when GPIO2 is
HIGH and decreases when GPIO1 is HIGH. The amount
that the output voltage of the power supply changes
with voltage margining, will be equal to 2.468V x the
ratio of the external feedback resistor and the external
resistor tied to VMSET. Figure 11 shows the positive
and negative margining for a 3.3V output, using a
20.5k feedback resistor and using various VMSET
resistor values.
FIGURE 12B.
3.7
3.6
3.5
3.4
3.3
3.2
3.1
VIN = 12V, V
OUT
= 3.3V, I = 10A
OUT
3.0
2.9
2.8
150 175 200 225 250 275 300 325 350 375 400
RVMSET (k)
FIGURE 11. VOLTAGE MARGINING vs. VMSET
RESISTANCE
FIGURE 13. PGOOD DELAY
FN9169 Rev 4.00
December 4, 2009
Page 14 of 21
ISL6420A
The slew time of the current is set by an external
capacitor on the CDEL pin, which is charged and
discharged with a 100µA current source. The change in
voltage on the capacitor is 2.5V. This same capacitor is
used to set the PGOOD active delay after soft-start.
When PGOOD is low, the internal PGOOD circuitry uses
the capacitor and when PGOOD is high, the voltage
margining circuit uses the capacitor. The slew time for
voltage margining can be in the range of 300µs to
2ms.
Undervoltage
If the voltage on the FB pin is less than 15% of the
reference voltage for 8 consecutive PWM cycles, then
the circuit enters into soft-start hiccup mode. This
mode is identical to the overcurrent hiccup mode.
Overvoltage Protection
If the voltage on the FB pin exceeds the reference
voltage by 15%, the lower gate driver is turned on
continuously to discharge the output voltage. If the
overvoltage condition continues for 32 consecutive
PWM cycles, then the chip is turned off with the gate
drivers tri-stated. The voltage on the FB pin will fall
and reach the 15% undervoltage threshold. After 8
clock cycles, the chip will enter soft-start hiccup
mode. This mode is identical to the overcurrent hiccup
mode.
External Reference/DDR Supply
The voltage margining can be disabled by connecting
the VMSET/MODE to VCC5. In this mode, the chip can
be configured to work with an external reference input
and provide a buffered reference output.
If VMSET/MODE pin and the GPIO1/REFIN pin are both
tied to VCC5, then the internal 0.6V reference is used
as the error amplifier non-inverting input. The buffered
reference output on REFOUT will be 0.6V ±0.01V,
capable of sourcing 20mA and sinking up to 50µA
current with a 2.2µF capacitor connected to the
REFOUT pin.
Gate Control Logic
The gate control logic translates generated PWM
control signals into the MOSFET gate drive signals
providing necessary amplification, level shifting and
shoot-through protection. Also, it has functions that
help optimize the IC performance over a wide range of
operational conditions.
If the VMSET/MODE pin is tied to high but
GPIO1/REFIN is connected to an external voltage
source between 0.6V to 1.25V, then this external
voltage is used as the reference voltage at the positive
input of the error amplifier. The buffered reference
output on REFOUT will be Vrefin ±0.01V, capable of
sourcing 20mA and sinking up to 50µA current with a
2.2µF capacitor on the REFOUT pin.
Since MOSFET switching time can vary dramatically
from type to type and with the input voltage, the gate
control logic provides adaptive dead time by
monitoring the gate-to-source voltages of both upper
and lower MOSFETs. The lower MOSFET is not turned
on until the gate-to-source voltage of the upper
MOSFET has decreased to less than approximately 1V.
Similarly, the upper MOSFET is not turned on until the
gate-to-source voltage of the lower MOSFET has
decreased to less than approximately 1V. This allows a
wide variety of upper and lower MOSFETs to be used
without a concern for simultaneous conduction, or
shoot-through.
Power-Good
The PGOOD pin can be used to monitor the status of
the output voltage. PGOOD will be true (open drain)
when the FB pin is within ±10% of the reference and
the ENSS pin has completed its soft-start ramp.
Additionally, a capacitor on the CDEL pin will set a
delay for the PGOOD signal. After the ENSS pin
completes its soft-start ramp, a 2µA current begins
charging the CDEL capacitor to 2.5V. The capacitor will
be quickly discharged before PGOOD goes high. The
programmable delay can be used to sequence multiple
converters or as a LOW-true reset signal.
Start-up into Pre-Biased Load
The ISL6420A is designed to power-up into a
pre-biased load. This is achieved by transitioning from
Diode Emulation mode to a Forced Continuous
Conduction mode during start-up. The lower gate turns
ON for a short period of time and the voltage on the
phase pin is sensed. When this goes negative the lower
gate is turned OFF and remains OFF till the next cycle.
As a result, the inductor current will not go negative
during soft-start and thus will not discharge the
pre-biased load. The waveform for this condition is
shown in Figure 14.
If the voltage on the FB pin exceeds ±10% of the
reference, then PGOOD will go low after 1µs of noise
filtering.
Over-Temperature Protection
The IC is protected against over-temperature
conditions. When the junction temperature exceeds
+150°C, the PWM shuts off. Normal operation is
resumed when the junction temperature is cooled
down to +130°C.
Shutdown
When ENSS pin is below 1V, the regulator is disabled
with the PWM output drivers tri-stated. When disabled,
the IC power will be reduced.
FN9169 Rev 4.00
December 4, 2009
Page 15 of 21
ISL6420A
VIN = 12V, V
= 3.3V at 25mA LOAD
OUT
VIN
ISL6420A
UGATE
Q1
Q2
L
O
V
OUT
PHASE
C
IN
C
D2
O
LGATE
GND
RETURN
FIGURE 15. PRINTED CIRCUIT BOARD POWER AND
GROUND PLANES OR ISLANDS
FIGURE 14. PREBIASED OUTPUT AT 25mA LOAD
Application Guidelines
Layout Considerations
+VIN
BOOT
As in any high frequency switching converter, layout is
very important. Switching current from one power
device to another can generate voltage transients
across the impedances of the interconnecting bond
wires and circuit traces. These interconnecting
impedances should be minimized by using wide, short
printed circuit traces. The critical components should
be located as close together as possible using ground
plane construction or single point grounding.
D
1
Q
1
L
O
C
BOOT
PHASE
+5V
CC
V
OUT
ISL6420A
ENSS
C
Q
O
2
V
C
VCC
C
SS
GND
Figure 15 shows the critical power components of the
converter. To minimize the voltage overshoot the
interconnecting wires indicated by heavy lines should
be part of ground or power plane in a printed circuit
board. The components shown in Figure 15 should be
located as close together as possible. Please note that
FIGURE 16. PRINTED CIRCUIT BOARD SMALL
SIGNAL LAYOUT GUIDELINES
Feedback Compensation
Figure 17 highlights the voltage-mode control loop for
a synchronous-rectified buck converter. The output
the capacitors C and C each represent numerous
IN
O
physical capacitors. Locate the ISL6420A within 3
inches of the MOSFETs, Q and Q . The circuit traces
1
2
voltage (V
) is regulated to the Reference voltage
OUT
for the MOSFETs’ gate and source connections from the
ISL6420A must be sized to handle up to 1A peak
current.
level. The error amplifier (Error Amp) output (V ) is
E/A
compared with the oscillator (OSC) triangular wave to
provide a pulse-width modulated (PWM) wave with an
amplitude of VIN at the PHASE node. The PWM wave
Figure 16 shows the circuit traces that require
additional layout consideration. Use single point and
ground plane construction for the circuits shown.
Minimize any leakage current paths on the ENSS PIN
is smoothed by the output filter (L and C ).
O
O
The modulator transfer function is the small-signal
transfer function of V /V . This function is
OUT E/A
dominated by a DC Gain and the output filter (L and
and locate the capacitor, C close to the SS pin
ss
O
because the internal current source is only 10µA.
C ), with a double pole break frequency at F and a
O
LC
Provide local V
decoupling between VCC and GND
pins. Locate the capacitor, C as close as practical
CC
zero at F
. The DC Gain of the modulator is simply
ESR
BOOT
the input voltage (VIN) divided by the peak-to-peak
oscillator voltage V
to the BOOT and PHASE pins.
.
OSC
FN9169 Rev 4.00
December 4, 2009
Page 16 of 21
ISL6420A
Compensation Break Frequency Equations
VIN
OSC
DRIVER
DRIVER
1
----------------------------------
F
=
PWM
(EQ. 6)
(EQ. 7)
Z1
P1
2 R2 C1
L
O
COMPARATOR
V
OUT
-
PHASE
1
+
V
------------------------------------------------------
F
=
OSC
C
O
C1 C2
C1 + C2
----------------------
2 R2
ESR
(PARASITIC)
Z
FB
1
-----------------------------------------------------
F
F
=
V
(EQ. 8)
(EQ. 9)
E/A
Z2
P2
2 R1 + R3 C3
Z
-
IN
+
REFERENCE
ERROR
AMP
1
----------------------------------
=
2 R3 C3
1. Pick Gain (R2/R1) for desired converter bandwidth
ST
DETAILED COMPENSATION COMPONENTS
2. Place 1
(~75% F
Zero Below Filter’s Double Pole
)
LC
ND
Z
FB
V
OUT
3. Place 2
4. Place 1
5. Place 2
Zero at Filter’s Double Pole
Pole at the ESR Zero
C
2
Z
IN
ST
C
C
R
1
3
R
3
ND
2
Pole at Half the Switching Frequency
6. Check Gain against Error Amplifier’s Open-Loop
Gain
R
1
COMP
FB
7. Estimate Phase Margin - Repeat if Necessary
-
+
R
Figure 18 shows an asymptotic plot of the DC/DC
converter’s gain vs. frequency. The actual Modulator
Gain has a high gain peak due to the high Q factor of
the output filter and is not shown in Figure 18. Using
the previously mentioned guidelines should give a
Compensation Gain similar to the curve plotted. The
open loop error amplifier gain bounds the
4
ISL6420A
REF
R
1
V
= V
1 + ------
OUT
REF
R
4
FIGURE 17. VOLTAGE - MODE BUCK CONVERTER
COMPENSATION DESIGN
compensation gain. Check the compensation gain at
F
with the capabilities of the error amplifier. The Loop
P2
Gain is constructed on the log-log graph of Figure 18
by adding the Modulator Gain (in dB) to the
Compensation Gain (in dB). This is equivalent to
multiplying the modulator transfer function to the
compensation transfer function and plotting the gain.
Modulator Break Frequency Equations
1
--------------------------------------
F
=
(EQ. 4)
LC
2
L
C
O
O
1
100
--------------------------------------------
F
=
(EQ. 5)
F
F
P1
F
ESR
F
P2
Z2
Z1
2 ESR C
O
80
60
OPEN LOOP
ERROR AMP GAIN
The compensation network consists of the error
amplifier (internal to the ISL6420A) and the
40
impedance networks Z and Z . The goal of the
20LOG
(R2/R1)
IN FB
compensation network is to provide a closed loop
transfer function with the highest 0dB crossing
20
20LOG
(VIN/V
)
OSC
0
frequency (f
) and adequate phase margin. Phase
0dB
COMPENSATION
GAIN
margin is the difference between the closed loop phase
at f and 180°. The following equations relate the
MODULATOR
GAIN
-20
-40
-60
0dB
compensation network’s poles, zeros and gain to the
components (R , R , R , C , C , and C ) in Figure 17.
CLOSED LOOP
GAIN
F
LC
F
ESR
100k
FREQUENCY (Hz)
1
2
3
1
2
3
Use the following guidelines for locating the poles and
10
100
1k
10k
1M
10M
zeros of the compensation network.
FIGURE 18. ASYMPTOTIC BODE PLOT OF CONVERTER
GAIN
FN9169 Rev 4.00
December 4, 2009
Page 17 of 21
ISL6420A
The compensation gain uses external impedance
current and the output capacitors ESR. The ripple
voltage and current are approximated by Equations 10
and 11:
networks Z and Z to provide a stable, high
FB IN
bandwidth (BW) overall loop. A stable control loop has
a gain crossing with -20dB/decade slope and a phase
margin greater than 45°. Include worst case
V
- V
V
OUT OUT
IN
------------------------------- ---------------
I =
(EQ. 10)
L
Fs x L
V
IN
component variations when determining phase margin.
V
= I ESR
(EQ. 11)
OUT
L
Component Selection
Guidelines
Increasing the value of inductance reduces the ripple
current and voltage. However, larger inductance values
reduce the converter’s response time to a load
transient.
Output Capacitor Selection
An output capacitor is required to filter the output and
supply the load transient current. The filtering
requirements are a function of the switching frequency
and the ripple current. The load transient requirements
are a function of the slew rate (di/dt) and the
magnitude of the transient load current. These
requirements are generally met with a mix of
capacitors and careful layout.
One of the parameters limiting the converter’s
response to a load transient is the time required to
change the inductor current. Given a sufficiently fast
control loop design, the ISL6420A will provide either
0% or 100% duty cycle in response to a load transient.
The response time is the time required to slew the
inductor current from an initial current value to the
transient current level. During this interval the
difference between the inductor current and the
transient current level must be supplied by the output
capacitor. Minimizing the response time can minimize
the output capacitance required.
Modern microprocessors produce transient load rates
above 1A/ns. High frequency capacitors initially supply
the transient and slow the current load rate seen by
the bulk capacitors. The bulk filter capacitor values are
generally determined by the ESR (effective series
resistance) and voltage rating requirements rather
than actual capacitance requirements.
The response time to a transient is different for the
application of load and the removal of load. Equations
12 and 13 give the approximate response time interval
for application and removal of a transient load:
High frequency decoupling capacitors should be placed
as close to the power pins of the load as physically
possible. Be careful not to add inductance in the circuit
board wiring that could cancel the usefulness of these
low inductance components. Consult with the
manufacturer of the load on specific decoupling
requirements. For example, Intel recommends that the
high frequency decoupling for the Pentium Pro be
composed of at least forty (40) 1.0µF ceramic
capacitors in the 1206 surface-mount package.
L
I
TRAN
O
-------------------------------
t
=
(EQ. 12)
(EQ. 13)
RISE
FALL
V
– V
OUT
IN
L
I
O
TRAN
------------------------------
t
=
V
OUT
where: I
TRAN
the response time to the application of load, and t
is the transient load current step, t
RISE
is
FALL
is the response time to the removal of load. With a +5V
input source, the worst case response time can be
either at the application or removal of load and
dependent upon the output voltage setting. Be sure to
check both of these equations at the minimum and
maximum output levels for the worst case response
time.
Use only specialized low-ESR capacitors intended
for switching-regulator applications for the bulk
capacitors. The bulk capacitor’s ESR will determine
the output ripple voltage and the initial voltage drop
after a high slew-rate transient. An aluminum
electrolytic capacitor's ESR value is related to the
case size with lower ESR available in larger case
sizes. However, the equivalent series inductance (ESL)
of these capacitors increases with case size and can
reduce the usefulness of the capacitor to high slew-rate
transient loading. Unfortunately, ESL is not a specified
parameter. Work with your capacitor supplier and
measure the capacitor’s impedance with frequency to
select a suitable component. In most cases, multiple
electrolytic capacitors of small case size perform better
than a single large case capacitor.
Input Capacitor Selection
Use a mix of input bypass capacitors to control the
voltage overshoot across the MOSFETs. Use small
ceramic capacitors for high frequency decoupling and
bulk capacitors to supply the current needed each
time Q turns on. Place the small ceramic capacitors
1
physically close to the MOSFETs and between the
drain of Q and the source of Q .
1
2
The important parameters for the bulk input capacitor
are the voltage rating and the RMS current rating. For
reliable operation, select the bulk capacitor with
voltage and current ratings above the maximum input
voltage and largest RMS current required by the circuit.
The capacitor voltage rating should be at least 1.25 x
Output Inductor Selection
The output inductor is selected to meet the output
voltage ripple requirements and minimize the
converter’s response time to the load transients. The
inductor value determines the converter’s ripple
current and the ripple voltage is a function of the ripple
FN9169 Rev 4.00
December 4, 2009
Page 18 of 21
ISL6420A
greater than the maximum input voltage and a voltage
rating of 1.5 x is a conservative guideline. The RMS
current rating requirement for the input capacitor of a
buck regulator is approximately 1/2 the DC load current.
Equation 14 shows a more specific formula for
determining the input ripple:
Schottky Selection
Rectifier D is a clamp that catches the negative
2
inductor swing during the dead time between turning off
the lower MOSFET and turning on the upper MOSFET.
The diode must be a Schottky type to prevent the
parasitic MOSFET body diode from conducting. It is
acceptable to omit the diode and let the body diode of
the lower MOSFET clamp the negative inductor swing,
but efficiency will drop one or two percent as a result.
The diode's rated reverse breakdown voltage must be
greater than the maximum input voltage.
2
I
= I
MAX
D – D
(EQ. 14)
RMS
For a through hole design, several electrolytic capacitors
(Panasonic HFQ series or Nichicon PL series or Sanyo
MV-GX or equivalent) may be needed. For surface mount
designs, solid tantalum capacitors can be used, but
caution must be exercised with regard to the capacitor
surge current rating. These capacitors must be capable
of handling the surge-current at power-up. The TPS
series available from AVX, and the 593D series from
Sprague are both surge current tested.
MOSFET Selection/Considerations
The ISL6420A requires 2 N-Channel power MOSFETs.
These should be selected based upon r
, gate
DS(ON)
supply requirements, and thermal management
requirements.
In high-current applications, the MOSFET power
dissipation, package selection and heatsink are the
dominant design factors. The power dissipation includes
two loss components; conduction loss and switching loss.
The conduction losses are the largest component of
power dissipation for both the upper and the lower
MOSFETs. These losses are distributed between the two
MOSFETs according to duty factor (see Equations 15 and
16). Only the upper MOSFET has switching losses, since
the Schottky rectifier clamps the switching node before
the synchronous rectifier turns on.
2
O
1
2
--
P
P
= I r
D +
I
V t f
IN sw sw
(EQ. 15)
UFET
LFET
DSON
DSON
O
2
= I r
1 – D
(EQ. 16)
O
Where D is the duty cycle = Vo/VIN, t
SW
is the switching
is the switching frequency.
interval, and f
SW
These equations assume linear voltage-current
transitions and do not adequately model power loss due
the reverse recovery of the lower MOSFET’s body diode.
The gate-charge losses are dissipated by the ISL6420A
and don't heat the MOSFETs. However, large
gate-charge increases the switching interval, t
which
SW
increases the upper MOSFET switching losses. Ensure
that both MOSFETs are within their maximum junction
temperature at high ambient temperature by calculating
the temperature rise according to package
thermal-resistance specifications. A separate heatsink
may be necessary depending upon MOSFET power,
package type, ambient temperature and air flow.
FN9169 Rev 4.00
December 4, 2009
Page 19 of 21
ISL6420A
Package Outline Drawing
L20.4x4
20 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 1, 11/06
4X
2.0
4.00
0.50
16X
A
6
B
16
20
PIN #1 INDEX AREA
6
PIN 1
INDEX AREA
1
15
2 . 10 ± 0 . 15
11
5
(4X)
0.15
6
10
0.10 M
C
A B
4
0.25 +0.05 / -0.07
TOP VIEW
20X 0.6 +0.15 / -0.25
BOTTOM VIEW
SEE DETAIL "X"
C
0.10
0 . 90 ± 0 . 1
C
BASE PLANE
( 3. 6 TYP )
(
SEATING PLANE
0.08 C
( 20X 0 . 5 )
2. 10 )
SIDE VIEW
0 . 2 REF
( 20X 0 . 25 )
( 20X 0 . 8)
5
C
0 . 00 MIN.
0 . 05 MAX.
TYPICAL RECOMMENDED LAND PATTERN
DETAIL "X"
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
Tiebar shown (if present) is a non-functional feature.
5.
6.
The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 indentifier may be
either a mold or mark feature.
FN9169 Rev 4.00
December 4, 2009
Page 20 of 21
ISL6420A
Shrink Small Outline Plastic Packages (SSOP)
Quarter Size Outline Plastic Packages (QSOP)
M20.15
N
20 LEAD SHRINK SMALL OUTLINE PLASTIC PACKAGE
(0.150” WIDE BODY)
INDEX
AREA
0.25(0.010M) B M
H
E
GAUGE
PLANE
INCHES
MIN
MILLIMETERS
SYM-
BOL
-B-
MAX
0.069
0.010
0.061
0.012
0.010
0.344
0.157
MIN
1.35
0.10
-
MAX NOTES
A
A1
A2
B
0.053
0.004
-
1.75
0.25
1.54
0.30
0.25
8.74
3.98
-
1
2
3
-
L
0.25
0.010
SEATING PLANE
A
-
-A-
0.008
0.007
0.337
0.150
0.20
0.18
8.56
3.81
9
D
h x 45°
C
-
-C-
D
E
3
4
A2
e
A1
C
e
0.025 BSC
0.635 BSC
-
B
0.10(0.004)
H
h
0.228
0.0099
0.016
0.244
0.0196
0.050
5.80
0.26
0.41
6.19
0.49
1.27
-
0.17(0.007M) C A M B S
5
NOTES:
L
6
1. Symbols are defined in the “MO Series Symbol List” in Sec-
tion 2.2 of Publication Number 95.
N
20
20
7
0°
8°
0°
8°
-
2. Dimensioning and tolerancing per ANSI Y14.5M-1982.
Rev. 1 6/04
3. Dimension “D” does not include mold flash, protrusions or
gate burrs. Mold flash, protrusion and gate burrs shall not
exceed 0.15mm (0.006 inch) per side.
4. Dimension “E” does not include interlead flash or protru-
sions. Interlead flash and protrusions shall not exceed
0.25mm (0.010 inch) per side.
5. The chamfer on the body is optional. If it is not present, a
visual index feature must be located within the cross-
hatched area.
6. “L” is the length of terminal for soldering to a substrate.
7. “N” is the number of terminal positions.
8. Terminal numbers are shown for reference only.
9. Dimension “B” does not include dambar protrusion. Allow-
able dambar protrusion shall be 0.10mm (0.004 inch) total
in excess of “B” dimension at maximum material condition.
10. Controlling dimension: INCHES. Converted millimeter di-
mensions are not necessarily exact.
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Intersil products are sold by description only. Intersil may modify the circuit design and/or specifications of products at any time without notice, provided that such
modification does not, in Intersil's sole judgment, affect the form, fit or function of the product. Accordingly, the reader is cautioned to verify that datasheets are
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For information regarding Intersil Corporation and its products, see www.intersil.com
FN9169 Rev 4.00
December 4, 2009
Page 21 of 21
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