AFE5803_14 [TI]
Fully Integrated, 8-Channel Ultrasound Analog Front End, 0.75 nV/rtHz, 14/12-Bit, 65 MSPS, 158 mW/CH;型号: | AFE5803_14 |
厂家: | TEXAS INSTRUMENTS |
描述: | Fully Integrated, 8-Channel Ultrasound Analog Front End, 0.75 nV/rtHz, 14/12-Bit, 65 MSPS, 158 mW/CH |
文件: | 总52页 (文件大小:789K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
AFE5803
www.ti.com
SLOS763A –JANUARY 2012–REVISED JANUARY 2012
Fully Integrated, 8-Channel Ultrasound Analog Front End,
0.75 nV/rtHz, 14/12-Bit, 65 MSPS, 158 mW/CH
Check for Samples: AFE5803
1
FEATURES
DESCRIPTION
The AFE5803 is a highly integrated Analog Front-End
•
8-Channel Complete Analog Front-End
LNA, VCAT, PGA, LPF, ADC
(AFE) solution specifically designed for ultrasound
systems in which high performance and small size
are required. The AFE5803 integrates a complete
time-gain-control (TGC) imaging path. It also enables
users to select one of various power/noise
combinations to optimize system performance.
Therefore, the AFE5803 is a suitable ultrasound
analog front end solution for portable systems.
–
•
Programmable Gain Low-Noise Amplifier
(LNA)
–
–
–
–
24/18/12 dB Gain
0.25/0.5/1 VPP Linear Input Range
0.63/0.7/0.9 nV/rtHz Input Referred Noise
Programmable Active Termination
The AFE5803 contains eight channels of voltage
controlled amplifier (VCA), 14/12-bit Analog-to-Digital
Converter (ADC). The VCA includes Low noise
Amplifier (LNA), Voltage controlled Attenuator
(VCAT), Programmable Gain Amplifier (PGA), and
Low-Pass Filter (LPF). The LNA gain is
programmable to support 250 mVPP to 1 VPP input
signals. Programmable active termination is also
supported by the LNA. The ultra-low noise VCAT
provides an attenuation control range of 40 dB and
improves overall low gain SNR which benefits
harmonic imaging and near field imaging. The PGA
provides gain options of 24 dB and 30 dB. Before the
ADC, a LPF can be configured as 10 MHz, 15 MHz,
20 MHz or 30 MHz to support ultrasound applications
with different frequencies. The high-performance
14bit/65 MSPS ADC in the AFE5803 achieves
77dBFS SNR. It ensures excellent SNR at low chain
gain. The ADC’s LVDS outputs enable flexible system
integration desired for miniaturized systems.
•
40 dB Low Noise Voltage Controlled
Attenuator (VCAT)
•
•
24/30 dB Programmable Gain Amplifier (PGA)
3rd Order Linear Phase Low-Pass Filter (LPF)
–
10, 15, 20, 30 MHz
•
•
•
14-bit Analog to Digital Converter (ADC)
–
–
77 dBFS SNR at 65 MSPS
LVDS Outputs
Noise/Power Optimizations (Full Chain)
–
–
158 mW/CH at 0.75 nV/rtHz, 65 MSPS
101 mW/CH at 1.1 nV/rtHz, 40 MSPS
Excellent Device-to-Device Gain Matching
±0.5 dB (Typical) and ±0.9 dB (Max)
–
•
•
•
Low Harmonic Distortion
Fast and Consistent Overload Recovery
Small Package: 15 mm x 9 mm, 135-BGA
The AFE5803 is available in a 15mm × 9mm, 135-pin
BGA package and it is specified for operation from
0°C to 85°C. It is also pin-to-pin compatible to the
AFE5807, AFE5808 and AFE5808A.
APPLICATIONS
•
•
Medical Ultrasound Imaging
Nondestructive Evaluation Equipments
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2012, Texas Instruments Incorporated
AFE5803
SLOS763A –JANUARY 2012–REVISED JANUARY 2012
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
AFE5803 (1 of 8 Channels)
SPI OUT
SPI IN
SPI Logic
3rd LP Filter
10, 15, 20,
30 MHz
PGA
24, 30dB
VCAT
0 to -40dB
14Bit
ADC
LVDS
LNA
LNA IN
Reference
Reference
Differential
TGC Vcntl
EXT/INT
REFs
Figure 1. Block Diagram
PACKAGING/ORDERING INFORMATION(1)
TRANSPORT MEDIA,
QUANTITY
PRODUCT
AFE5803
PACKAGE TYPE
OPERATING
ORDERING NUMBER
ZCF
0°C to 85°C
AFE5803ZCF
Tray, 160
(1) For the most current package and ordering information see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
2
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SLOS763A –JANUARY 2012–REVISED JANUARY 2012
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted)(1)
VALUE
UNIT
MIN
–0.3
–0.3
–0.3
–0.3
–0.3
–0.3
MAX
3.9
2.2
6
AVDD
V
V
AVDD_ADC
AVDD_5V
DVDD
Supply voltage
range
V
2.2
0.3
V
Voltage between AVSS and LVSS
Voltage at analog inputs and digital inputs
Peak solder temperature(2)
V
min [3.6,AVDD+0.3]
V
260
105
150
85
°C
°C
°C
°C
V
Maximum junction temperature (TJ), any condition
Storage temperature range
–55
Operating temperature range
0
Human Body Model (HBM)
ESD Ratings
2000
500
Charged Device Model (CDM)
V
(1) Stresses above those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied Exposure to absolute maximum rated conditions for extended periods may degrade device reliability.
(2) Device complies with JSTD-020D.
THERMAL INFORMATION
AFE5803
THERMAL METRIC(1)
BGA
135 PINS
34.1
5
UNITS
θJA
Junction-to-ambient thermal resistance
θJCtop
θJB
Junction-to-case (top) thermal resistance
Junction-to-board thermal resistance
11.5
0.2
°C/W
ψJT
Junction-to-top characterization parameter
Junction-to-board characterization parameter
Junction-to-case (bottom) thermal resistance
ψJB
10.8
n/a
θJCbot
(1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
RECOMMENDED OPERATING CONDITIONS
PARAMETER
MIN
3.15
1.7
1.7
4.75
0
MAX
3.6
1.9
1.9
5.5
85
UNIT
V
AVDD
AVDD_ADC
DVDD
V
V
AVDD_5V
V
Ambient Temperature, TA
°C
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DEVICE INFORMATION
PIN CONFIGURATION
Top View
ZCF (BGA-135)
1
AVDD
2
INP8
3
INP7
4
5
6
INP4
7
INP3
8
INP2
9
A
B
C
D
E
F
INP6
INP5
INP1
CM_BYP
ACT8
ACT7
ACT6
INM6
ACT5
INM5
AVSS
AVSS
AVSS
AVSS
AVSS
AVSS
VHIGH
DNC
ACT4
INM4
ACT3
INM3
ACT2
ACT1
INM1
AVDD
AVDD
DNC
AVSS
INM8
INM7
INM2
AVSS
AVSS
AVSS
AVSS
AVSS
AVSS
AVSS
AVSS
VCNTLP
AVDD_ADC
VREF_IN
DVDD
D5M
AVSS
AVSS
AVSS
AVSS
AVSS
AVSS
VCNTLM
REFM
REFP
DNC
AVSS
AVSS
AVSS
AVSS
AVSS
AVDD_ADC
AVSS
DNC
AVSS
AVSS
AVSS
AVSS
AVSS
AVDD_ADC
DNC
AVDD
CH7_TEST_OUTP
CH7_BUFFER_OUTM
AVSS
CH7_TEST_OUTM
CH7_BUFFER_OUTP
AVSS
AVDD
DNC
G
H
J
DNC
DNC
CH8_BUFFER_OUTM
CH8_TEST_OUTP
AVDD
CH8_BUFFER_OUTP
CH8_TEST_OUTM
AVDD_5V
CLKM_ADC
AVDD_ADC
D8M
PDN_GLOBAL
PDN_VCA
AVDD_ADC
PDN_ADC
DNC
RESET
SCLK
SDATA
SEN
K
L
CLKP_ADC
AVDD_ADC
D8P
DNC
M
N
P
R
DNC
DNC
DNC
SDOUT
D1P
DVSS
DVSS
DVSS
DNC
DVDD
D4M
D1M
D7M
D6M
FCLKM
FCLKP
DCLKM
DCLKP
D3M
D2M
D7P
D6P
D5P
D4P
D3P
D2P
PIN FUNCTIONS
PIN
DESCRIPTION
NO.
NAME
ACT1...ACT8
B9~ B2
Active termination input pins for CH1~8. 1 μF capacitors are recommended. See the Applicaiton
Information section.
A1, D8, D9, E8, AVDD
E9, K1
3.3 V Analog supply for LNA, VCAT, PGA, LPF blocks.
K2
AVDD_5V
5 V Analog supply for LNA, VCAT, PGA, LPF blocks.
1.8 V Analog power supply for ADC.
J6, J7, K8, L3, AVDD_ADC
M1, M2
C1, D1~D7,
E3~E7, F3~F7,
G1~G7,
H3~H7,J3~J5,
K6
AVSS
Analog ground.
Negative input of differential ADC clock. In the single-end clock mode, it can be tied to GND directly or
through a 0.1 µF capacitor.
L2
L1
B1
CLKM_ADC
CLKP_ADC
CM_BYP
Positive input of differential ADC clock. In the single-end clock mode, it can be tied to clock signal
directly or through a 0.1 µF capacitor.
Bias voltage and bypass to ground. ≥1µF is recommended. To suppress the ultra low frequency noise,
10µF can be used.
E2
E1
CH7_TEST_OUTM
CH7_TEST_OUTP
CH7 PGA negative output when PGA test mode is enabled. Can be floated if not used.
CH7 PGA positive output when PGA test mode is enabled. Can be floated if not used.
Negative differential output for the buffer amplifier when PGA test mode is enabled. Can be floated if not
used. See the TEST MODES in the application information section.
F1
F2
CH7_BUFFER_OUTM
CH7_BUFFER_OUTP
Positive differential output for the buffer amplifier when PGA test mode is enabled. Can be floated if not
used. See the TEST MODES in the application information section.
J2
J1
CH8_TEST_OUTM
CH8_TEST_OUTP
CH8 PGA negative output when PGA test mode is enabled. Can be floated if not used.
CH8 PGA positive output when PGA test mode is enabled. Can be floated if not used.
Negative differential output for the buffer amplifier when PGA test mode is enabled. Can be floated if not
used. See the TEST MODES in the application information section.
H1
H2
CH8_BUFFER_OUTM
CH8_BUFFER_OUTP
D1M~D8M
Positive differential output for the buffer amplifier when PGA test mode is enabled. Can be floated if not
used. See the TEST MODES in the application information section.
N8, P9~P7,
P3~P1, N2
ADC CH1~8 LVDS negative outputs
N9, R9~R7,
R3~R1, N1
D1P~D8P
DCLKM
ADC CH1~8 LVDS positive outputs
LVDS bit clock (7x) negative output
P6
4
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PIN FUNCTIONS (continued)
PIN
DESCRIPTION
NO.
NAME
R6
DCLKP
LVDS bit clock (7x) positive output
Do not connect. Must leave floated
F8, F9, G8, G9,
K7,
L5~L7,M5~M8,
N4, N6
DNC
N3, N7
N5, P5, R5
P4
DVDD
DVSS
ADC digital and I/O power supply, 1.8 V
ADC digital ground
FCLKM
FCLKP
LVDS frame clock (1X) negative output
LVDS frame clock (1X) positive output
R4
CH1~8 complimentary analog inputs. Bypass to ground with ≥ 0.015 µF capacitors. The HPF response
of the LNA depends on the capacitors.
C9~C2
INM1…INM8
A9~A2
L8
INP1...INP8
PDN_ADC
PDN_VCA
CH1~8 analog inputs. AC couple to inputs with ≥ 0.1µF capacitors.
ADC partial (fast) power down control pin with an internal pull down resistor of 100 kΩ. Active High.
VCA partial (fast) power down control pin with an internal pull down resistor of 20 kΩ. Active High.
J8
Global (complete) power-down control pin for the entire chip with an internal pull down resistor of 20kΩ.
Active High.
H8
L4
PDN_GLOBAL
REFM
0.5 V reference output in the internal reference mode. Must leave floated in the internal reference mode.
Adding test point on PCB is recommended for monitoring the reference output.
1.5 V reference output in the internal reference mode. Must leave floated in the internal reference mode.
Adding test point on PCB is recommended for monitoring the reference output.
M4
REFP
H9
J9
RESET
SCLK
Hardware reset pin with an internal pull-down resistor of 20 kΩ. Active high.
Serial interface clock input with an internal pull-down resistor of 2 0kΩ
Serial interface data input with an internal pull-down resistor of 20 kΩ
Serial interface data readout. High impedance when readout is disabled.
Serial interface enable with an internal pull up resistor of 20 kΩ. Active low.
Negative differential attenuation control pin.
K9
M9
L9
SDATA
SDOUT
SEN
K4
K3
K5
M3
VCNTLM
VCNTLP
VHIGH
VREF_IN
Positive differential attenuation control pin
Bias voltage; bypass to ground with ≥1µF.
ADC 1.4 V reference input in the external reference mode; bypass to ground with 0. 1 µF.
F8, F9, G8, G9,
K7, L5~L7,
M5~M8, N4,
N6
DNC
Do not connect. Must leave floated
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ELECTRICAL CHARACTERISTICS
AVDD_5 V = 5 V, AVDD = 3.3 V, AVDD_ADC = 1.8 V, DVDD = 1.8 V, AC-coupled with 0.1µF at INP and bypassed to ground
with 15nF at INM, No active termination, VCNTL = 0 V, fIN = 5 MHz, LNA = 18 dB, PGA = 24 dB, 14Bit, sample rate = 65
MSPS, LPF Filter = 15 MHz, low noise mode, VOUT = –1 dBFS, ADC configured in internal reference mode, single-ended
VCNTL mode, VCNTLM = GND, at ambient temperature TA = 25°C, unless otherwise noted. Min and max values are
specified across full-temperature range with AVDD_5 V = 5 V, AVDD = 3.3 V, AVDD_ADC = 1.8 V, DVDD = 1.8 V
PARAMETER
TEST CONDITION
MIN
TYP
MAX
UNITS
TGC FULL SIGNAL CHANNEL (LNA+VCAT+LPF+ADC)
Rs = 0 Ω, f = 2 MHz, LNA =24/18/12 dB, PGA = 2 4dB
Rs = 0 Ω, f = 2 MHz,LNA =24/18/12 dB, PGA = 30 dB
Rs = 0 Ω, f = 2 MHz,LNA =24/18/12 dB, PGA = 24 dB
Rs = 0 Ω, f = 2 MHz, LNA =24/18/12 dB, PGA = 30 dB
Rs = 0 Ω, f = 2 MHz,LNA = 24/18/12 dB, PGA = 24 dB
Rs = 0 Ω, f = 2 MHz, LNA = 24/18/12 dB, PGA = 30 dB
Low Noise Mode/Medium Power Mode/Low Power Mode
Rs = 200 Ω, 200 Ω active termination, PGA = 24dB,LNA = 12/18/24 dB
Rs = 100 Ω, 100 Ω active termination, PGA = 24dB,LNA = 12/18/24 dB
LNA gain = 24/18/12 dB
0.76/0.83/1.16
0.75/0.86/1.12
1.1/1.2/1.45
1.1/1.2/1.45
1/1.05/1.25
0.95/1.0/1.2
2.7/2.1/2
Input voltage noise over LNA Gain(low
noise mode)
nV/rtHz
nV/rtHz
nV/rtHz
Input voltage noise over LNA Gain(low
power mode)
en (RTI)
Input Voltage Noise over LNA
Gain(Medium Power Mode)
Input referred current noise
pA/rtHz
dB
3.85/2.4/1.8
5.3/3.1/2.3
250/500/1000
350/600/1150
24/30
NF
Noise figure
dB
VMAX
Maximum Linear Input Voltage
Clamp Voltage
mVPP
dB
VCLAMP
Reg52[10:9] = 0, LNA = 24/18/12 dB
Low noise mode
PGA Gain
Medium/Low power mode
24/28.5
LNA = 24 dB, PGA = 30 dB, Low noise mode
LNA = 2 4dB, PGA = 30 dB, Med power mode
LNA = 24 dB, PGA = 30 dB, Low power mode
54
Total gain
52.5
dB
52.5
Ch-CH Noise Correlation Factor without
Signal
Summing of 8 channels
0
(1)
Full band (VCNTL = 0/0.8)
0.15/0.17
0.18/0.75
70
Ch-CH Noise Correlation Factor with
Signal
(1)
1MHz band over carrier (VCNTL= 0/0.8)
VCNTL = 0.6V(22 dB total channel gain)
VCNTL = 0, LNA = 18 dB, PGA =24 dB
VCNTL = 0, LNA = 24 dB, PGA = 24 dB
68
Signal to Noise Ratio (SNR)
59.3
63
dBFS
dBFS
58
SNR over 2 MHz band around carrier at VCNTL = 0.6 V ( 22 dB total
gain)
Narrow Band SNR
75
77
Input Common-mode Voltage
At INP and INM pins
2.4
V
kΩ
Ω
8
50/100/200/400
20
Input resistance
Preset active termination enabled
Input capacitance
pF
V
Input Control Voltage
Common-mode voltage
Gain Range
VCNTLP - VCNTLM
0
1.5
VCNTLP and VCNTLM
0.75
V
-40
dB
dB/V
KΩ
pF
µs
Gain Slope
VCNTL= 0.1 V to 1.1 V
35
Input Resistance
Between VCNTLP and VCNTLM
Between VCNTLP and VCNTLM
VCNT L= 0 V to 1.5 V step function
200
Input Capacitance
1
TGC Response Time
3rd order-Low-pass Filter
Settling time for change in LNA gain
1.5
10, 15, 20, 30
14
MHz
µs
Settling time for change in active
termination setting
1
µs
(1) Noise correlation factor is defined as Nc/(Nu+Nc), where Nc is the correlated noise power in single channel; and Nu is the uncorrelated
noise power in single channel. Its measurement follows the below equation, in which the SNR of single channel signal and the SNR of
summed eight channel signal are measured.
8CH_SNR
-
10
N
10
1
1
7
C
=
x
-
1CH_SNR
10
Nu + N
56
-
C
10
6
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ELECTRICAL CHARACTERISTICS (continued)
AVDD_5 V = 5 V, AVDD = 3.3 V, AVDD_ADC = 1.8 V, DVDD = 1.8 V, AC-coupled with 0.1µF at INP and bypassed to ground
with 15nF at INM, No active termination, VCNTL = 0 V, fIN = 5 MHz, LNA = 18 dB, PGA = 24 dB, 14Bit, sample rate = 65
MSPS, LPF Filter = 15 MHz, low noise mode, VOUT = –1 dBFS, ADC configured in internal reference mode, single-ended
VCNTL mode, VCNTLM = GND, at ambient temperature TA = 25°C, unless otherwise noted. Min and max values are
specified across full-temperature range with AVDD_5 V = 5 V, AVDD = 3.3 V, AVDD_ADC = 1.8 V, DVDD = 1.8 V
PARAMETER
TEST CONDITION
MIN
TYP
MAX
UNITS
AC ACCURACY
LPF Bandwidth tolerance
CH-CH group delay variation
CH-CH Phase variation
±5%
2
2 MHz to 15 MHz
15 MHz signal
ns
11
Degree
0 V < VCNTL < 0.1 V (Dev-to-Dev)
0.1 V < VCNTL < 1.1 V(Dev-to-Dev)
0.1 V < VCNTL < 1.1 V(Dev-to-Dev) Temp = 0°C and 85°C
1.1 V < VCNTL < 1.5 V(Dev-to-Dev)
Channel-to-Channel
±0.5
±0.5
±0.5
±0.5
±0.25
–0.9
–1.1
0.9
1.1
Gain matching
dB
Gain matching
Output offset
dB
Vcntl= 0, PGA = 30 dB, LNA = 24 dB
–75
75
LSB
AC PERFORMANCE
Fin = 2 MHz; VOUT = -1 dBFS
Fin = 5 MHz; VOUT = -1 dBFS
–60
–60
Fin = 5 MHz; VIN= 500 mVpp,
VOUT = –1dBFS, LNA = 18dB, VCNTL=0.88 V
HD2
Second-Harmonic Distortion
dBc
–55
–55
Fin = 5 MHz; Vin = 250 mVpp,
VOUT =–1 dBFS, LNA = 24dB, VCNTL= 0.88 V
Fin = 2 MHz; VOUT = –1dBFS
Fin = 5 MHz; VOUT = –1dBFS
–55
–55
Fin = 5 MHz; VIN = 500 mVpp,
VOUT = –1 dBFS, LNA = 18 dB, VCNTL= 0.88 V
HD3
THD
Third-Harmonic Distortion
dBc
–55
–55
Fin = 5 MHz; VIN = 2 50 mVpp,
VOUT = –1 dBFS, LNA = 24 dB, VCNTL= 0.88 V
Fin = 2 MHz; VOUT=–1 dBFS
Fin = 5 MHz; VOUT=–1 dBFS
–55
–55
Total Harmonic Distortion
Intermodulation distortion
dBc
dBc
f1 = 5 MHz at –1dBFS,
f2 = 5.01 MHz at –27 dBFS
IMD3
–60
XTALK
Cross-talk
Fin = 5 MHz; VOUT= –1 dBFS
–65
dB
Phase Noise
1 kHz off 5 MHz (VCNTL=0V)
–132
dBc/Hz
LNA
Input Referred Voltage Noise
High-Pass Filter
Rs = 0 Ω, f = 2MHz, Rin = High Z, Gain = 24/18/12 dB
0.63/0.70/0.9
50/100/150/200
4
nV/rtHz
KHz
-3 dB Cut-off Frequency
LNA linear output
VPP
VCAT+ PGA
VCAT Input Noise
0dB/-40 dB Attenuation
24 dB/30 dB
2/10.5
1.75
80
nV/rtHz
nV/rtHz
KHz
PGA Input Noise
-3dB HPF cut-off Frequency
ADC SPECIFICATIONS
Sample rate
10
65
MSPS
dBFS
V
SNR
Signal-to-noise ratio
Idle channel SNR of ADC 14b
REFP
77
1.5
0.5
1.4
50
Internal reference mode
External reference mode
REFM
V
VREF_IN Voltage
VREF_IN Current
V
µA
ADC input full-scale range
LVDS Rate
2
Vpp
Mbps
65MSPS at 14 bit
910
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ELECTRICAL CHARACTERISTICS (continued)
AVDD_5 V = 5 V, AVDD = 3.3 V, AVDD_ADC = 1.8 V, DVDD = 1.8 V, AC-coupled with 0.1µF at INP and bypassed to ground
with 15nF at INM, No active termination, VCNTL = 0 V, fIN = 5 MHz, LNA = 18 dB, PGA = 24 dB, 14Bit, sample rate = 65
MSPS, LPF Filter = 15 MHz, low noise mode, VOUT = –1 dBFS, ADC configured in internal reference mode, single-ended
VCNTL mode, VCNTLM = GND, at ambient temperature TA = 25°C, unless otherwise noted. Min and max values are
specified across full-temperature range with AVDD_5 V = 5 V, AVDD = 3.3 V, AVDD_ADC = 1.8 V, DVDD = 1.8 V
PARAMETER
POWER DISSIPATION
AVDD Voltage
TEST CONDITION
MIN
TYP
MAX
UNITS
3.15
1.7
3.3
1.8
5
3.6
1.9
5.5
1.9
190
V
V
V
V
AVDD_ADC Voltage
AVDD_5V Voltage
DVDD Voltage
4.75
1.7
1.8
158
145
114
101.5
202
126
99
TGC low noise mode, 65 MSPS
TGC low noise mode, 40 MSPS
TGC medium power mode, 40 MSPS
TGC low power mode, 40 MSPS
TGC low noise mode, no signal
TGC medium power mode, no signal
TGC low power mode, no signal
TGC low noise mode, 500 mVPP Input,1% duty cycle
TGC medium power mode, 500 mVPP Input, 1% duty cycle
TGC low power, 500 mVPP Input, 1% duty cycle
TGC mode no signal
Total power dissipation per channel
mW/CH
240
AVDD (3.3V) Current
AVDD_5V Current
mA
mA
210
133
105
25.5
16.5
99
35
TGC mode, 500 mVPP Input,1% duty cycle
TGC low noise mode, no signal
TGC medium power mode, no signal
TGC low power mode, no signal
TGC low noise mode, 500 mVPP input,1% duty cycle
TGC medium power mode, 500 mVPP Input, 1% duty cycle
TGC low power mode, 500 mVPP input,1% duty cycle
65 MSPS
121
68
55.5
102.5
71
VCA Power dissipation
mW/CH
59.5
187
77
AVDD_ADC(1.8V) Current
DVDD(1.8V) Current
205
110
69
mA
mA
65 MSPS
65 MSPS
59
50 MSPS
51
ADC Power dissipation/CH
mW/CH
mW/CH
40 MSPS
46
20 MSPS
35
Power dissipation in power down mode
PDN_VCA = High, PDN_ADC = High
Complete power-down PDN_Global=High
Time taken to enter power down
VCA power down
25
0.6
1
Power-down response time
Power-up response time
µs
µs
2µs+1% of PDN
time
ADC power down
1
Complete power down
2.5
ms
fin = 5 MHz, at 50 mVpp noise at 1 KHz on supply(2)
fin = 5 MHz, at 50 mVpp noise at 50 KHz on supply(2)
f = 10 kHz,VCNTL = 0 V (high gain), AVDD
f = 10 kHz,VCNTL = 0 V(high gain), AVDD_5V
f = 10 kHz,VCNTL = 1 V (low gain), AVDD
–65
–65
–40
–55
–50
dBc
dBc
dBc
dBc
dBc
Power supply modulation ratio, AVDD and
AVDD_5V
Power supply rejection ratio
(2) PSMR specification is with respect to input signal amplitude.
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SLOS763A –JANUARY 2012–REVISED JANUARY 2012
DIGITAL CHARACTERISTICS
Typical values are at 25°C, AVDD = 3.3 V, AVDD_5 = 5 V and AVDD_ADC = 1.8 V, DVDD = 1.8 V, 14 bit sample rate = 65
MSPS, unless otherwise noted. Minimum and maximum values are across the full temperature range: TMIN = 0°C to TMAX
=
+85°C
PARAMETER
CONDITION
MIN
TYP
MAX UNITS(1)
DIGITAL INPUTS/OUTPUTS
VIH
VIL
Logic high input voltage
Logic low input voltage
Logic high input current
Logic low input current
Input capacitance
2
0
3.3
0.3
V
V
200
200
5
µA
µA
pF
V
VOH
VOL
Logic high output voltage
Logic low output voltage
SDOUT pin
SDOUT pin
DVDD
0
V
LVDS OUTPUTS
with 100 ohms external differential
termination
Output differential voltage
400
mV
Output offset voltage
FCLKP and FCLKM
DCLKP and DCLKM
Common-mode voltage
1X clock rate
1100
mV
MHz
MHz
MHz
ps
10
70
60
65
455
390
7X clock rate
6X clock rate
tsu
th
Data setup time(2)
Data hold time(2)
350
350
ps
ADC INPUT CLOCK
CLOCK frequency
10
45%
0.5
65
MSPS
Clock duty cycle
50%
55%
Sine-wave, ac-coupled
LVPECL, ac-coupled
LVDS, ac-coupled
biased internally
Vpp
Vpp
Vpp
V
Clock input amplitude,
differential(VCLKP_ADC–VCLKM_ADC
1.6
0.7
1
)
Common-mode voltage
Clock input amplitude VCLKP_ADC
(single-ended)
CMOS CLOCK
1.8
Vpp
(1) The DC specifications refer to the condition where the LVDS outputs are not switching, but are permanently at a valid logic level 0 or 1
with 100Ω external termination.
(2) Setup and hold time specifications take into account the effect of jitter on the output data and clock. These specifications also assume
that the data and clock paths are perfectly matched within the receiver. Any mismatch in these paths within the receiver would appear
as reduced timing margins
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TYPICAL CHARACTERISTICS
AVDD_5 V = 5 V, AVDD = 3.3 V, AVDD_ADC = 1.8 V, DVDD = 1.8 V, ac-coupled with 0.1µF caps at INP and 1
5nF caps at INM, No active termination, VCNTL = 0 V, FIN = 5 MHz, LNA = 18 dB, PGA = 24 dB, 14 Bit, sample
rate = 65 MSPS, LPF Filter = 15 MHz, low noise mode, VOUT = -1dBFS, ADC is configured in internal reference
mode, single-ended VCNTL mode, VCNTLM = GND, at ambient temperature TA = 25°C, unless otherwise noted.
45
40
35
30
25
20
15
10
5
45
40
35
30
25
20
15
10
5
Low noise
Medium power
Low power
−40 deg C
25 deg C
85 deg C
0
0
0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5
Vcntl (V)
0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
Vcntl (V)
Figure 2. Gain vs. VCNTL, LNA = 18 dB and PGA = 24 dB
Figure 3. Gain Variation vs. Temperature, LNA = 18 dB and
PGA = 24 dB
9000
8000
7000
6000
5000
4000
3000
2000
1000
0
9000
8000
7000
6000
5000
4000
3000
2000
1000
0
Gain (dB)
Gain (dB)
G004
G005
Figure 4. Gain Matching Histogram,
VCNTL= 0.3V (34951 channels)
Figure 5. Gain Matching Histogram,
VCNTL = 0.6V (34951 channels)
8000
7000
6000
5000
4000
3000
2000
1000
0
120
110
100
90
80
70
60
50
40
30
20
10
0
Gain (dB)
ADC Output
G005
G058
Figure 6. Gain Matching Histogram,
VCNTL = 0.9V (34951 channels)
Figure 7. Output Offset Histogram, VCNTL = 0V (1247
channels)
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SLOS763A –JANUARY 2012–REVISED JANUARY 2012
TYPICAL CHARACTERISTICS (continued)
Impedance Magnitude Response
Impedance Phase Response
10
0
Open
Open
12000
10000
8000
6000
4000
2000
−10
−20
−30
−40
−50
−60
−70
−80
−90
500k
4.5M
8.5M
12.5M
16.5M
20.5M
500k
4.5M
8.5M
12.5M
16.5M
20.5M
Frequency (Hz)
Frequency (Hz)
Figure 8. Input Impedance without Active Termination
(Magnitude)
Figure 9. Input Impedance without Active Termination
(Phase)
Impedance Magnitude Response
Impedance Phase Response
500
10
50 Ohms
450
400
350
300
250
200
150
100
50
0
100 Ohms
200 Ohms
400 Ohms
−10
−20
−30
−40
−50
−60
−70
−80
−90
50 Ohms
100 Ohms
200 Ohms
400 Ohms
0
500k
4.5M
8.5M
12.5M
16.5M
20.5M
500k
4.5M
8.5M
12.5M
16.5M
20.5M
Frequency (Hz)
Frequency (Hz)
Figure 10. Input Impedance with Active Termination
(Magnitude)
Figure 11. Input Impedance with Active Termination
(Phase)
LNA INPUT HPF CHARECTERISTICS
5
10MHz
15MHz
20MHz
30MHz
3
0
0
−5
−3
−6
−9
−10
−15
−20
−25
−30
−12
−15
−18
−21
−24
−27
−30
01
00
11
10
0
10
20
30
40
50
60
10
100
500
Frequency (MHz)
Frequency (KHz)
Figure 12. Low-Pass Filter Response
Figure 13. LNA High-Pass Filter Response vs. Reg59[3:2]
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TYPICAL CHARACTERISTICS (continued)
HPF CHARECTERISTICS (LNA+VCA+PGA+ADC)
60
50
40
30
20
10
0
LNA 12 dB
LNA 18 dB
LNA 24 dB
5
0
−5
−10
−15
−20
−25
−30
−35
−40
0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
Vcntl (V)
10
100
500
Frequency (KHz)
Figure 14. Full Channel High-Pass Filter Response at
Default Register Setting
Figure 15. IRN, PGA = 24 dB and Low Noise Mode
3.5
70
LNA 12 dB
LNA 18 dB
LNA 24 dB
2.5
LNA 12 dB
LNA 18 dB
LNA 24 dB
3.0
60
50
40
30
20
10
0
2.0
1.5
1.0
0.5
0.0
0.0
0.1
0.2
0.3
0.4
0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
Vcntl (V)
Vcntl (V)
Figure 16. IRN, PGA = 24 dB and Low Noise Mode
Figure 17. IRN, PGA = 24 dB and Medium Power Mode
4.0
70
LNA 12 dB
LNA 18 dB
LNA 24 dB
LNA 12 dB
LNA 18 dB
LNA 24 dB
3.5
3.0
2.5
2.0
1.5
1.0
0.5
60
50
40
30
20
10
0
0.0
0.1
0.2
0.3
0.4
0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
Vcntl (V)
Vcntl (V)
Figure 18. IRN, PGA = 24 dB and Medium Power Mode
Figure 19. IRN, PGA = 24 dB and Low Power Mode
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SLOS763A –JANUARY 2012–REVISED JANUARY 2012
TYPICAL CHARACTERISTICS (continued)
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
220
210
LNA 12 dB
LNA 18 dB
LNA 24 dB
LNA 12 dB
LNA 18 dB
LNA 24 dB
190
170
150
130
110
90
70
50
30
0.0
0.1
0.2
0.3
0.4
0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
Vcntl (V)
Vcntl (V)
Figure 20. IRN, PGA = 24 dB and Low Power Mode
Figure 21. ORN, PGA = 24 dB and Low Noise Mode
300
340
LNA 12 dB
LNA 18 dB
LNA 24 dB
LNA 12 dB
LNA 18 dB
LNA 24 dB
320
300
280
260
240
220
200
180
160
140
120
100
80
280
260
240
220
200
180
160
140
120
100
80
60
60
40
40
0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
Vcntl (V)
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
Vcntl (V)
1
1.1 1.2
Figure 22. ORN, PGA = 24 dB and Medium Power Mode
Figure 23. ORN, PGA = 24 dB and Low Power Mode
1.5
1.4
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
180.0
160.0
140.0
120.0
100.0
80.0
60.0
0.3
40.0
1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0
1.0
3.0
5.0
7.0
9.0
11.0 12.0
Frequency (MHz)
Frequency (MHz)
Figure 24. IRN, PGA = 24 dB and Low Noise Mode
Figure 25. ORN, PGA = 24 dB and Low Noise Mode
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TYPICAL CHARACTERISTICS (continued)
75
70
65
60
55
75
70
65
60
55
24 dB PGA gain
30 dB PGA gain
24 dB PGA gain
30 dB PGA gain
0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
Vcntl (V)
0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2
Vcntl (V)
Figure 26. SNR, LNA = 18 dB and Low Noise Mode
Figure 27. SNR, LNA = 18 dB and Low Power Mode
73
9
Low noise
Low power
100 ohm act term
8
7
6
5
4
3
2
1
0
71
69
67
65
63
61
59
57
200 ohm act term
400 ohm act term
Without Termination
0
3
6
9
12 15 18 21 24 27 30 33 36 39 42
Gain (dB)
50
100
150
200
250
300
350
400
Source Impedence (Ω)
Figure 28. SNR vs. Different Power Modes
Figure 29. Noise Figure, LNA = 12 dB and Low Noise Mode
10
9
8
7
6
5
4
3
2
1
0
8
50 ohm act term
100 ohm act term
200 ohm act term
400 ohm act term
Without Termination
50 ohm act term
7
6
5
4
3
2
1
0
100 ohm act term
200 ohm act term
400 ohm act term
No Termination
50
100
150
200
250
300
350
400
50
100
150
200
250
300
350
400
Source Impedence (Ω)
Source Impedence (Ω)
Figure 30. Noise Figure, LNA = 18 dB and Low Noise Mode
Figure 31. Noise Figure, LNA = 24 dB and Low Noise Mode
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SLOS763A –JANUARY 2012–REVISED JANUARY 2012
TYPICAL CHARACTERISTICS (continued)
4.5
4
3
2
1
Low noise
Low power
Medium power
Low noise
Low power
Medium power
3.5
2.5
1.5
50
100
150
200
250
300
350
400
50
100
150
200
250
300
350
400
Source Impedence (Ω)
Source Impedence (Ω)
Figure 32. Noise Figure vs. Power Modes with 400 Ω
Termination
Figure 33. Noise Figure vs. Power Modes without
Termination
−50.0
−45
Low noise
Low noise
Low power
Medium power
Low power
Medium power
−55.0
−60.0
−65.0
−70.0
−75.0
−80.0
−50
−55
−60
−65
−70
−75
1
2
3
4
5
6
7
8
9
10
1
2
3
4
5
6
7
8
9
10
Frequency (MHz)
Frequency (MHz)
Figure 34. HD2 vs. Frequency, Vin = 500 mVPP and
VOUT = -1 dBFS
Figure 35. HD3 vs. Frequency, Vin = 500 mVpp and
VOUT = -1 dBFS
−40
−40
Low noise
Low power
Medium power
Low noise
Low power
Medium power
−45
−50
−55
−60
−65
−70
−75
−80
−85
−90
−50
−60
−70
−80
−90
6
12
18
24
30
36
6
12
18
24
30
36
Gain (dB)
Gain (dB)
Figure 36. HD2 vs. Gain, LNA = 12 dB and
PGA = 24 dB and VOUT = -1 dBFS
Figure 37. HD3 vs. Gain, LNA = 12 dB and PGA = 24 dB
and VOUT = -1 dBFS
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TYPICAL CHARACTERISTICS (continued)
−40
−50
−60
−70
−80
−90
−40
−50
−60
−70
−80
−90
Low noise
Low power
Medium power
Low noise
Low power
Medium power
12
18
24
30
36
42
12
18
24
30
36
42
Gain (dB)
Gain (dB)
Figure 38. HD2 vs. Gain, LNA = 18 dB and PGA = 24 dB
and VOUT = -1 dBFS
Figure 39. HD3 vs. Gain, LNA = 18 dB and PGA = 24 dB
and VOUT = -1 dBFS
−40
−40
Low noise
Low power
Low noise
Low power
−45
Medium power
Medium power
−50
−55
−60
−65
−70
−75
−80
−85
−90
−50
−60
−70
−80
−90
18
24
30
36
42
48
18
21
24
27
30
33
36
39
42
45
48
Gain (dB)
Gain (dB)
Figure 40. HD2 vs. Gain, LNA = 24 dB and PGA = 24 dB
and VOUT = -1 dBFS
Figure 41. HD3 vs. Gain, LNA = 24 dB and PGA = 24 dB
and VOUT = -1 dBFS
−50
−50
Fin1=2MHz, Fin2=2.01MHz
Fin1=2MHz, Fin2=2.01MHz
Fin1=5MHz, Fin2=5.01MHz
−54
Fin1=5MHz, Fin2=5.01MHz
−54
−58
−62
−66
−70
−58
−62
−66
−70
14
18
22
26
30
34
38
42
14
18
22
26
30
34
38
42
Gain (dB)
Gain (dB)
G001
G001
Figure 42. IMD3, Fout1 = -1 dBFS and Fout2 = -21 dBFS
Figure 43. IMD3, Fout1 = -7dBFS and Fout2 = -7dBFS
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SLOS763A –JANUARY 2012–REVISED JANUARY 2012
TYPICAL CHARACTERISTICS (continued)
PSMR vs SUPPLY FREQUENCY
PSMR vs SUPPLY FREQUENCY
−60
−65
−70
−75
−55
−60
−65
−70
−75
−80
Vcntl = 0
Vcntl = 0
Vcntl = 0.3
Vcntl = 0.6
Vcntl = 0.9
Vcntl = 0.3
Vcntl = 0.6
Vcntl = 0.9
5
10
100
1000 2000
5
10
100
1000 2000
Supply frequency (kHz)
Supply frequency (kHz)
Figure 44. AVDD Power Supply Modulation Ratio,
100 mVPP Supply Noise with Different Frequencies
Figure 45. AVDD_5V Power Supply Modulation Ratio,
100mVPP Supply Noise with Different Frequencies
3V PSRR vs SUPPLY FREQUENCY
5V PSRR vs SUPPLY FREQUENCY
−20
−20
Vcntl = 0
Vcntl = 0
Vcntl = 0.3
Vcntl = 0.6
Vcntl = 0.9
Vcntl = 0.3
Vcntl = 0.6
Vcntl = 0.9
−30
−40
−50
−60
−70
−80
−90
−30
−40
−50
−60
−70
−80
−90
5
10
100
1000 2000
5
10
100
1000 2000
Supply frequency (kHz)
Supply frequency (kHz)
Figure 46. AVDD Power Supply Rejection Ratio, 100mVPP
Supply Noise with Different Frequencies
Figure 47. AVDD_5 V Power Supply Rejection Ratio,
100mVPP Supply Noise with Different Frequencies
20000.0
1.3
1.2
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0.0
−0.1
20000.0
18000.0
16000.0
14000.0
12000.0
10000.0
8000.0
6000.0
4000.0
2000.0
0.0
1.3
1.2
Output Code
Vcntl
18000.0
16000.0
14000.0
12000.0
10000.0
8000.0
6000.0
4000.0
2000.0
0.0
1.1
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0.0
−0.1
Output Code
Vcntl
0.0
0.5
1.0
1.5
Time (µs)
2.0
2.5
3.0
0.0 0.2 0.5 0.8 1.0 1.2 1.5 1.8 2.0 2.2 2.5
Time (µs)
Figure 48. VCNTL Response Time, LNA = 18 dB
and PGA = 24 dB
Figure 49. VCNTL Response Time, LNA = 18 dB
and PGA = 24 dB
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TYPICAL CHARACTERISTICS (continued)
1.2
1.0
1.2
1.0
0.8
0.8
0.6
0.6
0.4
0.4
0.2
0.2
0.0
0.0
−0.2
−0.4
−0.6
−0.8
−1.0
−1.2
−0.2
−0.4
−0.6
−0.8
−1.0
−1.2
0.0 2.0 4.0 6.0 8.0 10.0 12.0 14.0 16.0 18.0 20.0
Time (µs)
0.0 2.0 4.0 6.0 8.0 10.0 12.0 14.0 16.0 18.0 20.0
Time (µs)
Figure 50. Pulse Inversion Asymmetrical Positive Input
Figure 51. Pulse Inversion Asymmetrical Negative Input
10000.0
10000
Positive overload
Negative overload
Average
47nF
15nF
8000.0
6000.0
4000.0
2000.0
0.0
8000
6000
4000
2000
0
−2000.0
−4000.0
−6000.0
−8000.0
−10000.0
−2000
−4000
−6000
−8000
−10000
0.0
1.0
2.0
3.0
4.0
5.0
6.0
0
0.5
1
1.5
2
2.5
3
3.5
4
4.5
5
Time (µs)
Time (µs)
Figure 52. Pulse Inversion, VIN = 2 VPP, PRF = 1 KHz, Gain
= 21 dB
Figure 53. Overload Recovery Response vs. INM
Capacitor, VIN = 50 mVPP/100 µVPP, Max Gain
10
2000
1600
47nF
15nF
5
0
1200
800
−5
−10
−15
−20
−25
−30
−35
−40
k=2
k=3
k=4
k=5
k=6
k=7
k=8
k=9
k=10
400
0
−400
−800
−1200
−1600
−2000
0
0.2 0.4 0.6 0.8
1
1.2 1.4 1.6 1.8
2
Frequency (MHz)
1
1.5
2
2.5
3
3.5
4
4.5
5
G000
Time (µs)
Figure 54. Overload Recovery Response vs. INM Capacitor
Figure 55. Digital High-Pass Filter Response
(Zoomed), VIN = 50 mVPP/100 µVPP, Max Gain
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TIMING CHARACTERISTICS(1)
Typical values are at 25°C, AVDD_5V = 5 V, AVDD = 3.3 V, AVDD_ADC = 1.8 V, DVDD = 1.8 V, Differential clock, CLOAD
=
5pF, RLOAD = 100 Ω, 14Bit, sample rate = 65MSPS, unless otherwise noted. Minimum and maximum values are across the
full temperature range TMIN = 0°C to TMAX = 85°C with AVDD_5V = 5 V, AVDD = 3.3 V, AVDD_ADC = 1.8 V, DVDD = 1.8 V
PARAMETER
TEST CONDITIONS
MIN TYP MAX
UNIT
The delay in time between the rising edge of the input sampling
clock and the actual time at which the sampling occurs
ta
tj
Aperture delay
0.7
3
ns
Aperture delay
matching
Across channels within the same device
±150
ps
Aperture jitter
450
Fs rms
Input
clock
ADC latency
Default, after reset, or / 0 x 2 [12] = 1, LOW_LATENCY = 1
11/8
cycles
Data and frame clock
delay
Input clock rising edge (zero cross) to frame clock rising edge (zero
cross) minus 3/7 of the input clock period (T).
tdelay
3
5.4
7
1
ns
Δtdelay
tRISE
Delay variation
At fixed supply and 20°C T difference. Device to device
–1
ns
ns
Rise time measured from –100 mV to 100 mV Fall time measured
from 100 mV to –100 mV 10 MHz < fCLKIN < 65 MHz
0.14
0.15
0.14
0.15
Data rise time Data fall
time
tFALL
tFCLKRISE
tFCLKFALL
Frame clock rise time
Frame clock fall time
Rise time measured from –100mV to 100mV Fall time measured
from 100 mV to –100 mV 10 MHz < fCLKIN < 65MHz
ns
ns
Frame clock duty cycle Zero crossing of the rising edge to zero crossing of the falling edge 48% 50% 52%
tDCLKRISE
tDCLKFALL
Rise time measured from –100mV to 100mV Fall time measured
from 100 mV to –100 mV 10 MHz < fCLKIN < 65MHz
0.13
0.12
Bit clock rise time Bit
clock fall time
Zero crossing of the rising edge to zero crossing of the falling edge
10 MHz < fCLKIN < 65 MHz
Bit clock duty cycle
46%
54%
(1) Timing parameters are ensured by design and characterization; not production tested.
OUTPUT INTERFACE TIMING(1)(2)(3)
Setup Time (tsu), ns
(for output data and frame clock)
Hold Time (th), ns
(for output data and frame clock)
tPROG = (3/7)x T + tdelay, ns
fCLKIN
,
Input Clock
Frequency
Input Clock Zero-Cross (rising edge)
to Frame Clock Zero-Cross (rising
edge)
Data Valid to Input Clock
Zero-Crossing
Input Clock Zero-Crossing to Data
Invalid
MHz
MIN
0.24
0.41
0.55
0.87
1.30
TYP
0.37
0.54
0.70
1.10
1.56
MAX
MIN
0.24
0.46
0.61
0.94
1.46
TYP
0.38
0.57
0.73
1.1
MAX
MIN
11
TYP
12
MAX
12.5
14.4
16.7
20.1
27.3
65/14bit
50/14bit
40/14bit
30/14bit
20/14bit
13
13.9
16
15
18.5
25.7
19.5
26.7
1.6
(1) FCLK timing is the same as for the output data lines. It has the same relation to DCLK as the data pins. Setup and hold are the same
for the data and the frame clock.
(2) Data valid is logic HIGH = +100mV and logic LOW = -100mV
(3) Timing parameters are ensured by design and characterization; not production tested.
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LVDS Setup and Hold Timing
12-Bit 6x serialization mode
14-Bit 7x serialization mode
Figure 56. LVDS Timing Diagrams
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LVDS Output Interface Description
AFE5803 has LVDS output interface which supports multiple output formats. The ADC resolutions can be
configured as 12bit or 14bit as shown in the LVDS timing diagrams Figure 56. The ADCs in the AFE5803 are
running at 14bit; 2 LSBs are removed when 12-bit output is selected; and two 0s are added at LSBs when 16-bit
output is selected. Appropriate ADC resolutions can be selected for optimizing system performance-cost
effectiveness. When the devices run at 16bit mode, higher end FPGAs are required to process higher rate of
LVDS data. Corresponding register settings are listed in Table 1.
Table 1. Corresponding Register Settings
LVDS Rate
Reg 3 [14:13]
Reg 4 [2:0]
Description
12 bit (6X DCLK)
14 bit (7X DCLK)
16 bit (8X DCLK)
11
010
00
01
000
000
N/A
2 LSBs removed
2 0s added at LSBs
SERIAL REGISTER TIMING
Serial Register Write Description
Programming of different modes can be done through the serial interface formed by pins SEN (serial interface
enable), SCLK (serial interface clock), SDATA (serial interface data) and RESET. All these pins have a pull-down
resistor to GND of 100kΩ. Serial shift of bits into the device is enabled when SEN is low. Serial data SDATA is
latched at every rising edge of SCLK when SEN is active (low). The serial data is loaded into the register at
every 24th SCLK rising edge when SEN is low. If the word length exceeds a multiple of 24 bits, the excess bits
are ignored. Data can be loaded in multiple of 24-bit words within a single active SEN pulse (there is an internal
counter that counts groups of 24 clocks after the falling edge of SEN). The interface can work with the SCLK
frequency from 20 MHz down to low speeds (few Hertz) and even with non-50% duty cycle SCLK. The data is
divided into two main portions: a register address (8 bits) and the data itself (16 bits), to load on the addressed
register. When writing to a register with unused bits, these should be set to 0. Figure 57 illustrates this process.
Start Sequence
End Sequence
SEN
t6
t7
t1
t2
Data Latched On Rising Edge of SCLK
SCLK
t3
SDATA
A7 A6 A5 A4 A3 A2 A1 A0 D15 D14 D13 D12 D11 D10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0
t4
t5
Start Sequence
End Sequence
RESET
T0384-01
Figure 57. SPI Timing
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SPI Timing Characteristics
Minimum values across full temperature range TMIN = 0°C to TMAX = 85°C, AVDD_5V =5.0V, AVDD=3.3V,
AVDD_ADC=1.8V, DVDD=1.8V
PARAMETER
DESCRIPTION
MIN
50
20
20
5
TYP
MAX
UNIT
ns
t1
t2
t3
t4
t5
t6
t7
t8
SCLK period
SCLK high time
SCLK low time
Data setup time
Data hold time
SEN fall to SCLK rise
ns
ns
ns
5
ns
8
ns
Time between last SCLK rising edge to SEN rising edge
SDOUT delay
8
ns
12
20
28
ns
Register Readout
The device includes an option where the contents of the internal registers can be read back. This may be useful
as a diagnostic test to verify the serial interface communication between the external controller and the AFE.
First, the <REGISTER READOUT ENABLE> bit (Reg0[1]) needs to be set to '1'. Then user should initiate a
serial interface cycle specifying the address of the register (A7-A0) whose content has to be read. The data bits
are "don’t care". The device will output the contents (D15-D0) of the selected register on the SDOUT pin.
SDOUT has a typical delay t8 of 20nS from the falling edge of the SCLK. For lower speed SCLK, SDOUT can be
latched on the rising edge of SCLK. For higher speed SCLK,e.g. the SCLK period lesser than 60nS, it would be
better to latch the SDOUT at the next falling edge of SCLK. The following timing diagram shows this operation
(the time specifications follow the same information provided. In the readout mode, users still can access the
<REGISTER READOUT ENABLE> through SDATA/SCLK/SEN. To enable serial register writes, set the
<REGISTER READOUT ENABLE> bit back to '0'.
Start Sequence
End Sequence
SEN
t6
t7
t1
t2
SCLK
SDATA
SDOUT
t3
A7 A6 A5 A4 A3 A2 A1 A0
x
x
x
x
x
x
x
x
x
x
x
x
x
x
x
x
t4
t8
t5
D15 D14 D13 D12 D11 D10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0
Figure 58. Serial Interface Register Read
The AFE5803 SDOUT buffer is tri-stated and will get enabled only when 0[1] (REGISTER READOUT ENABLE)
is enabled. SDOUT pins from multiple AFE5803s can be tied together without any pull-up resistors. Level shifter
SN74AUP1T04 can be used to convert 1.8V logic to 2.5V/3.3V logics if needed.
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t1
AVDD
AVDD_5V
AVDD_ADC
t2
t3
DVDD
t4
t7
t5
RESET
t6
Device Ready for
Serial Register Write
SEN
Device Ready for
Data Conversion
Start of Clock
CLKP_ADC
t8
10 µs < t1 < 50 ms, 10 µs < t2 < 50 ms, –10 ms < t3 < 10 ms, t4 > 10 ms, t5 > 100 ns, t6 > 100 ns, t7 > 10 ms, and
t8 > 100 µs.
The AVDDx and DVDD power-on sequence does not matter as long as –10ms < t3 < 10ms. Similar considerations
apply while shutting down the device.
Figure 59. Recommended Power-up Sequencing and Reset Timing
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REGISTER MAP
ADC Register Map
A reset process is required at the AFE5803 initialization stage. Initialization can be done in one of two ways:
1. Through a hardware reset, by applying a positive pulse at the RESET pin
2. Through a software reset, using the serial interface, by setting the SOFTWARE RESET bit to high. Setting
this bit initializes the internal registers to the respective default values (all zeros) and then self-resets the
SOFTWARE RESET bit to low. In this case, the RESET pin can stay low (inactive).
After reset, all ADC and VCA registers are set to ‘0’, i.e. default settings. During register programming, all
reserved/unlisted register bits need to be set as ‘0’. Register settings are maintained when the AFE5803 is in
either partial power down mode or complete power down mode.
Table 2. ADC Register Map
ADDRESS
(DEC)
ADDRESS
(HEX)
Default
Value
FUNCTION
DESCRIPTION
0[0]
0x0[0]
0x0[1]
0x1[0]
0x1[1]
0x1[9:2]
0
0
0
0
0
SOFTWARE_RESET
0: Normal operation;
1: Resets the device and self-clears the bit to '0'
REGISTER_READOUT_ENABLE 0:Disables readout;
1: enables readout of register at SDOUT Pin
0: Normal
0[1]
1[0]
ADC_COMPLETE_PDN
LVDS_OUTPUT_DISABLE
ADC_PDN_CH<7:0>
1: Complete Power down
1[1]
0: Output Enabled;
1: Output disabled
1[9:2]
0: Normal operation;
1: Power down. Power down Individual ADC channels.
1[9]→CH8…1[2]→CH1
1[10]
1[11]
1[13]
0x1[10]
0x1[11]
0x1[13]
0
0
0
PARTIAL_PDN
0: Normal Operation;
1: Partial Power Down ADC
LOW_FREQUENCY_
NOISE_SUPPRESSION
0: No suppression;
1: Suppression Enabled
EXT_REF
0: Internal Reference;
1: External Reference. VREF_IN is used. Both 3[15] and 1[13] should be set
as 1 in the external reference mode
1[14]
1[15]
0x1[14]
0x1[15]
0
0
LVDS_OUTPUT_RATE_2X
SINGLE-ENDED_CLK_MODE
0: 1x rate;
1: 2x rate. Combines data from 2 channels on 1 LVDS pair. When ADC clock
rate is low, this feature can be used
0: Differential clock input;
1: Single-ended clock input
2[2:0]
0x2[2:0]
0
0
RESERVED
Set to 0
2[10:3]
0x2[10:3]
POWER-DOWN_LVDS
0: Normal operation;
1: PDN Individual LVDS outputs. 2[10]→CH8…2[3]→CH1
2[11]
0x2[11]
0x2[12]
0x2[15:3]
0
0
0
AVERAGING_ENABLE
LOW_LATENCY
0: No averaging;
1: Average 2 channels to increase SNR
2[12]
0: Default Latency with digital features supported, 11 cycle latency
1: Low Latency with digital features bypassed, 8 cycle latency
2[15:13]
TEST_PATTERN_MODES
000: Normal operation;
001: Sync;
010: De-skew;
011: Custom;
100:All 1's;
101: Toggle;
110: All 0's;
111: Ramp
3[7:0]
3[8]
0x3[7:0]
0x3[8]
0
0
INVERT_CHANNELS
0: No inverting;
1:Invert channel digital output. 3[7]→CH8;3[0]→CH1
CHANNEL_OFFSET_
0: No offset subtraction;
SUBSTRACTION_ENABLE
1: Offset value Subtract Enabled
3[9:11]
3[12]
0x3[9:11]
0x3[12]
0
0
RESERVED
Set to 0
DIGITAL_GAIN_ENABLE
0: No digital gain;
1: Digital gain Enabled
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Table 2. ADC Register Map (continued)
ADDRESS
(DEC)
ADDRESS
(HEX)
Default
Value
FUNCTION
DESCRIPTION
3[14:13]
0x3[14:13]
0
SERIALIZED_DATA_RATE
Serialization factor
00: 14x
01: 16x
10: reserved
11: 12x
when 4[1]=1. In the 16x serialization rate, two 0s are filled at two LSBs (see
Table 1)
3[15]
0x3[15]
0
ENABLE_EXTERNAL_
REFERENCE_MODE
0: Internal reference mode;
1: Set to external reference mode
Note: both 3[15] and 1[13] should be set as 1 when configuring the device in
the external reference mode
4[1]
4[3]
4[4]
0x4[1]
0x4[3]
0x4[4]
0
0
0
ADC_RESOLUTION_SELECT
ADC_OUTPUT_FORMAT
LSB_MSB_FIRST
0: 14bit;
1: 12bit
0: 2's complement;
1: Offset binary
0: LSB first;
1: MSB first
5[13:0]
10[8]
0x5[13:0]
0xA[8]
0
0
CUSTOM_PATTERN
SYNC_PATTERN
Custom pattern data for LVDS output (2[15:13]=011)
0: Test pattern outputs of 8 channels are NOT synchronized.
1: Test pattern outputs of 8 channels are synchronized.
13[9:0]
0xD[9:0]
0
0
0
0
0
0
0
0
0
OFFSET_CH1
Value to be subtracted from channel 1 code
0 dB to 6 dB in 0. 2dB steps
13[15:11]
15[9:0]
0xD[15:11]
0xF[9:0]
DIGITAL_GAIN_CH1
OFFSET_CH2
value to be subtracted from channel 2 code
0dB to 6dB in 0.2 dB steps
15[15:11]
17[9:0]
0xF[15:11]
0x11[9:0]
0x11[15:11]
0x13[9:0]
0x13[15:11]
0x15[0]
DIGITAL_GAIN_CH2
OFFSET_CH3
value to be subtracted from channel 3 code
0 dB to 6 dB in 0.2 dB steps
17[15:11]
19[9:0]
DIGITAL_GAIN_CH3
OFFSET_CH4
value to be subtracted from channel 4 code
0 dB to 6 dB in 0. 2dB steps
19[15:11]
21[0]
DIGITAL_GAIN_CH4
DIGITAL_HPF_FILTER_ENABLE 0: Disable the digital HPF filter;
_ CH1-4
1: Enable for 1-4 channels
21[4:1]
0x15[4:1]
0
DIGITAL_HPF_FILTER_K_CH1-4 Set K for the high-pass filter (k from 2 to 10, i.e. 0010B to 1010B).
This group of four registers controls the characteristics of a digital high-pass
transfer function applied to the output data, following the formula:
y(n) = 2k/(2k + 1) [x(n) – x(n – 1) + y(n – 1)] (see Table 3 and Figure 55)
25[9:0]
0x19[9:0]
0
0
0
0
0
0
0
0
0
OFFSET_CH8
value to be subtracted from channel 8 code
0 dB to 6 dB in 0.2dB steps
25[15:11]
27[9:0]
0x19[15:11]
0x1B[9:0]
0x1B[15:11]
0x1D[9:0]
0x1D[15:11]
0x1F[9:0]
DIGITAL_GAIN_CH8
OFFSET_CH7
value to be subtracted from channel 7 code
0 dB to 6dB in 0.2 dB steps
27[15:11]
29[9:0]
DIGITAL_GAIN_CH7
OFFSET_CH6
value to be subtracted from channel 6 code
0 dB to 6 dB in 0.2 dB steps
29[15:11]
31[9:0]
DIGITAL_GAIN_CH6
OFFSET_CH5
value to be subtracted from channel 5 code
0 dB to 6 dB in 0.2 dB steps
31[15:11]
33[0]
0x1F[15:11]
0x21[0]
DIGITAL_GAIN_CH5
DIGITAL_HPF_FILTER_ENABLE 0: Disable the digital HPF filter;
_ CH5-8
1: Enable for 5-8 channels
33[4:1]
0x21[4:1]
0
DIGITAL_HPF_FILTER_K_CH5-8 Set K for the high-pass filter (k from 2 to 10, 010B to 1010B)
This group of four registers controls the characteristics of a digital high-pass
transfer function applied to the output data, following the formula:
y(n) = 2k/(2k + 1) [x(n) – x(n – 1) + y(n – 1)] (see Table 3 and Figure 55)
66[15]
0x42[15]
0
DITHER
0: Disable dither function.
1: Enable dither function. Improve the ADC linearity with slight noise
degradation.
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ADC Register/Digital Processing Description
The ADC in the AFE5803 has extensive digital processing functionalities which can be used to enhance
ultrasound system performance. The digital processing blocks are arranged as in Figure 60.
Final
Digital
Output
Channel
Average
Default=No
Digital
Gain
Default=0
Digital HPF
Default = No
ADC
Output
12/14b
12/14b
Digital Offset
Default=No
Figure 60. ADC Digital Block Diagram
AVERAGING_ENABLE: Address: 2[11]
When set to 1, two samples, corresponding to two consecutive channels, are averaged (channel 1 with 2, 3 with
4, 5 with 6, and 7 with 8). If both channels receive the same input, the net effect is an improvement in SNR. The
averaging is performed as:
•
•
•
•
Channel 1 + channel 2 comes out on channel 3
Channel 3 + channel 4 comes out on channel 4
Channel 5 + channel 6 comes out on channel 5
Channel 7 + channel 8 comes out on channel 6
ADC_OUTPUT_FORMAT: Address: 4[3]
The ADC output, by default, is in 2’s-complement mode. Programming the ADC_OUTPUT_FORMAT bit to 1
inverts the MSB, and the output becomes straight-offset binary mode.
DIGITAL_GAIN_ENABLE: Address: 3[12]
Setting this bit to 1 applies to each channel i the corresponding gain given by DIGTAL_GAIN_CHi <15:11>. The
gain is given as 0dB + 0.2dB × DIGTAL_GAIN_CHi<15:11>. For instance, if DIGTAL_GAIN_CH5<15:11> = 3,
channel 5 is increased by 0.6dB gain. DIGTAL_GAIN_CHi <15:11> = 31 produces the same effect as
DIGTAL_GAIN_CHi <15:11> = 30, setting the gain of channel i to 6dB.
DIGITAL_HPF_ENABLE
•
•
CH1-4: Address 21[0]
CH5-8: Address 33[0]
DIGITAL_HPF_FILTER_K_CHX
•
•
CH1-4: Address 21[4:1]
CH5-8: Address 3[4:1]
This group of registers controls the characteristics of a digital high-pass transfer function applied to the output
data, following Equation 1.
2k
2k +1
é
ù
x n - x n -1 + y n -1
y n =
( )
( ) ( ) (
)
ë
û
(1)
These digital HPF registers (one for the first four channels and one for the second group of four channels)
describe the setting of K. The digital high pass filter can be used to suppress low frequency noise which
commonly exists in ultrasound echo signals. The digital filter can significantly benefit near field recovery time due
to T/R switch low frequency response. Table 3 shows the cut-off frequency vs K, also see Figure 55.
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Table 3. Digital HPF –1dB Corner Frequency vs K and Fs
k
2
3
4
40 MSPS
2780 KHz
1490 KHz
770 KHz
50 MSPS
3480 KHz
1860 KHz
960 KHz
65 MSPS
4520 KHz
2420 KHz
1250 KHz
LOW_FREQUENCY_NOISE_SUPPRESSION: Address: 1[11]
The low-frequency noise suppression mode is especially useful in applications where good noise performance is
desired in the frequency band of 0MHz to 1MHz (around dc). Setting this mode shifts the low-frequency noise of
the AFE5803 to approximately Fs/2, thereby moving the noise floor around dc to a much lower value. Register bit
1[11] is used for enabling or disabling this feature. When this feature is enabled, power consumption of the
device will be increased by approximate 1 mW/CH.
LVDS_OUTPUT_RATE_2X: Address: 1[14]
The output data always uses a DDR format, with valid/different bits on the positive as well as the negative edges
of the LVDS bit clock, DCLK. The output rate is set by default to 1X (LVDS_OUTPUT_RATE_2X = 0), where
each ADC has one LVDS stream associated with it. If the sampling rate is low enough, two ADCs can share one
LVDS stream, in this way lowering the power consumption devoted to the interface. The unused outputs will
output zero. To avoid consumption from those outputs, no termination should be connected to them. The
distribution on the used output pairs is done in the following way:
•
•
•
•
Channel 1 and channel 2 come out on channel 3. Channel 1 comes out first.
Channel 3 and channel 4 come out on channel 4. Channel 3 comes out first.
Channel 5 and channel 6 come out on channel 5. Channel 5 comes out first.
Channel 7 and channel 8 come out on channel 6. Channel 7 comes out first
CHANNEL_OFFSET_SUBSTRACTION_ENABLE: Address: 3[8]
Setting this bit to 1 enables the subtraction of the value on the corresponding OFFSET_CHx<9:0> (offset for
channel i) from the ADC output. The number is specified in 2s-complement format. For example,
OFFSET_CHx<9:0> = 11 1000 0000 means subtract –128. For OFFSET_CHx<9:0> = 00 0111 1111 the effect is
to subtract 127. In effect, both addition and subtraction can be performed. Note that the offset is applied before
the digital gain (see DIGITAL_GAIN_ENABLE). The whole data path is 2s-complement throughout internally, with
digital gain being the last step. Only when ADC_OUTPUT_FORMAT=1 (straight binary output format) is the
2s-complement word translated into offset binary at the end.
SERIALIZED_DATA_RATE: Address: 3[14:13]
See Table 1 for detail description.
TEST_PATTERN_MODES: Address: 2[15:13]
The AFE5803 can output a variety of test patterns on the LVDS outputs. These test patterns replace the normal
ADC data output. The device may also be made to output 6 preset patterns:
1. Ramp: Setting Register 2[15:13]=111causes all the channels to output a repeating full-scale ramp pattern.
The ramp increments from zero code to full-scale code in steps of 1LSB every clock cycle. After hitting the
full-scale code, it returns back to zero code and ramps again.
2. Zeros: The device can be programmed to output all zeros by setting Register 2[15:13]=110;
3. Ones: The device can be programmed to output all 1s by setting Register 2[15:13]=100;
4. Deskew Patten: When 2[15:13]=010; this mode replaces the 14-bit ADC output with the 01010101010101
word.
5. Sync Pattern: When 2[15:13]=001, the normal ADC output is replaced by a fixed 11111110000000 word.
6. Toggle: When 2[15:13]=101, the normal ADC output is alternating between 1's and 0's. The start state of
ADC word can be either 1's or 0's.
7. Custom Pattern: It can be enabled when 2[15:13]= 011;. Users can write the required VALUE into register
bits <CUSTOM PATTERN> which is Register 5[13:0]. Then the device will output VALUE at its outputs,
about 3 to 4 ADC clock cycles after the 24th rising edge of SCLK. So, the time taken to write one value is 24
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SCLK clock cycles + 4 ADC clock cycles. To change the customer pattern value, users can repeat writing
Register 5[13:0] with a new value. Due to the speed limit of SPI, the refresh rate of the custom pattern may
not be high. For example, 128 points custom pattern will take approximately 128 x (24 SCLK clock cycles + 4
ADC clock cycles).
NOTE
only one of the above patterns can be active at any given instant.
SYNC_PATTERN: Address: 10[8]
By enabling this bit, all channels' test pattern outputs are synchronized. When 10[8] is set as 1, the ramp
patterns of all 8 channels start simultaneously.
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VCA Register Map
Table 4. VCA Register Map
ADDRESS
(DEC)
ADDRESS
(HEX)
Default
Value
FUNCTION
DESCRIPTION
51[0]
0x33[0]
0
0
RESERVED
0
51[3:1]
0x33[3:1]
LPF_PROGRAMMABILITY
000: 15MHz,
010: 20MHz,
011: 30MHz,
100: 10MHz
51[4]
0x33[4]
0
0
PGA_INTEGRATOR_DISABLE
(PGA_HPF_DISABLE)
0: Enable
1: Disables offset integrator for PGA. See explanation for the PGA
integrator function in APPLICATION INFORMATION section
51[6:5]
0x33[6:5]
PGA_CURRENT_CLAMP_LEVEL
00: –2dBFS;
10: 0dBFS;
01:–4dBFS when 51[7]=0
Note: the current clamp circuit makes sure that PGA output is in linear
range. For example, at 00 setting, PGA output HD3 will be worsen by
3dB at –2dBFS ADC input. In normal operation, the current clamp
function can be set as 00
51[7]
0x33[7]
0
0
PGA_CURRENT_CLAMP_DISABLE
PGA_GAIN_CONTROL
0:Enables the PGA current clamp circuit;
1:Disables the PGA current clamp circuit before the PGA outputs.
51[6:5] determines the current clamp level
51[13]
0x33[13]
0:24dB;
1:30dB.
52[4:0]
52[5]
0x34[4:0]
0x34[5]
0
0
ACTIVE_TERMINATION_
INDIVIDUAL_RESISTOR_CNTL
See Table 6 Reg 52[5] should be set as '1' to access these bits
ACTIVE_TERMINATION_
0: Disables;
INDIVIDUAL_RESISTOR_ENABLE
1: Enables internal active termination individual resistor control
52[7:6]
0x34[7:6]
0
PRESET_ACTIVE_ TERMINATIONS
00: 50ohm,
01: 100ohm,
10: 200ohm,
11: 400ohm.
(Note: the device will adjust resistor mapping (52[4:0]) automatically.
50ohm active termination is NOT supported in 12dB LNA setting.
Instead, '00' represents high impedance mode when LNA gain is 12dB)
52[8]
0x34[8]
0
0
ACTIVE TERMINATION ENABLE
LNA_INPUT_CLAMP_SETTING
0: Disables;
1: Enables active termination
52[10:9]
0x34[10:9]
00: Auto setting,
01: 1.5Vpp,
10: 1.15Vpp and
11: 0.6Vpp
52[11]
52[12]
0x34[11]
0x34[12]
0
0
RESERVED
Set to 0
LNA_INTEGRATOR_DISABLE
(LNA_HPF_DISABLE)
0: Enables;
1: Disables offset integrator for LNA. See the explanation for this
function in the following section
52[14:13]
0x34[14:13]
0
LNA_GAIN
00: 18dB;
01: 24dB;
10: 12dB;
11: Reserved
52[15]
0x34[15]
0x35[7:0]
0
0
LNA_INDIVIDUAL_CH_CNTL
0: Disable;
1: Enable LNA individual channel control. See Register 57 for details
53[7:0]
PDN_CH<7:0>
0: Normal operation;
1: Powers down corresponding channels. Bit7→CH8,
Bit6→CH7…Bit0→CH1. PDN_CH will shut down whichever blocks are
active
53[8]
53[9]
53[10]
0x35[8]
0x35[9]
0x35[10]
0
0
0
RESERVED
RESERVED
LOW_POWER
Set to 0
Set to 0
0: Low noise mode;
1: Sets to low power mode (53[11]=0). At 30dB PGA, total chain gain
may slightly change.
See typical characteristics
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Table 4. VCA Register Map (continued)
ADDRESS
(DEC)
ADDRESS
(HEX)
Default
Value
FUNCTION
DESCRIPTION
53[11]
0x35[11]
0
MED_POWER
0: Low noise mode;
1: Sets to medium power mode(53[10]=0). At 30dB PGA, total chain
gain may slightly change.
See typical characteristics
53[12]
53[13]
53[14]
53[15]
0x35[12]
0x35[13]
0x35[14]
0x35[15]
0
0
0
0
PDN_VCAT_PGA
PDN_LNA
0: Normal operation;
1: Powers down VCAT (voltage-controlled-attenuator) and PGA
0: Normal operation;
1: Powers down LNA only
VCA_PARTIAL_PDN
VCA_COMPLETE_PDN
0: Normal operation;
1: Powers down LNA, VCAT, and PGA partially(fast wake response)
0: Normal operation;
1: Powers down LNA, VCAT, and PGA completely (slow wake
response). This bit can overwrite 53[14].
54[4:0]
54[7]
0x36[4:0]
0x36[7]
0
0
0
0
0
0
BUFFER_AMP_GAIN_CNTL
RESERVED
Selects Feedback resistor for the buffer amplifier see Table 7
Set to 0
Set to 0
Set to 0
54[8]
0x36[8]
RESERVED
54[9]
0x36[9]
RESERVED
57[1:0]
57[3:2]
0x39[1:0]
0x39[3:2]
CH1_LNA_GAIN_CNTL
00: 18dB;
01: 24dB;
10: 12dB;
11: Reserved
REG52[15] should be set as '1'
CH2_LNA_GAIN_CNTL
57[5:4]
0x39[5:4]
0
0
0
0
0
0
0
CH3_LNA_GAIN_CNTL
CH4_LNA_GAIN_CNTL
CH5_LNA_GAIN_CNTL
CH6_LNA_GAIN_CNTL
CH7_LNA_GAIN_CNTL
CH8_LNA_GAIN_CNTL
HPF_LNA
00: 18dB;
01: 24dB;
10: 12dB;
11: Reserved
REG52[15] should be set as '1'
57[7:6]
0x39[7:6]
57[9:8]
0x39[9:8]
57[11:10]
57[13:12]
57[15:14]
59[3:2]
0x39[11:10]
0x39[13:12]
0x39[15:14]
0x3B[3:2]
00: 100kHz;
01: 50kHz;
10: 200kHz;
11: 150kHz with 0.015uF on INMx
59[6:4]
0x3B[6:4]
0
DIG_TGC_ATT_GAIN
000: 0dB attenuation;
001: 6dB attenuation;
N: ~N×6dB attenuation when 59[7] = 1
59[7]
59[8]
59[9]
0x3B[7]
0x3B[8]
0x3B[9]
0
0
0
DIG_TGC_ATT
0: disable digital TGC attenuator;
1: enable digital TGC attenuator
BUFFER_AMP_PDN
PGA_TEST_MODE
0: Power down;
1: Normal operation
0: Normal operation;
1: PGA outputs appear at test outputs
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AFE5803 VCA Register Description
LNA Input Impedances Configuration (Active Termination Programmability)
Different LNA input impedances can be configured through the register 52[4:0]. By enabling and disabling the
feedback resistors between LNA outputs and ACTx pins, LNA input impedance is adjustable accordingly. Table 5
describes the relationship between LNA gain and 52[4:0] settings.
The AFE5803 also has 4 preset active termination impedances as described in 52[7:6]. An internal decoder is
used to select appropriate resistors corresponding to different LNA gain.
Table 5. Register 52[4:0] Description
52[4:0]/0x34[4:0]
00000
FUNCTION
No feedback resistor enabled
Enables 450 Ω feedback resistor
Enables 900 Ω feedback resistor
Enables 1800 Ω feedback resistor
Enables 3600 Ω feedback resistor
Enables 4500 Ω feedback resistor
00001
00010
00100
01000
10000
Table 6. Register 52[4:0] vs LNA Input Impedances
52[4:0]/0x34[4:0]
LNA:12dB
00000
High Z
High Z
High Z
00001
150 Ω
90 Ω
00010
300 Ω
180 Ω
100 Ω
00011
100 Ω
60 Ω
00100
600 Ω
360 Ω
200 Ω
00101
120 Ω
72 Ω
00110
200 Ω
120 Ω
66.67 Ω
00111
86 Ω
51 Ω
29 Ω
LNA:18dB
LNA:24dB
50 Ω
33 Ω
40 Ω
52[4:0]/0x34[4:0]
LNA:12dB
01000
1200 Ω
720 Ω
400 Ω
01001
133 Ω
80 Ω
01010
240 Ω
144 Ω
80 Ω
01011
92 Ω
55 Ω
31 Ω
01100
400 Ω
240 Ω
133 Ω
01101
109 Ω
65 Ω
01110
171 Ω
103 Ω
57 Ω
01111
80 Ω
48 Ω
27 Ω
LNA:18dB
LNA:24dB
44 Ω
36 Ω
52[4:0]/0x34[4:0]
LNA:12dB
10000
1500 Ω
900 Ω
500 Ω
10001
136 Ω
82 Ω
10010
250 Ω
150 Ω
83 Ω
10011
94 Ω
56 Ω
31 Ω
10100
429 Ω
257 Ω
143 Ω
10101
111 Ω
67 Ω
10110
176 Ω
106 Ω
59 Ω
10111
81 Ω
49 Ω
27 Ω
LNA:18dB
LNA:24dB
45 Ω
37 Ω
52[4:0]/0x34[4:0]
LNA:12dB
11000
667 Ω
400 Ω
222 Ω
11001
122 Ω
73 Ω
11010
207 Ω
124 Ω
69 Ω
11011
87 Ω
52 Ω
29 Ω
11100
316 Ω
189 Ω
105 Ω
11101
102 Ω
61 Ω
11110
154 Ω
92 Ω
11111
76 Ω
46 Ω
25 Ω
LNA:18dB
LNA:24dB
41 Ω
34 Ω
51 Ω
Programmable Feedback Resistors for Buffer Amplifier in TGC Test Mode
Different feedback resistors can be configured for the buffer amplifier through the register 54[4:0] when TGC test
mode is enabled. Therefore, certain gain can be applied to the PGA test outputs for CH7 and CH8. Table 7
describes the relationship between the resistor configurations and 54[4:0] settings. Note these 5 resistor are put
in parallel. When multiple bits are enabled, the resistance is reduced.
Table 7. Register 54[4:0] Description
54[4:0]
00000
00001
00010
00100
01000
10000
FUNCTION
No feedback resistor enabled
Enables 250 Ω feedback resistor
Enables 250 Ω feedback resistor
Enables 500 Ω feedback resistor
Enables 1000 Ω feedback resistor
Enables 2000 Ω feedback resistor
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THEORY OF OPERATION
AFE5803 OVERVIEW
The AFE5803 is a highly integrated Analog Front-End (AFE) solution specifically designed for ultrasound
systems in which high performance and small size are required. The AFE5803 integrates a complete
time-gain-control (TGC) imaging path. It also enables users to select one of various power/noise combinations to
optimize system performance. The AFE5803 contains eight channels; each channels includes a Low-Noise
Amplifier (LNA), a Voltage Controlled Attenuator (VCAT), a Programmable Gain Amplifier (PGA), a Low-pass
Filter (LPF), and a 14-bit Analog-to-Digital Converter (ADC).
In addition, multiple features in the AFE5803 are suitable for ultrasound applications, such as active termination,
individual channel control, fast power up/down response, programmable clamp voltage control, fast and
consistent overload recovery, etc. Therefore the AFE5803 brings premium image quality to ultra–portable,
handheld systems all the way up to high-end ultrasound systems. Its simplified function block diagram is listed in
Figure 61.
AFE5803 (1 of 8 Channels)
SPI OUT
SPI IN
SPI Logic
3rd LP Filter
10, 15, 20,
30 MHz
PGA
24, 30dB
VCAT
0 to -40dB
14Bit
ADC
LVDS
LNA
LNA IN
Reference
Reference
Differential
TGC Vcntl
EXT/INT
REFs
Figure 61. Functional Block Diagram
LOW-NOISE AMPLIFIER (LNA)
In many high-gain systems, a low noise amplifier is critical to achieve overall performance. Using a new
proprietary architecture, the LNA in the AFE5803 delivers exceptional low-noise performance, while operating on
a low quiescent current compared to CMOS-based architectures with similar noise performance. The LNA
performs single-ended input to differential output voltage conversion. It is configurable for a programmable gain
of 24/18/12dB and its input-referred noise is only 0.63/0.70/0.9nV/√Hz respectively. Programmable gain settings
result in a flexible linear input range up to 1Vpp, realizing high signal handling capability demanded by new
transducer technologies. Larger input signal can be accepted by the LNA; however, the signal can be distorted
since it exceeds the LNA’s linear operation region. Combining the low noise and high input range, a wide input
dynamic range is achieved consequently for supporting the high demands from various ultrasound imaging
modes.
The LNA input is internally biased at approximately +2.4 V; the signal source should be ac-coupled to the LNA
input by an adequately-sized capacitor, e.g. ≥0.1 µF. To achieve low DC offset drift, the AFE5803 incorporates a
DC offset correction circuit for each amplifier stage. To improve the overload recovery, an integrator circuit is
used to extract the DC component of the LNA output and then fed back to the LNA’s complementary input for DC
offset correction. This DC offset correction circuit has a high-pass response and can be treated as a high-pass
filter. The effective corner frequency is determined by the capacitor CBYPASS connected at INM. With larger
capacitors, the corner frequency is lower. For stable operation at the highest HP filer cut-off frequency, a ≥15 nF
capacitor can be selected. This corner frequency scales almost linearly with the value of the CBYPASS. For
example, 15 nF gives a corner frequency of approximately 100 kHz, while 47 nF can give an effective corner
frequency of 33 KHz. The DC offset correction circuit can also be disabled/enabled through register 52[12].
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The AFE5803 can be terminated passively or actively. Active termination is preferred in ultrasound application for
reducing reflection from mismatches and achieving better axial resolution without degrading noise figure too
much. Active termination values can be preset to 50, 100, 200, 400Ω; other values also can be programmed by
users through register 52[4:0]. A feedback capacitor is required between ACTx and the signal source as
Figure 62 shows. On the active termination path, a clamping circuit is also used to create a low impedance path
when overload signal is seen by the AFE5803. The clamp circuit limits large input signals at the LNA inputs and
improves the overload recovery performance of the AFE5803. The clamp level can be set to 350 mVpp, 600
mVpp, 1.15 Vpp automatically depending on the LNA gain settings when register 52[10:9]=0. Other clamp
voltages, such as 1.15Vpp, 0.6 Vpp, and 1.5 Vpp, are also achievable by setting register 52[10:9]. This clamping
circuit is also designed to obtain good pulse inversion performance and reduce the impact from asymmetric
inputs.
CLAMP
AFE
CACT
ACTx
CIN
INPx
INPUT
CBYPSS
LNAx
INMx
DC Offset
Correction
Figure 62. AFE5803 LNA with DC Offset Correction Circuit
VOLTAGE-CONTROLLED ATTENUATOR
The voltage-controlled attenuator is designed to have a linear-in-dB attenuation characteristic; that is, the
average gain loss in dB (refer to Figure 2) is constant for each equal increment of the control voltage (VCNTL) as
shown in Figure 63. A differential control structure is used to reduce common mode noise. A simplified attenuator
structure is shown in the following Figure 63 and Figure 64.
The attenuator is essentially a variable voltage divider that consists of the series input resistor (RS) and seven
shunt FETs placed in parallel and controlled by sequentially activated clipping amplifiers (A1 through A7). VCNTL
is the effective difference between VCNTLP and VCNTLM. Each clipping amplifier can be understood as a
specialized voltage comparator with a soft transfer characteristic and well-controlled output limit voltage.
Reference voltages V1 through V7 are equally spaced over the 0V to 1.5Vcontrol voltage range. As the control
voltage increases through the input range of each clipping amplifier, the amplifier output rises from a voltage
where the FET is nearly OFF to VHIGH where the FET is completely ON. As each FET approaches its ON state
and the control voltage continues to rise, the next clipping amplifier/FET combination takes over for the next
portion of the piecewise-linear attenuation characteristic. Thus, low control voltages have most of the FETs
turned OFF, producing minimum signal attenuation. Similarly, high control voltages turn the FETs ON, leading to
maximum signal attenuation. Therefore, each FET acts to decrease the shunt resistance of the voltage divider
formed by Rs and the parallel FET network.
Additionally, a digitally controlled TGC mode is implemented to achieve better phase-noise performance in the
AFE5803. The attenuator can be controlled digitally instead of the analog control voltage VCNTL. This mode can
be set by the register bit 59[7]. The variable voltage divider is implemented as a fixed series resistance and FET
as the shunt resistance. Each FET can be turned ON by connecting the switches SW1-7. Turning on each of the
switches can give approximately 6dB of attenuation. This can be controlled by the register bits 59[6:4]. This
digital control feature can eliminate the noise from the VCNTL circuit and ensure the better SNR and phase noise
for TGC path.
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A1 - A7 Attenuator Stages
RS
Attenuator
Input
Attenuator
Output
Q1
A1
Q2
Q3
Q4
Q5
Q6
Q7
VB
A1
A1
A1
A1
A1
A1
C1
C2
C3
C4
C5
C6
C7
V1
V2
V3
V4
V5
V6
V7
VCNTL
C1 - C8 Clipping Amplifiers
Control
Input
Figure 63. Simplified Voltage Controlled Attenuator (Analog Structure)
RS
Attenuator
Input
Attenuator
Output
Q1
Q2
Q3
Q4
Q5
Q6
Q7
VB
SW4
SW1
SW2
SW3
SW5
SW6
SW7
VHIGH
Figure 64. Simplified Voltage Controlled Attenuator (Digital Structure)
The voltage controlled attenuator’s noise follows a monotonic relationship to the attenuation coefficient. At higher
attenuation, the input-referred noise is higher and vice-versa. The attenuator’s noise is then amplified by the PGA
and becomes the noise floor at ADC input. In the attenuator’s high attenuation operating range, i.e. VCNTL is
high, the attenuator’s input noise may exceed the LNA’s output noise; the attenuator then becomes the dominant
noise source for the following PGA stage and ADC. Therefore the attenuator’s noise should be minimized
compared to the LNA output noise. The AFE5803’s attenuator is designed for achieving low noise even at high
attenuation (low channel gain) and realizing better SNR in near field. The input referred noise for different
attenuations is listed in the below table:
Table 8. Voltage-Controlled-Attenuator noise vs Attenuation
Attenuation (dB)
Attenuator Input Referred noise (nV/rtHz)
–40
–36
–30
–24
–18
–12
–6
10.5
10
9
8.5
6
4
3
0
2
PROGRAMMABLE GAIN AMPLIFIER (PGA)
After the voltage controlled attenuator, a programmable gain amplifier can be configured as 24dB or 30dB with a
constant input referred noise of 1.75 nV/rtHz. The PGA structure consists of a differential voltage-to-current
converter with programmable gain, current clamp( bias control) circuits, a transimpedance amplifier with a
programmable low-pass filter, and a DC offset correction circuit. Its simplified block diagram is shown below:
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Current Clamp
Current Clamp
To ADC
From attenuator
I/V
LPF
V/I
DC Offset
Correction Loop
Figure 65. Simplified Block Diagram of PGA
Low input noise is always preferred in a PGA and its noise contribution should not degrade the ADC SNR too
much after the attenuator. At the minimum attenuation (used for small input signals), the LNA noise dominates; at
the maximum attenuation (large input signals), the PGA and ADC noise dominates. Thus 24 dB gain of PGA
achieves better SNR as long as the amplified signals can exceed the noise floor of the ADC.
The PGA current clamp circuit can be enabled (register 51) to improve the overload recovery performance of the
AFE. If we measure the standard deviation of the output just after overload, for 0.5 V VCNTL, it is about 3.2 LSBs
in normal case, i.e the output is stable in about 1 clock cycle after overload. With the current clamp circuit
disabled, the value approaches 4 LSBs meaning a longer time duration before the output stabilizes; however,
with the current clamp circuit enabled, there will be degradation in HD3 for PGA output levels > -2dBFS. For
example, for a –2dBFS output level, the HD3 degrades by approximately 3dB.
The AFE5803 integrates an anti-aliasing filter in the form of a programmable low-pass filter (LPF) in the
transimpedance amplifier. The LPF is designed as a differential, active, 3rd order filter with a typical 18dB per
octave roll-off. Programmable through the serial interface, the –1dB frequency corner can be set to one of
10MHz, 15 MHz, 20 MHz, and 30MHz. The filter bandwidth is set for all channels simultaneously.
A selectable DC offset correction circuit is implemented in the PGA as well. This correction circuit is similar to the
one used in the LNA. It extracts the DC component of the PGA outputs and feeds back to the PGA’s
complimentary inputs for DC offset correction. This DC offset correction circuit also has a high-pass response
with a cut-off frequency of 80 KHz.
ANALOG TO DIGITAL CONVERTER
The analog-to-digital converter (ADC) of the AFE5803 employs a pipelined converter architecture that consists of
a combination of multi-bit and single-bit internal stages. Each stage feeds its data into the digital error correction
logic, ensuring excellent differential linearity and no missing codes at the 14-bit level. The 14 bits given out by
each channel are serialized and sent out on a single pair of pins in LVDS format. All eight channels of the
AFE5803 operate from a common input clock (CLKP/M). The sampling clocks for each of the eight channels are
generated from the input clock using a carefully matched clock buffer tree. The 14x clock required for the
serializer is generated internally from the CLKP/M pins. A 7x and a 1x clock are also given out in LVDS format,
along with the data, to enable easy data capture. The AFE5803 operates from internally-generated reference
voltages that are trimmed to improve the gain matching across devices. The nominal values of REFP and REFM
are 1.5V and 0.5V, respectively. Alternately, the device also supports an external reference mode that can be
enabled using the serial interface.
Using serialized LVDS transmission has multiple advantages, such as a reduced number of output pins (saving
routing space on the board), reduced power consumption, and reduced effects of digital noise coupling to the
analog circuit inside the AFE5803.
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EQUIVALENT CIRCUITS
CM
CM
(a) INP
(b) INM
(c) ACT
S0492-01
Figure 66. Equivalent Circuits of LNA inputs
S0493-01
Figure 67. Equivalent Circuits of VCNTLP/M
VCM
5 kΩ
5 kΩ
CLKP
CLKM
ADC Input Clocks
Figure 68. Equivalent Circuits of Clock Inputs
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AFE5803
+
–
+Vdiff
OUTP
External
100-W Load
+
–
OUTM
Rout
+
–
Vcommon
–Vdiff
Switch impedance is
nominally 50 W (±10%)
Figure 69. Equivalent Circuits of LVDS Outputs
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APPLICATION INFORMATION
5VA
10μF
3.3VA
1.8VA
1.8VD
10μF
10μF
10μF
N*0.1μF
AVSS
N*0.1μF
AVSS
N*0.1μF
DVSS
0.1μF
AVSS
Clock termination
depends on clock types
LVDS, PECL, or CMOS
1μF ACT1
D1P
0.1 μF IN1P
15nF IN1M
1μF ACT2
0.1μF IN2P
15nF IN2M
1μF ACT3
0.1μF IN3P
15nF IN3M
1μF ACT4
0.1μF IN4P
15nF IN4M
1μF ACT5
0.1μF IN5P
15nF IN5M
1μF ACT6
0.1μF IN6P
15nF IN6M
1μF ACT7
0.1μF IN7P
15nF IN7M
1μF ACT8
0.1μF IN8P
D1M
D2P
IN CH1
IN CH2
IN CH3
IN CH4
IN CH5
IN CH6
IN CH7
IN CH8
0.1μF
CLKP
D2M
D3P
0.1μF
AFE5803
CLKM
CLOCK
INPUTS
D3M
D4P
D4M
D5P
D5M
D6P
SOUT
SDATA
SCLK
SEN
D6M
D7P
AFE5803
D7M
D8P
RESET
DIGITAL
INPUTS
AFE5803
PDN_VCA
D8M
DCLKP
DCLKM
FCLKP
FCLKM
PDN_ADC
ANALOG INPUTS
ANALOG OUTPUTS
REF/BIAS DECOUPLING
LVDS OUTPUTS
PDN_GLOBAL
CH7_TEST_OUTP
CH7_BUFFER_OUTM
CH7_TEST_OUTM
CH7_BUFFER_OUTP
15nF IN8M
CM_BYP
R (optional)
CH7_OUTM
>1μF
>1μF
VHIGH
R (optional)
CH7_OUTP
RVCNTL
CVCNTL
470pF
200Ω
VCNTLP IN
VCNTLM IN
VCNTLP
CH8_TEST_OUTP
CH8_BUFFER_OUTM
CH8_TEST_OUTM
CH8_BUFFER_OUTP
VCNTLM
CVCNTL
RVCNTL
200Ω
470pF
VREF_IN
R (optional)
R (optional)
CH8_OUTM
CH8_OUTP
1.4V
0.1μF
REFM
REFP
DNCs
DVSS
AVSS
Figure 70. Application Circuit
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A typical application circuit diagram is listed above. The configuration for each block is discussed below.
LNA CONFIGURATION
LNA Input Coupling and Decoupling
The LNA closed-loop architecture is internally compensated for maximum stability without the need of external
compensation components. The LNA inputs are biased at 2.4 V and AC coupling is required. A typical input
configuration is shown in Figure 71. CIN is the input AC coupling capacitor. CACT is a part of the active termination
feedback path. Even if the active termination is not used, the CACT is required for the clamp functionality.
Recommended values for CACT = 1 µF and CIN are ≥ 0.1 µF. A pair of clamping diodes is commonly placed
between the T/R switch and the LNA input. Schottky diodes with suitable forward drop voltage (e.g. the
BAT754/54 series, the BAS40 series, the MMBD7000 series, or similar) can be considered depending on the
transducer echo amplitude.
AFE
CLAMP
CACT
ACTx
CIN
INPx
Input
CBYPASS
LNAx
INMx
DC Offset
Correction
S0498-01
Figure 71. LNA Input Configurations
This architecture minimizes any loading of the signal source that may otherwise lead to a frequency-dependent
voltage divider. The closed-loop design yields low offsets and offset drift. CBYPASS (≥0.015 µF) is used to set the
high-pass filter cut-off frequency and decouple the complimentary input. Its cut-off frequency is inversely
proportional to the CBYPASS value, The HPF cut-off frequency can be adjusted through the register 59[3:2] a
Table 9 lists. Low frequency signals at T/R switch output, such as signals with slow ringing, can be filtered out. In
addition, the HPF can minimize system noise from DC-DC converters, pulse repetition frequency (PRF) trigger,
and frame clock. Most ultrasound systems’ signal processing unit includes digital high-pass filters or band-pass
filters (BPFs) in FPGAs or ASICs. Further noise suppression can be achieved in these blocks. In addition, a
digital HPF is available in the AFE5803 ADC. If low frequency signal detection is desired in some applications,
the LNA HPF can be disabled.
Table 9. LNA HPF Settings (CBYPASS = 15 nF)
Reg59[3:2] (0x3B[3:2])
Frequency
100 KHz
50 KHz
00
01
10
11
200 KHz
150 KHz
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CM_BYP and VHIGH pins, which generate internal reference voltages, need to be decoupled with ≥1uF
capacitors. Bigger bypassing capacitors (>2.2uF) may be beneficial if low frequency noise exists in system.
LNA Noise Contribution
The noise spec is critical for LNA and it determines the dynamic range of entire system. The LNA of the
AFE5803 achieves low power and an exceptionally low-noise voltage of 0.63 nV/√Hz, and a low current noise of
2.7 pA/√Hz.
Typical ultrasonic transducer’s impedance Rs varies from tens of ohms to several hundreds of ohms. Voltage
noise is the dominant noise in most cases; however, the LNA current noise flowing through the source
impedance (Rs) generates additional voltage noise.
LNA _Noisetotal = VL2NAnoise + Rs2 ´IL2NAnoise
(2)
The AFE5803 achieves low noise figure (NF) over a wide range of source resistances as shown in Figure 29,
Figure 30, andFigure 31.
Active Termination
In ultrasound applications, signal reflection exists due to long cables between transducer and system. The
reflection results in extra ringing added to echo signals in PW mode. Since the axial resolution depends on echo
signal length, such ringing effect can degrade the axial resolution. Hence, either passive termination or active
termination, is preferred if good axial resolution is desired. Figure 72 shows three termination configurations:
Rs
Rs
Rs
LNA
(a) No Termination
Rf
LNA
(b) Active Termination
LNA
Rt
(c) Passive Termination
S0499-01
Figure 72. Termination Configurations
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Under the no termination configuration, the input impedance of the AFE5803 is about 6 KΩ (8 K//20 pF) at 1
MHz. Passive termination requires external termination resistor Rt, which contributes to additional thermal noise.
The LNA supports active termination with programmable values, as shown in Figure 73 .
450Ω
900Ω
1800Ω
ACTx
3600Ω
4500Ω
INPx
Input
LNAx
INMx
AFE
S0500-01
Figure 73. Active Termination Implementation
The AFE5803 has four pre-settings 50,100, 200 and 400 Ω which are configurable through the registers. Other
termination values can be realized by setting the termination switches shown in Figure 73. Register [52] is used
to enable these switches. The input impedance of the LNA under the active termination configuration
approximately follows:
Rf
ZIN =
AnLNA
1+
2
(3)
Table 5 lists the LNA RINs under different LNA gains. System designers can achieve fine tuning for different
probes.
The equivalent input impedance is given by Equation 4 where RIN (8 K) and CIN (20 pF) are the input resistance
and capacitance of the LNA.
Rf
ZIN =
/ /CIN / /RIN
AnLNA
1+
2
(4)
Therefore the ZIN is frequency dependent and it decreases as frequency increases shown in Figure 10. Since 2
MHz~10 MHz is the most commonly used frequency range in medical ultrasound, this rolling-off effect doesn’t
impact system performance greatly. Since each ultrasound system includes multiple transducers with different
impedances, the flexibility of impedance configuration is a great plus.
Figure 29, Figure 30, and Figure 31 shows the NF under different termination configurations. It indicates that no
termination achieves the best noise figure; active termination adds less noise than passive termination. Thus
termination topology should be carefully selected based on each use scenario in ultrasound.
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LNA Gain Switch Response
The LNA gain is programmable through SPI. The gain switching time depends on the SPI speed as well as the
LNA gain response time. During the switching, glitches might occur and they can appear as artifacts in images.
LNA gain switching in a single imaging line may not be preferred, although digital signal processing might be
used here for glitch suppression.
VOLTAGE-CONTROLLED-ATTENUATOR
The attenuator in the AFE5803 is controlled by a pair of differential control inputs, the VCNTLM/P pins. The
differential control voltage spans from 0 V to 1.5 V. This control voltage varies the attenuation of the attenuator
based on its linear-in-dB characteristic. Its maximum attenuation (minimum channel gain) appears at VCNTLP
-
VCNTLM = 1.5 V, and minimum attenuation (maximum channel gain) occurs at VCNTLP - VCNTLM = 0. The typical
gain range is 40 dB and remains constant, independent of the PGA setting.
When only single-ended VCNTL signal is available, this 1.5 VPP signal can be applied on the VCNTLP pin with the
VCNTLM pin connected to ground. As shown in Figure 74, TGC gain curve is inversely proportional to the VCNTLP
-
VCNTLM
.
1.5V
VCNTLP
VCNTLM = 0V
X+40dB
TGC Gain
XdB
(a) Single-Ended Input at VCNTLP
1.5V
VCNTLP
0.75V
VCNTLM
0V
X+40dB
TGC Gain
XdB
(b) Differential Inputs at VCNTLP and VCNTLM
W0004-01
Figure 74. VCNTLP and VCNTLM Configurations
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As discussed in the theory of operation, the attenuator architecture uses seven attenuator segments that are
equally spaced in order to approximate the linear-in-dB gain-control slope. This approximation results in a
monotonic slope; the gain ripple is typically less than ±0.5 dB.
The control voltage input (VCNTLM/P pins) represents a high-impedance input. The VCNTLM/P pins of multiple
AFE5803 devices can be connected in parallel with no significant loading effects. When the voltage level (VCNTLP
- VCNTLM) is above 1.5 V or below 0 V, the attenuator continues to operate at its maximum attenuation level or
minimum attenuation level respectively. It is recommended to limit the voltage from -0.3 V to 2 V.
The AFE5803 gain-control input has a –3 dB bandwidth of approximately 800KHz. This wide bandwidth, although
useful in many applications (e.g. fast VCNTL response), can also allow high-frequency noise to modulate the gain
control input and finally affect the Doppler performance. In practice, this modulation can be avoided by additional
external filtering (RVCNTL and CVCNTL) at VCNTLM/P pins as Figure 69 shows. However, the external filter's cutoff
frequency cannot be kept too low as this results in low gain response time. Without external filtering, the gain
control response time is typically less than 1 μs to settle within 10% of the final signal level of 1VPP (–6 dBFS)
output as indicated in Figure 48 and Figure 49.
Typical VCNTLM/P signals are generated by an 8bit to 12bit 10MSPS digital to analog converter (DAC) and a
differential operation amplifier. TI’s DACs, such as TLV5626 and DAC7821/11 (10 MSPS/12bit), could be used to
generate TGC control waveforms. Differential amplifiers with output common mode voltage control (e.g.
THS4130 and OPA1632) can connect the DAC to the VCNTLM/P pins. The buffer amplifier can also be configured
as an active filter to suppress low frequency noise. More information can be found in the documents SLOS318
and SBAA150. The VCNTL vs Gain curves can be found in Figure 2. The below table also shows the absolute
gain vs. VCNTL, which may help program DAC correspondingly.
In PW Doppler and color Doppler modes, VCNTL noise should be minimized to achieve the best close-in phase
noise and SNR. Digital VCNTL feature is implemented to address this need in the AFE5803. In the digital VCNTL
mode, no external VCNTL is needed.
Table 10. VCNTLP–VCNTLM vs Gain Under Different LNA and PGA Gain Settings (Low Noise Mode)
Gain (dB)
LNA = 12 dB
PGA = 24 dB
Gain (dB)
LNA = 18 dB
PGA = 24 dB
Gain (dB)
LNA = 24 dB
PGA = 24 dB
Gain (dB)
LNA = 12 dB
PGA = 30 dB
Gain (dB)
LNA = 18 dB
PGA = 30 dB
Gain (dB)
LNA = 24 dB
PGA = 30 dB
V
CNTLP–VCNTLM
(V)
0
36.45
33.91
30.78
27.39
23.74
20.69
17.11
13.54
10.27
6.48
42.45
39.91
36.78
33.39
29.74
26.69
23.11
19.54
16.27
12.48
9.16
48.45
45.91
42.78
39.39
35.74
32.69
29.11
25.54
22.27
18.48
15.16
11.65
9.52
42.25
39.71
36.58
33.19
29.54
26.49
22.91
19.34
16.07
12.28
8.96
48.25
45.71
42.58
39.19
35.54
32.49
28.91
25.34
22.07
18.28
14.96
11.45
9.32
54.25
51.71
48.58
45.19
41.54
38.49
34.91
31.34
28.07
24.28
20.96
17.45
15.32
14.22
13.79
13.8
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1.0
1.1
1.2
1.3
1.4
1.5
3.16
–0.35
–2.48
–3.58
–4.01
–4
5.65
5.45
3.52
3.32
2.42
8.42
2.22
8.22
1.99
7.99
1.79
7.79
2
8
1.8
7.8
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ADC OPERATION
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ADC Clock Distribution
To ensure that the aperture delay and jitter are the same for all channels, the AFE5803 uses a clock tree
network to generate individual sampling clocks for each channel. The clock, for all the channels, are matched
from the source point to the sampling circuit of each of the eight internal ADCs. The variation on this delay is
described in the aperture delay parameter of the output interface timing. Its variation is given by the aperture jitter
number of the same table.
FPGA Clock/
Noisy Clock
n×(20~65)MHz
TI Jitter Cleaner
CDCE72010/
CDCM7005
20~65 MHz
ADC CLK
CDCE72010 has 10
CDCLVP1208
outputs thus the buffer
1-to-8
may not be needed for
CLK Buffer
64CH systems
8 Synchronized
ADC CLKs
B0437-01
Figure 75. ADC Clock Distribution Network
In the single-end case, it is recommended that the use of low jitter square signals (LVCMOS levels, 1.8 V
amplitude). See TI document SLYT075 for further details on the theory.
The jitter cleaner CDCM7005 or CDCE72010 is suitable to generate the AFE5803’s ADC clock and ensure the
performance for the14 bit ADC with 77 dBFS SNR. A clock distribution network is shown in Figure 75.
The AFE5803 can accept differential LVDS, LVPECL, and other differential clock inputs as well as single-ended
clock. AC coupling is required between clock drivers and the AFE5803 clock inputs. When single-ended clock is
used, CLKM should be tied to ground. Common clock configurations are illustrated in Figure 76. Appropriate
termination is recommended to achieve good signal integrity.
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3.3 V
130 Ω
83 Ω
0.1 μF
0.1 μF
3.3 V
130 Ω
AFE
CLOCKs
CDCM7005
CDCE72010
LVPECL
(a) LVPECL Configuration
0.1 μF
100 Ω
0.1 μF
AFE
CLOCKs
CDCE72010
LVDS
(b) LVDS Configuration
0.1μF
0.1μF
0.1μF
AFE
CLOCKs
CLOCK
SOURCE
50 Ω
0.1μF
(c) Transformer Based Configuration
AFE
CMOS CLK
CMOS CLK
Driver
CMOS
(d) CMOS Configuration
S0503-01
Figure 76. Clock Configurations
Special considerations should be applied in such a clock distribution network design. In typical ultrasound
systems, it is preferred that all clocks are generated from a same clock source, such as CW clocks, audio ADC
clocks, RF ADC clock, pulse repetition frequency signal, frame clock and etc. By doing this, interference due to
clock asynchronization can be minimized.
ADC Reference Circuit
The ADC’s voltage reference can be generated internally or provided externally. When the internal reference
mode is selected, the REFP/M becomes output pins and should be floated. When 3[15] =1 and 1[13]=1, the
device is configured to operate in the external reference mode in which the VREF_IN pin should be driven with a
1.4V reference voltage and REFP/M must be left open. Since the input impedance of the VREF_IN is high, no
special drive capability is required for the 1.4V voltage reference
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The digital beam-forming algorithm in an ultrasound system relies on gain matching across all receiver channels.
A typical system would have about 12 octal AFEs on the board. In such a case, it is critical to ensure that the
gain is matched, essentially requiring the reference voltages seen by all the AFEs to be the same. Matching
references within the eight channels of a chip is done by using a single internal reference voltage buffer.
Trimming the reference voltages on each chip during production ensures that the reference voltages are
well-matched across different chips. When the external reference mode is used, a solid reference plane on a
printed circuit board can ensure minimal voltage variation across devices. More information on voltage reference
design can be found in the document SLYT339. The dominant gain variation in the AFE5803 comes from the
VCA gain variation. The gain variation contributed by the ADC reference circuit is much smaller than the VCA
gain variation. Hence, in most systems, using the ADC internal reference mode is sufficient to maintain good gain
matching among multiple AFE5803s. In addition, the internal reference circuit without any external components
achieves satisfactory thermal noise and phase noise performance.
POWER MANAGEMENT
Power/Performance Optimization
The AFE5803 has options to adjust power consumption and meet different noise performances. This feature
would be useful for portable systems operated by batteries when low power is more desired. See the electrical
characteristics table as well as the typical characteristic plots.
Power Management Priority
Power management plays a critical role to extend battery life and ensure long operation time. The AFE5803 has
fast and flexible power down/up control which can maximize battery life. The AFE5803 can be powered down/up
through external pins or internal registers. The following table indicates the affected circuit blocks and priorities
when the power management is invoked. The higher priority controls can overwrite the lower priority ones.In the
device, all the power down controls are logically ORed to generate final power down for different blocks. Thus,
the higher priority controls can cover the lower priority ones. The AFE5803 register settings are maintained when
the AFE5803 is in either partial power down mode or complete power down mode.
Table 11. Power Management Priority
Name
Blocks
All
Priority
High
Pin
PDN_GLOBAL
Pin
PDN_VCA
LNA + VCAT+ PGA
LNA + VCAT+ PGA
LNA + VCAT+ PGA
ADC
Medium
Low
Register
Register
Pin
VCA_PARTIAL_PDN
VCA_COMPLETE_PDN
PDN_ADC
Medium
Medium
Low
Register
Register
Register
Register
ADC_PARTIAL_PDN
ADC_COMPLETE_PDN
PDN_VCAT_PGA
PDN_LNA
ADC
ADC
Medium
Lowest
Lowest
VCAT + PGA
LNA
Partial Power-Up/Down Mode
The partial power up/down mode is also called as fast power up/down mode. In this mode, most amplifiers in the
signal path are powered down, while the internal reference circuits remain active as well as the LVDS clock
circuit, i.e. the LVDS circuit still generates its frame and bit clocks.
The partial power down function allows the AFE5803 to be wake up from a low-power state quickly. This
configuration ensures that the external capacitors are discharged slowly; thus a minimum wake-up time is
needed as long as the charges on those capacitors are restored. The VCA wake-up response is typically about 2
μs or 1% of the power down duration whichever is larger. The longest wake-up time depends on the capacitors
connected at INP and INM, as the wake-up time is the time required to recharge the caps to the desired
operating voltages. For 0.1μF at INP and 15nF at INM can give a wake-up time of 2.5ms. For larger capacitors
this time will be longer. The ADC wake-up time is about 1 μs. Thus the AFE5803 wake-up time is more
dependent on the VCA wake-up time. This also assumes that the ADC clock has been running for at least 50 µs
before normal operating mode resumes. The power-down time is instantaneous, less than 1.0µs.
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This fast wake-up response is desired for portable ultrasound applications in which the power saving is critical.
The pulse repetition frequency of a ultrasound system could vary from 50KHz to 500Hz, while the imaging depth
(i.e., the active period for a receive path) varies from 10 μs to hundreds of us. The power saving can be pretty
significant when a system’s PRF is low. In some cases, only the VCA would be powered down while the ADC
keeps running normally to ensure minimal impact to FPGAs.
In the partial power-down mode, the AFE5803 typically dissipates only 26mW/ch, representing an 80% power
reduction compared to the normal operating mode. This mode can be set using either pins (PDN_VCA and
PDN_ADC) or register bits (VCA_PARTIAL_PDN and ADC_PARTIAL_PDN).
Complete Power-Down Mode
To achieve the lowest power dissipation of 0.7 mW/CH, the AFE5803 can be placed into a complete power-down
mode. This mode is controlled through the registers ADC_COMPLETE_PDN, VCA_COMPLETE_PDN or
PDN_GLOBAL pin. In the complete power-down mode, all circuits including reference circuits within the
AFE5803 are powered down; and the capacitors connected to the AFE5803 are discharged. The wake-up time
depends on the time needed to recharge these capacitors. The wake-up time depends on the time that the
AFE5803 spends in shutdown mode. 0.1μF at INP and 15 nF at INM can give a wake-up time close to 2.5ms.
VCA TEST MODES
The AFE5803 includes multiple test modes to accelerate system development. The ADC test modes have been
discussed in the register description section.
The VCA has a test mode in which the CH7 and CH8 PGA outputs can be brought. By monitoring these PGA
outputs, the functionality of VCA operation can be verified. Some registers are related to this test mode. PGA
Test Mode Enable: Reg59[9]; Buffer Amplifier Power Down Reg59[8]; and Buffer Amplifier Gain Control
Reg54[4:0].
The PGA outputs are connected to the virtual ground pins of the buffer amplifier through 5 kΩ resistors. The PGA
outputs can be monitored at the buffer amplifier outputs when the buffer amplifer is enabled. Note that the
signals at the buffer amplifier outputs are attenuated due to the 5KΩ resistors. The attenuation coefficient is
RINT/EXT/5kΩ. See Table 7 for the RINT configuration.
An alternative way is to measure the PGA outputs directly at the CH8_TEST_OUTM/P and CH7_TEST_OUTM/P
when the buffer amplifier is powered down.
Based on the buffer amplifier configuration, the registers can be set in different ways:
Configuration 1:
In this configuration, the test outputs can be monitored at CH7/8_TEST_OUTP/M
Reg59[9]=1 ;Test mode enabled
Reg59[8]=0 ;Buffer amplifier powered down
Configuration 2:
In this configuration, the test outputs can be monitored at CH7/8_BUFFER_OUTP/M
Reg59[9]=1 ;Test mode enabled
Reg59[8]=1 ;Buffer amplifier powered on
Reg54[4:0]=10H; Internal feedback 2K resistor enabled. Different values can be used as well
Rint/Rext
5 kW
CH7/8_BUFFER_OUTM
CH7/8_BUFFER_OUTP
PGA CH7/8P
PGA CH7/8M
CH7/8_TEST_OUTP
CH7/8_TEST_OUTM
Buffer
Amp
Rint/Rext
5 kW
Figure 77. AFE5803 PGA Test Mode
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POWER SUPPLY, GROUNDING AND BYPASSING
In a mixed-signal system design, power supply and grounding design plays a significant role. The AFE5803
distinguishes between two different grounds: AVSS(Analog Ground) and DVSS(digital ground). In most cases, it
should be adequate to lay out the printed circuit board (PCB) to use a single ground plane for the AFE5803.
Care should be taken that this ground plane is properly partitioned between various sections within the system to
minimize interactions between analog and digital circuitry. Alternatively, the digital (DVDD) supply set consisting
of the DVDD and DVSS pins can be placed on separate power and ground planes. For this configuration, the
AVSS and DVSS grounds should be tied together at the power connector in a star layout. In addition, optical
isolator or digital isolators, such as ISO7240, can separate the analog portion from the digital portion completely.
Consequently they prevent digital noise to contaminate the analog portion. Table 11 lists the related circuit blocks
for each power supply.
Table 12. Supply vs Circuit Blocks
POWER SUPPLY
GROUND
CIRCUIT BLOCKS
LNA, attenuator, PGA with current clamp
and BPF, reference circuits, PGA test mode
buffer, VCA SPI
AVDD (3.3VA)
AVSS
AVDD_5V (5VA)
AVDD_ADC (1.8VA)
DVDD (1.8VD)
AVSS
AVSS
DVSS
LNA, reference circuits
ADC analog and reference circuits
LVDS and ADC SPI
All Bypassing and power supplies for the AFE5803 should be referenced to their corresponding ground planes.
All supply pins should be bypassed with 0.1 µF ceramic chip capacitors (size 0603 or smaller). In order to
minimize the lead and trace inductance, the capacitors should be located as close to the supply pins as possible.
Where double-sided component mounting is allowed, these capacitors are best placed directly under the
package. In addition, larger bipolar decoupling capacitors 2.2 µF to 10 µF, effective at lower frequencies) may
also be used on the main supply pins. These components can be placed on the PCB in proximity (< 0.5 in or
12.7 mm) to the AFE5803 itself.
The AFE5803 has a number of reference supplies needed to be bypassed, such CM_BYP, VHIGH, and
VREF_IN. These pins should be bypassed with at least 1µF; higher value capacitors can be used for better
low-frequency noise suppression. For best results, choose low-inductance ceramic chip capacitors (size 0402, >
1 µF) and place them as close as possible to the device pins.
High-speed mixed signal devices are sensitive to various types of noise coupling. One primary source of noise is
the switching noise from the serializer and the output buffer/drivers. For the AFE5803, care has been taken to
ensure that the interaction between the analog and digital supplies within the device is kept to a minimal amount.
The extent of noise coupled and transmitted from the digital and analog sections depends on the effective
inductances of each of the supply and ground connections. Smaller effective inductance of the supply and
ground pins leads to improved noise suppression. For this reason, multiple pins are used to connect each supply
and ground sets. It is important to maintain low inductance properties throughout the design of the PCB layout by
use of proper planes and layer thickness.
BOARD LAYOUT
Proper grounding and bypassing, short lead length, and the use of ground and power-supply planes are
particularly important for high-frequency designs. Achieving optimum performance with a high-performance
device such as the AFE5803 requires careful attention to the PCB layout to minimize the effects of board
parasitics and optimize component placement. A multilayer PCB usually ensures best results and allows
convenient component placement. In order to maintain proper LVDS timing, all LVDS traces should follow a
controlled impedance design. In addition, all LVDS trace lengths should be equal and symmetrical; it is
recommended to keep trace length variations less than 150mil (0.150 in or 3.81mm).
Additional details on BGA PCB layout techniques can be found in the Texas Instruments Application Report
MicroStar BGA Packaging Reference Guide (SSYZ015B), which can be downloaded from www.ti.com.
48
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Copyright © 2012, Texas Instruments Incorporated
Product Folder Link(s): AFE5803
AFE5803
www.ti.com
SLOS763A –JANUARY 2012–REVISED JANUARY 2012
REVISION HISTORY
Changes from Original (January 2012) to Revision A
Page
•
Changed Table 3 Digital HPF –1dB Corner Frequency vs K and Fs ................................................................................. 27
Copyright © 2012, Texas Instruments Incorporated
Submit Documentation Feedback
49
Product Folder Link(s): AFE5803
PACKAGE OPTION ADDENDUM
www.ti.com
30-Jan-2012
PACKAGING INFORMATION
Status (1)
Eco Plan (2)
MSL Peak Temp (3)
Samples
Orderable Device
Package Type Package
Drawing
Pins
Package Qty
Lead/
Ball Finish
(Requires Login)
AFE5803ZCF
ACTIVE
NFBGA
ZCF
135
160
Green (RoHS
& no Sb/Br)
SNAGCU Level-3-260C-168 HR
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
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In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
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