THS3112CDDAG3 [TI]

Dual, Low-Noise, High Output Current, 110-MHz Amplifier 8-SO PowerPAD 0 to 70;


型号：	THS3112CDDAG3
厂家：	TEXAS INSTRUMENTS
描述：	Dual, Low-Noise, High Output Current, 110-MHz Amplifier 8-SO PowerPAD 0 to 70 放大器光电二极管商用集成电路
文件：	总7页 (文件大小：208K)
中文：	中文翻译
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Amplifiers: Op Amps

Texas Instruments Incorporated

Expanding the usability of

current-feedback amplifiers

By Randy Stephens (Email: r-stephens@ti.com)

Systems Specialist, Member Group Technical Staff

Introduction

Figure 1. VFB test circuit

Although current-feedback (CFB) amplifiers have been

around as long as the widely utilized voltage-feedback (VFB)

amplifiers, their acceptance has been sporadic. One of the

reasons for this is quite simple—they have a different

name and therefore must be difficult and very hard to use.

= 220 pF

= 187 Ω

= 750 Ω

1, 2, 3

This is simply not true. There are numerous papers

comparing the differences between the two amplifier

types that show they are more similar to each other than

different. In fact, for numerous circuits, a CFB amplifier

may actually yield better results due to its inherent slew-

rate advantage, lack of a gain-bandwidth product, and

reasonably low noise for the performance.

+15 V

OUT

THS4012

Almost every paper written about CFB amplifiers cautions

readers that placing a capacitor directly in the feedback path,

without any resistance in series, will cause the CFB ampli-

fier to oscillate. This is true, as the compensation of the

amplifier is tied directly to the feedback impedance. Since a

capacitor has low impedance at high frequencies, this essen-

tially places a short in the feedback path that inadvertently

defeats amplifier compensation, resulting in instability.

Because of this limitation, there are a handful of common

circuits that are not recommended for use with a CFB

amplifier. These include integrators, some types of filters,

and special feedback-compensation techniques. But what

if there was a way to make these circuits work? And what

if the solution was as simple as adding a single component?

This would make it feasible to implement a CFB amplifier

for just about every application for which a VFB amplifier

could be used, with the benefits of the CFB amplifier.

100 Ω

Term

–15 V

50 Ω

Figure 2. CFB test circuit with simple

modification

= 220 pF

= 187 Ω

= 750 Ω

+15 V

Compensation

This article does not explain the compensation theory of

VFB and CFB amplifiers, as there are many papers written

on this topic. The only thing that is important is that

there must be resistance, or impedance, in the feedback

path at the open-loop intersection point to make the CFB

amplifier stable.

OUT

THS3112

100 Ω

Term

50 Ω

–15 V

Figure 1 shows a traditional VFB amplifier, a THS4012,

configured in a noninverting gain of +5 with a simple low-

pass gain filter set at approximately 1 MHz by the straight-

forward 1/(2πR C ) formula.

If a CFB amplifier like the THS3112 is simply dropped

into this circuit, it will oscillate and the circuit will

become useless. A method of compensating the CFB

amplifier in this circuit is to insert a resistance, or imped-

ance (Z), in the feedback path as shown in Figure 2.

It can easily be seen that regardless of the impedance

amplifier, can now be essentially any resistance desired.

The reader should keep in mind that this is still a high-

speed amplifier with speeds over 100 MHz; so the feedback

resistance should always be kept less than a few kilohms

to minimize the effects of parasitic capacitances on the

overall circuit. Conversely, minimizing the resistance too

much will place too much of a load on the amplifier,

typically degrading performance.

of the feedback path represented by R and C , the

impedance Z is in the amplifier’s feedback loop dictating

the compensation of the amplifier. The interesting thing

about this configuration is that the feedback resistance

One of the drawbacks of adding the impedance Z in this

manner is that the summing node at the inverting terminal

is now separated from the virtual summing node. This can

(R ), which normally dictates the compensation of the

Analog Applications Journal

3Q 2003

www.ti.com/sc/analogapps

Analog and Mixed-Signal Products

Amplifiers: Op Amps

Texas Instruments Incorporated

introduce errors into the system due to

the bias current and the dynamic signal

current flowing through this impedance;

but these effects are reasonably small as

long as the impedance is minimized.

Adding impedance Z can affect input

offset voltage due to the dc input bias

current, which is typically 1 to 10 µA,

multiplied by the impedance Z. This

resulting voltage gets multiplied by the

noise gain of the circuit. Additionally, when

a signal appears at the output, the CFB

amplifier (as the name implies) relies on

an error current flowing through the

Figure 3. Frequency responses with resistors (gain = +5)

Z = 200 Ω

Z = 475 Ω

Z = 681 Ω

inverting node through the impedance Z,

producing a signal error. However, since

the transimpedance of most CFB ampli-

fiers is well over 100 kΩ and sometimes as

high as several megohms, this error is also

minimized if the impedance is kept low. The

drift of this circuit now also relies on the

temperature characteristics of impedance

Z and should not be used as a precision

amplifier; but most CFB amplifiers are not

used as precision amplifiers anyway due to

–5

–10

–15

Z = 1 kΩ

THS4012

10 k

100 k

1 M

10 M

Frequency (Hz)

100 M

1 G

their inherent topology limitations. Overall, these issues

are minimal and, for most systems, can be effectively

ignored in favor of the CFB amplifier’s advantages as

previously stated.

stated previously. This shows that there is a reasonably wide

range of acceptable values for Z and does not imply that the

selection for Z is highly critical. Figure 3 also illustrates a

common trait for current-feedback amplifiers—as the feed-

back impedance is decreased, the peaking will increase. If

the impedance is too low, there is a good chance that the

circuit will become unstable and oscillate, as illustrated by

the response when Z = 200 Ω.

Testing with different Z values

The easiest way to see if the circuit is stable is to use a

network analyzer frequency sweep. Instability can typical-

ly be seen as sharp rises in the frequency response at the

amplifier’s bandwidth limitations. If the peaking is smooth,

or there is no peak, then the amplifier should be stable.

Figure 3 shows the frequency response of the system with

different values of resistors for the variable Z.

The response of the THS4012 is also shown for reference

to easily compare the performance of the two systems. It

is interesting that no matter what resistance is used for Z,

the responses below 20 MHz look identical to each other.

This is the ultimate goal of this configuration—no differ-

ences in signal performance. For the stability part of the

circuit, the area above 20 MHz must be examined.

Output noise

One element that may be very important in a system is the

output noise. Adding a resistance in the manner discussed

only makes the output noise worse. The inverting current

noise of the amplifier goes through the resistance at Z and

creates a voltage noise. This noise then becomes multiplied

by the circuit’s gain, which is frequency-dependent.

For a CFB amplifier, the inverting current noise is typi-

cally the highest noise component of the amplifier. Although

the CFB amplifier voltage noise is inherently very low,

——

typically less than 3 nV/ Hz, the inverting current noise of

√

——

Examining the circuits in Figures 1 and 2 shows us that

the feedback impedance is dictated by the capacitor CF.

Above 20 MHz, this impedance is very small—essentially

creating a short from the output to the summing node. This

configuration is commonly referred to as a unity buffer with

most CFB amplifiers is generally around 15 to 20 pA/ Hz.

√

The noninverting current noise is only noticeable if the

source impedance is high. Using a 50-Ω environment

minimizes the noninverting current noise.

The THS3112 was designed to have very low noise. The

——

the signal gain set to 1. The data sheet for the THS3112

voltage noise is 2.2 nV/ Hz, the noninverting current noise

√

——

recommends that, in a gain of +1 under the circuit condi-

tions utilized, the feedback resistance be 1 kΩ. Thus, it is

no surprise to see that when Z = 1 kΩ, the response looks

very smooth and well behaved, indicating a very stable

system. However, when Z = 681 Ω, the response also looks

very reasonable and helps minimize the potential issues

is 2.9 pA/ Hz, and the critical inverting current noise is a

√

——

low 10.8 pA/ Hz. However, multiplying the inverting current

noise by 1 kΩ and then multiplying by the gain can alone

produce a very substantial output noise of about 54 nV/ Hz

in the pass band. To quantify the output noise of the system,

the circuits shown in Figures 1 and 2 were tested for output

√

——

√

Analog Applications Journal

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Amplifiers: Op Amps

Texas Instruments Incorporated

noise (see Figure 4). For comparison, the THS4012, with a

——

Keep in mind that the THS3112 has very low overall

noise but that many other CFB amplifiers will probably

produce much higher noise. The only way to get around

this is if the unity-gain stability of the amplifier requires a

very small resistor of, say, only 500 Ω or less. But what if

there was another way to make the CFB amplifier stable

and have low noise at the same time?

Fundamentally speaking, the circuit needs high impedance

within the feedback path only at the amplifier’s bandwidth

limit. At frequencies below this point, it really does not

matter what the impedance is, and the amplifier will work

fine. The issues stated previously are also

respectable voltage noise of 7.5 nV/ Hz and both current

√

——

noises of 1 pA/ Hz, is also shown in Figure 4.

√

Note that the output noise of the THS4012 is the same

as when using the THS3112 with Z = 475 Ω. Again, these

responses are just like those of a VFB amplifier in the tradi-

tional configuration, showing that the basic functionality is

sound—there are no differences between a VFB amplifier

and this configuration. Figure 4 shows that although using

Z = 1 kΩ produces a very stable amplifier, the output

——

noise is 20 nV/ Hz higher than that of the THS4012.

√

minimized, resulting in an even better

system than one using pure resistors.

The first solution that comes to mind is

Figure 4. Output noise (gain = +5)

to use an inductor. Inductors have low

impedance at low frequencies and high

impedance at high frequencies—exactly

what is desired; but their relatively large

size and high cost are generally considered

Z = 1 kΩ

prohibitive. An alternative component

that minimizes these disadvantages and

still functions the same is the ferrite chip.

Z = 681 Ω

THS4012; also Z = 475 Ω

Testing with ferrite chips

used for Z

Ferrite chips have been available for several

years, are relatively low-cost, and are

available in very small sizes—0402 and

larger. Although several manufacturers

produce ferrite chips, testing was done

with what was available in the test lab—

Z = 200 Ω

ferrite chips from Murata’s BLM series.

Examining the impedance characteristics

of these ferrites revealed several possible

components that could be utilized.

10 k

100 k

Frequency (Hz)

1 M

10 M

The first factor in determining the proper

component was the ferrite’s impedance at

the amplifier’s bandwidth limit. For the

THS3112, this implied an impedance of

at least 600 Ω at about 150 MHz to meet

stability. This can vary, as the first test

results showed (see Figure 3).

Additionally, the Q of the ferrite chips

varies from grade to grade. Some have a

low Q with a fairly smooth rise to the

resonance point that then subsides due to

inherent properties and parasitics, while

other chips have a relatively high Q with a

sharp rise and fall in impedance associated

with them. Although either style may

meet the impedance requirements, testing

was required to see if this Q had an effect

on the circuit. Again, the best way to show

the results was to graph the frequency

response of the system, as shown in

Figure 5. The responses below 10 MHz

were all identical to the original configu-

ration. This figure concentrates on the

stability portion of the responses above

10 MHz. For comparison purposes, the

681-Ω, pure-resistance response is shown.

Figure 5. Frequency responses above 10 MHz with

ferrite chips (gain = +5)

Z = BLM18HD601SN1

Z = BLM18HG601SN1

Z = 681 Ω

–5

–10

–15

–20

Z = BLM18AG601SN1

10 M

100 M

Frequency (Hz)

1 G

Analog Applications Journal

3Q 2003

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Analog and Mixed-Signal Products

Amplifiers: Op Amps

Texas Instruments Incorporated

Although all of these ferrite chips have

the same impedance at 100 MHz (600 Ω),

they produced different results. The HD

series high-Q chip shows a very narrow

and large peak that will most likely result

in instability and oscillations. The AG and

HG series low-Q chips both performed

about the same, and either one would

probably produce acceptable results. The

only difference is that the HG series has

impedance at higher frequencies and

would probably be better suited for use

with very high-speed CFB amplifiers such

as the OPA685 or the THS3202.

Notice that the pure resistance has a

lower response peak than the ferrite chips.

Coupled with the fact that the HD series

has a high Q and a high peak, this implies

that the slope of the impedance at the

amplifier’s bandwidth is a factor for stabil-

ity. This makes a lot of sense; as it is well

known that for any amplifier, if a zero

intersects the amplifier’s open-loop

Figure 6. Responses with AG series ferrite chips (gain = +5)

Z = Ferrite Chip

BLM18AGxxxSN1 Series

xxx = 221

xxx = 471

xxx = 601

xxx = 102

–5

–10

–15

10 k

100 k

1 M

10 M

100 M

1 G

Frequency (Hz)

response at a rate of closure of 40 dB/

decade, large peaking and oscillations will

most likely result. For this circuit config-

uration, if the impedance of Z has a large

slope that intersects the transimpedance

curve at essentially a rate of closure of

40 dB/decade, peaking and oscillations

also will most likely occur. By comparison,

a resistor intersects the transimpedance

curve at a rate of closure of 20 dB/decade,

resulting in a stable response. Even though

the low-Q ferrite beads have some slope

related to their impedance, the rate of

closure is much lower than 40 dB/decade,

providing improved stability. Nevertheless,

minimizing this intersection rate of closure

as much as possible should produce

acceptable results.

Figure 7. Output noise comparison (gain = +5)

Z = 681 Ω

THS4012

Z = 332 Ω

Z = All Ferrite Chips

To further expand on the usefulness of

the ferrite chips, more testing was done

utilizing the AG series in the circuit, as

shown in Figure 6.

This figure shows that, just like the

10 k

100 k

1 M

10 M

results for the pure resistor, the higher

the impedance is, the lower the peaking.

How does this affect the output noise of

the system? Figure 7 shows the output

noise when the ferrite chips were used,

along with the output noise of the THS4012

and some of the original resistor configurations.

As expected, due to the low frequency impedance of the

ferrite chips, the noise is extremely low. This noise was the

same regardless of which ferrite was used. If noise above

10 MHz was important, the impedance of these ferrite

chips would start to increase the output noise to the same

extent as resistors. These tests show that there are several

advantages of using ferrite chips over resistors.

Frequency (Hz)

Inverting gain configuration

All of the testing discussed so far was done with the non-

inverting gain configuration. This configuration forces the

inverting node voltage to move proportionally to the input

voltage applied. So how does the system work in the

inverting gain configuration where the inverting node is

held at a virtual ground? The easy answer is that it works

Analog Applications Journal

Analog and Mixed-Signal Products

www.ti.com/sc/analogapps

3Q 2003

Amplifiers: Op Amps

Texas Instruments Incorporated

Figure 8. Inverting gain of 5 VFB configuration

Figure 9. Inverting gain of 5 CFB configuration

= 220 pF

= 750 Ω

+15 V

150 Ω

OUT

Term

THS4012

THS3112

75 Ω

100 Ω

–15 V

gain must be above unity gain, or 0 dB. As long as the peak

is below 0 dB, oscillations should not occur. As in the non-

inverting case, using 200 Ω shows a large narrow peak that

will most likely result in stability issues and/or oscillations.

However, notice that above 10 MHz the same general

shape occurs for both the CFB and VFB amplifiers. This is

caused by the amplifiers’ input and output impedances

becoming very high above their bandwidth limit. When

this occurs, there is a path for the input signal to flow

through R , through C , and then to feed forward to the

exactly the same as before. Figures 8 and 9 show the test

circuits for this configuration. The signal gain was kept at

a gain of 5.

The same concepts apply for this CFB configuration as

for the noninverting configuration. The advantage of this

circuit is that the attenuation is not limited to unity gain,

or 0 dB, like the noninverting gain circuit. Figure 10 shows

the frequency responses of this configuration with varying

pure resistor values for Z. The THS4012 response is shown

for comparison purposes.

load. Of course, the amplifiers’ own input and output

capacitances also affect the amount of feed-through in the

circuit; but it is important to remember that this occurs

above the amplifiers’ usable bandwidths.

Just as for the noninverting configuration, using ferrite

chips has several advantages for the inverting configuration.

As expected, the responses all look comparable to each

other below 10 MHz. Additionally, the resistance values

affect the stability and again show that the higher the

resistance is, the better the stability. Using a resistance as

low as 475 Ω actually shows respectable performance in this

configuration. Remember that for oscillations to occur, the

Figure 10. Frequency responses with resistors (gain = –5)

Z = 200 Ω

–5

THS4012

Z = 475 Ω

–10

–15

–20

Z = 1 kΩ

–

10 k

100 k

1 M 10 M

Frequency (Hz)

100 M

1 G

Analog Applications Journal

3Q 2003

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Analog and Mixed-Signal Products

Amplifiers: Op Amps

Texas Instruments Incorporated

Figure 11 shows the frequency responses

of several of these chips. Figure 12 shows

the results of using various ferrite chips

from the same AG family.

Figure 11. Frequency responses above 10 MHz

with ferrite chips (gain = –5)

As expected, all of these graphs show

the same type of results obtained with

the noninverting configuration. Using a

low-Q ferrite chip with high impedance

will result in a stable system. Although the

noise plots for this configuration are not

presented here, they will show the same

type of results as the noninverting configu-

ration; using ferrite chips will have the

lowest output noise of any configuration.

Z = BLM18HD601SN1

Z = BLM18AG601SN1

Z = BLM18HG601SN1

–5

–10

–15

–20

–25

Conclusion

Although this article shows only two con-

figurations with capacitors in the feedback

path, it shows the fundamental feasibility

of this compensation technique. While

resistors do work very well, producing the

most stable responses, the drawbacks of

the output noise coupled with the dc and

ac errors may limit some of the applications.

Using ferrite chips helps alleviate many

of these issues, producing the lowest noise

of all with no dc errors or in-band ac sig-

nal errors; and stability is almost as good

as when utilizing resistors. It is important

to choose the proper ferrite chip with the

amplifier; but this is considered normal

procedure for any circuit design and is no

more difficult than selecting the right

amplifier for the system.

Z = 681 Ω

10 M

100 M

Frequency (Hz)

1 G

Figure 12. Frequency responses with AG series

ferrite chips (gain = –5)

xxx = 221

xxx = 601

This simple technique helps eliminate

one of the major drawbacks of using the

CFB amplifier while allowing any system

to enjoy many of its benefits. Designers of

multiple feedback filters, for example, once

limited to the use of VFB amplifiers, can

now take advantage of the superior slew

rates and lack of gain-bandwidth product

characteristics found in the CFB amplifier.

xxx = 471

–5

–10

–15

–20

Z = Ferrite Chip

BLM18AGxxxSN1 Series

xxx = 102

References

–

For more information related to this article,

you can download an Acrobat Reader file

at www-s.ti.com/sc/techlit/litnumber and

replace “litnumber” with the TI Lit. #

for the materials listed below.

10 k

100 k

1 M

10 M

100 M

1 G

Frequency (Hz)

Document Title

1. “Voltage Feedback Vs. Current Feedback

Op Amps,” Application Report . . . . . . . . . . . . . .slva051

2. “The Current-Feedback Op Amp: A High-

Speed Building Block,” Application Bulletin . . .sboa076

3. “Current Feedback Amplifiers: Review,

TI Lit. #

Related Web sites

analog.ti.com

www.ti.com/sc/device/partnumber

Replace partnumber with OPA685, THS3112, THS3202 or

THS4012

Stability Analysis, and Applications,”

Application Bulletin . . . . . . . . . . . . . . . . . . . . . . .sboa081

4. “Low-Noise, High-Speed Current Feedback

Amplifiers,” Data Sheet . . . . . . . . . . . . . . . . . . . .slos385

5. “Effect of Parasitic Capacitance in Op Amp

Circuits,” Application Report . . . . . . . . . . . . . . .sloa013

Analog Applications Journal

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3Q 2003

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SLYT099

THS3112CDDAG3 [TI]

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