UCC21732 [TI]
UCC21732 10-A Source/Sink Reinforced Isolated Single Channel Gate Driver for SiC/IGBT with Active Protection, Isolated Analog Sensing and High-CMTI;型号: | UCC21732 |
厂家: | TEXAS INSTRUMENTS |
描述: | UCC21732 10-A Source/Sink Reinforced Isolated Single Channel Gate Driver for SiC/IGBT with Active Protection, Isolated Analog Sensing and High-CMTI 栅 双极性晶体管 |
文件: | 总55页 (文件大小:1876K) |
中文: | 中文翻译 | 下载: | 下载PDF数据表文档文件 |
UCC21732-Q1
SLUSD77 – AUGUST 2020
UCC21732 10-A Source/Sink Reinforced Isolated Single Channel Gate Driver
for SiC/IGBT with Active Protection, Isolated Analog Sensing and High-CMTI
1 Features
3 Description
•
•
•
•
•
•
•
•
5.7-kVRMS single channel isolated gate driver
SiC MOSFETs and IGBTs up to 2121Vpk
33-V maximum output drive voltage (VDD-VEE)
±10-A drive strength and split output
150-V/ns minimum CMTI
270-ns response time fast overcurrent protection
Internal 2-level turn-off when fault happens
Isolated analog sensor with PWM output for
– Temperature sensing with NTC, PTC or thermal
diode
– High voltage DC-Link or phase voltage
Alarm FLT on over current and reset from RST/EN
Fast enable/disable response on RST/EN
Reject <40-ns noise transient and pulse on input
pins
12-V VDD UVLO with power good on RDY
– VDD UVLO 12 V
Inputs/outputs with over/under-shoot transient
voltage Immunity up to 5 V
130-ns (maximum) propagation delay and 30-ns
(maximum) pulse/part skew
SOIC-16 DW package with creepage and
clearance distance > 8 mm
The UCC21732 is a galvanic isolated single channel
gate driver designed for SiC MOSFETs and IGBTs up
to 2121-V DC operating voltage with advanced
protection features, best-in-class dynamic
performance and robustness. UCC21732 has up to
±10-A peak source and sink current.
The input side is isolated from the output side with
SiO2 capacitive isolation technology, supporting up to
1.5-kVRMS working voltage, 12.8-kVPK surge immunity
with longer than 40 years Isolation barrier life, as well
as providing low part-to-part skew , >150V/ns
common mode noise immunity (CMTI).
•
•
•
The UCC21732 includes the state-of-art protection
features, such as fast overcurrent and short circuit
detection, shunt current sensing support, fault
reporting, active miller clamp, input and output side
power supply UVLO to optimize SiC and IGBT
switching behavior and robustness. The isolated
analog to PWM sensor can be utilized for easier
temperature or voltage sensing, further increasing the
drivers' versatility and simplifying the system design
effort, size and cost.
•
•
•
•
•
Device Information (1)
PART NUMBER
PACKAGE
BODY SIZE (NOM)
Operating junction temperature –40°C to 150°C
UCC21732
DW SOIC-16
10.3 mm × 7.5 mm
2 Applications
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
•
•
•
•
•
•
Traction inverter for EVs
On-board charger and charging pile
DC/DC Converter for HEV/EVs
Industrial motor drives
Server, telecom, and industrial power supplies
Uninterruptible power supplies (UPS)
Device Pin Configuration
APWM
VCC
RST/EN
FLT
AIN
OC
1
2
3
4
5
6
7
8
16
15
COM
OUTH
VDD
14
13
12
11
10
9
RDY
INÅ
OUTL
CLMPE
VEE
IN+
GND
Not to scale
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
UCC21732-Q1
SLUSD77 – AUGUST 2020
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Table of Contents
1 Features............................................................................1
2 Applications.....................................................................1
3 Description.......................................................................1
4 ...........................................................................................2
5 Pin Configuration and Functions...................................3
6 Specifications.................................................................. 5
6.1 Absolute Maximum Ratings........................................ 5
6.2 ESD Ratings............................................................... 5
6.3 Recommended Operating Conditions.........................5
6.4 Thermal Information....................................................6
6.5 Power Ratings.............................................................6
6.6 Insulation Specifications............................................. 7
6.7 Safety-Related Certifications...................................... 8
6.8 Safety Limiting Values.................................................8
6.9 Electrical Characteristics.............................................9
6.10 Switching Characteristics........................................11
6.11 Insulation Characteristics Curves............................12
6.12 Typical Characteristics............................................13
7 Parameter Measurement Information..........................18
7.1 Propagation Delay.................................................... 18
7.2 Input Deglitch Filter...................................................19
7.3 Active Miller Clamp................................................... 20
7.4 Under Voltage Lockout (UVLO)................................ 21
7.5 OC (Over Current) Protection................................... 23
8 Detailed Description......................................................24
8.1 Overview...................................................................24
8.2 Functional Block Diagram.........................................25
8.3 Feature Description...................................................25
8.4 Device Functional Modes..........................................32
9 Applications and Implementation................................33
9.1 Application Information............................................. 33
9.2 Typical Application.................................................... 34
10 Power Supply Recommendations..............................48
11 Layout...........................................................................49
11.1 Layout Guidelines................................................... 49
11.2 Layout Example...................................................... 50
12 Device and Documentation Support..........................51
12.1 Documentation Support.......................................... 51
12.2 Receiving Notification of Documentation Updates..51
12.3 Support Resources................................................. 51
12.4 Trademarks.............................................................51
12.5 Electrostatic Discharge Caution..............................51
12.6 Glossary..................................................................51
13 Mechanical, Packaging, and Orderable
Information.................................................................... 51
4
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
DATE
REVISION
NOTES
August 2020
*
Initial release
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5 Pin Configuration and Functions
APWM
VCC
RST/EN
FLT
AIN
OC
1
2
3
4
5
6
7
8
16
15
COM
OUTH
VDD
14
13
12
11
10
9
RDY
INÅ
OUTL
CLMPE
VEE
IN+
GND
Not to scale
Figure 5-1. UCC21710 DW SOIC (16) Top View
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Pin Functions
PIN
I/O(1)
DESCRIPTION
NAME
AIN
NO.
1
I
I
Isolated analog sensing input, parallel a small capacitor to COM for better noise immunity
OC
2
Over current detection pin, support lower threshold for SenseFET, DESAT, and Shunt resistor sensing
Common ground reference, connecting to emitter pin for IGBT and source pin for SiC-MOSFET
Gate driver output pull up
COM
OUTH
3
P
O
4
Positive supply rail for gate drive voltage, Bypassing a >220nF capacitor to COM to support specified gate
driver source peak current capability
VDD
5
P
OUTL
6
7
O
O
Gate driver output pull down
CLMPE
External Active miller clamp, connecting this pin to the gate of the external miller clamp MOSFET
Negative supply rail for gate drive voltage. Bypassing a >220nF capacitor to COM to support specified gate
driver sink peak current capability
VEE
8
P
GND
IN+
9
P
I
Input power supply and logic ground reference
Non-inverting gate driver control input
Inverting gate driver control input
10
11
IN–
I
Power good for VCC-GND and VDD-COM. RDY is open drain configuration and can be paralleled with other
RDY signals
RDY
FLT
12
13
O
O
Active low fault alarm output upon over current or short circuit. FLT is in open drain configuration and can be
paralleled with other faults
The RST/EN serves two purposes:
1) Enable / shutdown of the output side. The FET is turned off by a general turn-off, if terminal EN is set to
low;
RST/EN
14
I
2) Resets the OC condition signaled on FLT pin. if terminal RST/EN is set to low for more than 1000ns. A
reset of signal FLT is asserted at the rising edge of terminal RST/EN.
For automatic RESET function, this pin only serves as an EN pin. Enable / shutdown of the output side. The
FET is turned off by a general turn-off, if terminal EN is set to low.
VCC
15
16
P
Input power supply from 3V to 5.5V, bypassing a >100nF capacitor to GND
Isolated Analog Sensing PWM output
APWM
O
(1) P = Power, G = Ground, I = Input, O = Output
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)(1)
PARAMETER
MIN
–0.3
MAX
6
UNIT
V
VCC
VDD
VEE
VMAX
VCC – GND
VDD – COM
VEE – COM
VDD – VEE
–0.3
36
V
–17.5
0.3
V
–0.3
36
V
DC
GND–0.3
GND–5.0
–0.3
VCC
VCC+5.0
5
V
IN+, IN–, RST/EN
Transient, less than 100 ns(2)
V
AIN
OC
Reference to COM
Reference to COM
V
-0.3
6
DC
VEE–0.3
VEE–5.0
–0.3
VDD
VDD+5.0
5
V
V
OUTH, OUTL
Transient, less than 100 ns(2)
CLMPE
Reference to VEE
V
RDY, FLT, APWM
GND–0.3
VCC
20
V
I FLT , IRDY
IAPWM
TJ
FLT, and RDY pin input current
APWM pin output current
mA
mA
°C
°C
20
Junction temperature range
Storage temperature range
–40
–65
150
150
Tstg
(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) Values are verified by characterization on bench.
6.2 ESD Ratings
VALUE
±4000
±1500
UNIT
Human-body model (HBM), per AEC Q100-002(1)
Charged-device model (CDM), per AEC Q100-011
V(ESD)
Electrostatic discharge
V
(1) AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
PARAMETER
VCC
MIN
3.0
MAX
5.5
UNIT
VCC–GND
VDD–COM
VDD–VEE
V
V
V
VDD
13
33
VMAX
–
33
High level input voltage
Low level input voltage
0.7×VCC
0
VCC
0.3×VCC
4.5
IN+, IN–, RST/EN
Reference to GND
V
AIN
tRST/EN
TA
Reference to COM
0.6
V
Minimum pulse width that reset the fault
Ambient Temperature
1000
–40
ns
°C
°C
125
150
TJ
Junction temperature
–40
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UNIT
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6.4 Thermal Information
UCC21732
DW (SOIC)
16
THERMAL METRIC(1)
RθJA
RθJC(top)
RθJB
ψJT
Junction-to-ambient thermal resistance
68.3
°C/W
°C/W
°C/W
°C/W
°C/W
Junction-to-case (top) thermal resistance
Junction-to-board thermal resistance
27.5
32.9
Junction-to-top characterization parameter
Junction-to-board characterization parameter
14.1
ψJB
32.3
(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
6.5 Power Ratings
PARAMETER
TEST CONDITIONS
Value
UNIT
PD
Maximum power dissipation (both sides)
985
mW
Maximum power dissipation by
transmitter side
PD1
VCC = 5V, VDD-COM = 20V, COM-VEE = 5V, IN+/- = 5V, 150kHz,
50% Duty Cycle for 10nF load, Ta=25oC
20
mW
mW
Maximum power dissipation by receiver
side
PD2
965
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6.6 Insulation Specifications
PARAMETER
GENERAL
TEST CONDITIONS
VALUE
UNIT
CLR
CPG
External clearance(1)
Shortest terminal-to-terminal distance through air
> 8
> 8
mm
mm
Shortest terminal-to-terminal distance across the
package surface
External creepage(1)
Minimum internal gap (Internal clearance) of the
double insulation (2 × 0.0085 mm)
DTI
CTI
Distance through the insulation
> 17
µm
V
Comparative tracking index
Material group
DIN EN 60112 (VDE 0303-11); IEC 60112
According to IEC 60664–1
> 600
I
Rated mains voltage ≤ 300 VRMS
Rated mains voltage ≤ 600 VRMS
Rated mains voltage ≤ 1000 VRMS
I-IV
I-IV
I-III
Overvoltage Category per IEC 60664–1
DIN V VDE V 0884-11 (VDE V 0884-11):2017-01(2)
VIORM Maximum repetitive peak isolation voltage AC voltage (bipolar)
2121
1500
2121
VPK
VRMS
VDC
AC voltage (sine wave) Time dependent dielectric
breakdown (TDDB) test
VIOWM
Maximum isolation working voltage
DC voltage
VTEST=VIOTM, t = 60 s (qualification test)
VTEST=1.2 x VIOTM, t = 1 s (100% production test)
VIOTM
Maximum transient isolation voltage
Maximum surge isolation voltage(3)
8000
8000
VPK
VPK
Test method per IEC 62368-1, 1.2/50 µs waveform,
VTEST = 1.6 × VIOSM = 12800 VPK (qualification)
VIOSM
Method a: After I/O safety test subgroup 2/3, Vini
=
VIOTM, tini = 60 s; Vpd(m) = 1.2 × VIORM = 2545 VPK
tm = 10 s
,
≤ 5
≤ 5
≤ 5
Method a: After environmental tests subgroup 1,
Vini = VIOTM, tini = 60 s; Vpd(m) = 1.6 × VIORM = 3394
VPK, tm = 10 s
qpd
Apparent charge(4)
pC
Method b1: At routine test (100% production) and
preconditioning (type test) Vini = VIOTM, tini = 1 s;
Vpd(m) = 1.875 × VIORM = 3977 VPK, tm = 1 s
CIO
RIO
Barrier capacitance, input to output(5)
Insulation resistance, input to output(5)
VIO = 0.5 sin (2πft), f = 1 MHz
VIO = 500 V, TA = 25°C
~ 1
≥ 1012
≥ 1011
≥ 109
pF
Ω
VIO = 500 V, 100°C ≤ TA ≤ 125°C
VIO = 500 V at TS = 150°C
Pollution degree
Climatic category
2
40/125/21
UL 1577
VTEST = VISO = 5700 VRMS, t = 60 s (qualification);
VTEST = 1.2 × VISO = 6840 VRMS, t = 1 s (100%
production)
VISO
Withstand isolation voltage
5700
VRMS
(1) Apply creepage and clearance requirements according to the specific equipment isolation standards of an application. Care must be
taken to maintain the creepage and clearance distance of a board design to ensure that the mounting pads of the isolator on the
printed circuit board (PCB) do not reduce this distance. Creepage and clearance on a PCB become equal in certain cases. Techniques
such as inserting grooves and ribs on the PCB are used to help increase these specifications.
(2) This coupler is suitable for safe electrical insulation only within the safety ratings. Compliance with the safety ratings shall be ensured
by means of suitable protective circuits.
(3) Testing is carried out in air or oil to determine the intrinsic surge immunity of the isolation barrier.
(4) Apparent charge is electrical discharge caused by a partial discharge (pd).
(5) All pins on each side of the barrier tied together creating a two-terminal device
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6.7 Safety-Related Certifications
VDE
UL
CSA
CQC
TUV
Plan to certify according Plan to certify
to DIN V VDE V 0884-11 according to
Plan to certify according to
EN 61010-1:2010 (3rd Ed) and
EN 60950-1:2006/A11:2009/
A1:2010/
Plan to certify according to
CSA Component Acceptance Plan to certify according to
Notice 5A, IEC 60950-1, and
IEC 60601-1
(VDE V
UL 1577
0884-11):2017-01;
DIN EN 61010-1 (VDE
0411-1):2011-07
Component
Recognition
Program
GB4943.1-2011
A12:2011/A2:2013
Isolation Rating of 5700 VRMS
;
5700 VRMS Reinforced
insulation per
EN 61010-1:2010 (3rd Ed) up
to working voltage of 1000
VRMS
5700 VRMS Reinforced
insulation per
EN 60950-1:2006/A11:2009/
A1:2010/
Reinforced insulation
Maximum transient
isolation voltage, 8000
Reinforced insulation per CSA
60950-1- 07+A1+A2 and IEC
60950-1 (2nd Ed.), 1450 VRMS
max working voltage (pollution
degree 2, material group I) ;
2 MOPP (Means of Patient
Protection) per CSA
Reinforced Insulation, Altitude
≤ 5000m, Tropical climate, 400
VRMS maximum working
voltage
VPK
;
Single
Maximum repetitive peak protection,
isolation voltage, 2121
VPK
5700 VRMS
;
Maximum surge isolation
voltage, 8000 VPK
60601-1:14 and IEC 60601-1
Ed. 3.1, 250 VRMS (354 VPK
max working voltage
)
A12:2011/A2:2013 up to
working voltage of 1450 VRMS
Certification
Completed,
Certification
Number:
Certification Planned
Certification Planned
Certification Planned
Certification Planned
E181974 Vol 4
Sec 9
6.8 Safety Limiting Values
Safety limiting(1) intends to minimize potential damage to the isolation barrier upon failure of input or output
circuitry. A failure of the I/O can allow low resistance to ground or the supply and, without current limiting,
dissipate sufficient power to overheat the die and damage the isolation barrier, potentially leading to secondary
system failures.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNIT
RθJA =68.3°C/W, VDD = 15V, VEE=-5V, TJ = 150°C, TA
25°C
=
=
=
61
Safety input, output, or supply
current
IS
mA
49
RθJA =68.3°C/W, VDD = 20V, VEE=-5V, TJ = 150°C, TA
25°C
RθJA =68.3°C/W, VDD = 20V, VEE=-5V, TJ = 150°C, TA
25°C
PS
TS
Safety input, output, or total power
Safety temperature
1220
150
mW
°C
(1) The safety-limiting constraint is the maximum junction temperature specified in the data sheet. The power dissipation and junction-to-
air thermal impedance of the device installed in the application hardware determines the junction temperature. The assumed junction-
to-air thermal resistance in the Section 6.4 table is that of a device installed on a high-K test board for leaded surface-mount packages.
The power is the recommended maximum input voltage times the current. The junction temperature is then the ambient temperature
plus the power times the junction-to-air thermal resistance. These limits vary with the ambient temperature, the junction-to-air thermal
resistance, and the power supply voltages in different applications.
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6.9 Electrical Characteristics
VCC=3.3V or 5.0V, 1uF capacitor from VCC to GND, VDD–COM=20V, 18V or 15V, COM–VEE =0V, 5V, 8V or
15V, CL=100pF, –40°C<TJ<150°C (unless otherwise noted)(1) (2)
.
PARAMETER
VCC UVLO THRESHOLD AND DELAY
VVCC_ON
TEST CONDITIONS
MIN
TYP
MAX
UNIT
2.55
2.35
2.7
2.5
0.2
10
2.85
2.65
VVCC_OFF
VVCC_HYS
tVCCFIL
VCC–GND
V
VCC UVLO Deglitch time
tVCC+ to OUT
tVCC– to OUT
tVCC+ to RDY
tVCC– to RDY
VCC UVLO on delay to output high
VCC UVLO off delay to output low
VCC UVLO on delay to RDY high
VCC UVLO off delay to RDY low
28
5
37.8
10
50
15
50
15
IN+ = VCC, IN– = GND
µs
30
5
37.8
10
RST/EN = VCC
VDD UVLO THRESHOLD AND DELAY
VVDD_ON
10.5
9.9
12.0
10.7
0.8
5
12.8
11.8
VVDD_OFF
VVDD_HYS
tVDDFIL
VDD–COM
V
VDD UVLO Deglitch time
tVDD+ to OUT
tVDD– to OUT
tVDD+ to RDY
tVDD– to RDY
VDD UVLO on delay to output high
VDD UVLO off delay to output low
VDD UVLO on delay to RDY high
VDD UVLO off delay to RDY low
2
5
8
10
15
15
IN+ = VCC, IN– = GND
RST/EN = FLT=High
5
µs
10
10
VCC, VDD QUIESCENT CURRENT
OUT(H) = High, fS = 0Hz, AIN=2V
OUT(L) = Low, fS = 0Hz, AIN=2V
OUT(H) = High, fS = 0Hz, AIN=2V
OUT(L) = Low, fS = 0Hz, AIN=2V
2.5
1.45
3.6
3
2
4
2.75
5.9
IVCCQ VCC quiescent current
mA
mA
4
IVDDQ
VDD quiescent current
3.1
3.7
5.3
LOGIC INPUTS — IN+, IN– and RST/EN
VINH
VINL
VINHYS
IIH
Input high threshold
VCC=3.3V
1.85
1.52
0.33
90
2.31
V
V
Input low threshold
VCC=3.3V
0.99
Input threshold hysteresis
Input high level input leakage current
Input low level input leakage
Input pins pull down resistance
Input pins pull up resistance
VCC=3.3V
V
VIN = VCC
µA
µA
IIL
VIN = GND
–90
55
RIND
RINU
see Section 8 for more information
see Section 8 for more information
kΩ
55
IN+, IN– and RST/EN deglitch (ON and
OFF) filter time
TINFIL
fS = 50kHz
28
40
50
ns
ns
TRSTFIL
Deglitch filter time to reset /FLT
400
650
800
GATE DRIVER STAGE
IOUT, IOUTH Peak source current
IOUT, IOUTL
10
10
A
A
CL=0.18µF, fS=1kHz
Peak sink current
(3)
ROUTH
Output pull-up resistance
Output pull-down resistance
High level output voltage
Low level output voltage
IOUT = –0.1A
2.5
0.3
17.5
60
Ω
ROUTL
VOUTH
VOUTL
IOUT = 0.1A
Ω
IOUT = –0.2A, VDD=18V
IOUT = 0.2A
V
mV
ACTIVE PULLDOWN
IOUTL or IOUT = 0.1×IOUT(L)(tpy)
VDD=OPEN, VEE=COM
,
VOUTPD
Output active pull down on OUTL
1.5
2.0
2.5
V
EXTERNAL MILLER CLAMP
VCLMPTH
VCLMPE
ICLMPEH
Miller clamp threshold voltage
Reference to VEE
Reference to VEE
1.5
4.8
2.0
5
2.5
5.3
V
V
A
Output high voltage
Peak source current
CCLMPE = 10nF ; guaranteed by design
0.25
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VCC=3.3V or 5.0V, 1uF capacitor from VCC to GND, VDD–COM=20V, 18V or 15V, COM–VEE =0V, 5V, 8V or
15V, CL=100pF, –40°C<TJ<150°C (unless otherwise noted)(1) (2)
.
PARAMETER
Peak sink current
Rising time
TEST CONDITIONS
MIN
TYP
0.25
20
MAX
0.37
40
UNIT
ICLMPEL
tCLMPER
tDCLMPE
CCLMPE = 10nF
0.12
A
ns
CCLMPE = 330pF
Miller clamp ON delay time
40
70
ns
SHORT CIRCUIT CLAMPING
VCLP-OUT(H)
VCLP-OUT(L)
OC PROTECTION
IDCHG
VOUT–VDD, VOUTH–VDD
OUT = Low, IOUT(H) = 500mA, tCLP=10us
OUT = High, IOUT(L) = 500mA, tCLP=10us
0.9
1.8
0.99
1.98
V
V
VOUT–VDD, VOUTL–VDD
OC pull down current when
Detection Threshold
VOC = 1V
40
mA
V
VOCTH
0.63
0.7
0.77
Voltage when OUT(L) = LOW, Reference
to COM
VOCL
IOC = 5mA
0.13
V
tOCFIL
tOCOFF
tOCFLT
OC fault deglitch filter
95
150
300
120
270
530
180
400
750
ns
ns
ns
OC propagation delay to OUT(L) 90%
OC to FLT low delay
2-LEVEL TURNOFF (Triggered by OC)
V2LOFF
t2LOFF
ITL1
2LOFF voltage threshold
8.3
9.0
700
900
900
10.0
V
2LOFF voltage duration
500
1000
ns
High to 2-Level transition sink current
Soft turn-off current on fault conditions
mA
mA
ITL3
500
1200
ISOLATED TEMPERATURE SENSE AND MONITOR (AIN–APWM)
VAIN
Analog sensing voltage range
Internal current source
0.5
196
360
4.5
209
440
V
IAIN
VAIN=2.5V, -40°C< TJ< 150°C
VAIN=2.5V
203
400
10
µA
fAPWM
BWAIN
APWM output frequency
AIN–APWM bandwidth
kHz
kHz
VAIN = 0.6V
VAIN = 2.5V
VAIN = 4.5V
86.5
48.5
7.5
88
89.5
51.5
11.5
DAPWM
APWM Dutycycle
50
%
10
FLT AND RDY REPORTING
VDD UVLO RDY low minimum holding
time
tRDYHLD
0.55
0.55
1
1
ms
tFLTMUTE
RODON
VODL
Output mute time on fault
Open drain output on resistance
Open drain low output voltage
Reset fault through RST/EN
IODON = 5mA
ms
Ω
30
IODON = 5mA
0.3
V
COMMON MODE TRANSIENT IMMUNITY
CMTI Common-mode transient immunity
VCM = 1500 V
150
V/ns
(1) Current are positive into and negative out of the specified terminal.
(2) All voltages are referenced to COM unless otherwise notified.
(3) For internal PMOS only. Refer to Section 8.3 for effective pull-up resistance.
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6.10 Switching Characteristics
VCC=5.0V, 1uF capacitor from VCC to GND, VDD–COM=20V, 18V or 15V, COM–VEE = 3V, 5V or 8V,
CL=100pF, –40°C<TJ<150°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
130
130
30
UNIT
ns
tPDHL
tPDLH
PWD
tsk-pp
tr
Propagation delay time – High to Low
Propagation delay time – Low to High
60
90
60
90
Pulse width distortion |tPDHL – tPDLH
|
Part to Part skew
Rising or Falling Propagation Delay
30
Driver output rise time
CL=10nF
CL=10nF
28
tf
Driver output fall time
24
fMAX
Maximum switching frequency
1
MHz
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6.11 Insulation Characteristics Curves
1.E+12
1.E+11
1.E+10
54 Yrs
1.E+09
1.E+08
1.E+07
1.E+06
1.E+05
1.E+04
1.E+03
1.E+02
1.E+01
TDDB Line (< 1 ppm Fail Rate)
VDE Safety Margin Zone
1800VRMS
2200
200
1200
3200
4200
5200
6200
Applied Voltage (VRMS
)
Figure 6-1. Reinforced Isolation Capacitor Life Time Projection
100
80
60
40
20
0
2000
VDD=15V; VEE=-5V
VDD=20V; VEE=-5V
1500
1000
500
0
0
25
50
75
100
125
150
0
25
50
75
100
125
150
Ambient Temperature (oC)
Ambient Temperature (oC)
Safe
Safe
Figure 6-3. Thermal Derating Curve for Limiting
Power per VDE
Figure 6-2. Thermal Derating Curve for Limiting
Current per VDE
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6.12 Typical Characteristics
22
20
18
16
14
12
10
8
22
20
18
16
14
12
10
8
VDD/VEE = 18V/0V
VDD/VEE = 20V/-5V
VDD/VEE = 18V/0V
VDD/VEE = 20V/-5V
6
6
4
4
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D016
D017
Figure 6-4. Output High Drive Current vs.
Temperature
Figure 6-5. Output Low Driver Current vs.
Temperature
6
4
VCC = 3.3V
VCC = 5V
VCC = 3.3V
VCC = 5V
5.5
3.5
5
4.5
4
3
2.5
2
3.5
3
1.5
1
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D015
D014
A.
A.
IN+ = High
IN- = Low
IN+ = Low
IN- = Low
Figure 6-6. IVCCQ Supply Current vs. Temperature
Figure 6-7. IVCCQ Supply Current vs. Temperature
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5
4.5
4
6
5.5
5
VDD/VEE = 18V/0V
VDD/VEE = 20V/-5V
VDD/VEE = 18V/0V
VDD/VEE = 20V/-5V
3.5
3
4.5
4
2.5
2
3.5
3
30
70
110
150 190
Frequency (kHz)
230
270
310
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D018
D012
A.
Figure 6-8. IVCCQ Supply Current vs. Input
Frequency
IN+ = High
IN- = Low
Figure 6-9. IVDDQ Supply Current vs. Temperature
6
10
VDD/VEE = 18V/0V
VDD/VEE = 20V/-5V
VDD/VEE = 18V/0V
VDD/VEE = 20V/-5V
9
8
7
6
5
4
3
2
5.5
5
4.5
4
3.5
3
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
30
70
110
150
190
Frequency (kHz)
230
270
310
D013
D019
A.
Figure 6-11. IVDDQ Supply Current vs. Input
Frequency
IN+ = Low
IN- = Low
Figure 6-10. IVDDQ Supply Current vs. Temperature
4
3.5
3
14
13.5
13
12.5
12
2.5
2
11.5
11
10.5
10
1.5
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D002
D001
Figure 6-13. VDD UVLO vs. Temperature
Figure 6-12. VCC UVLO vs. Temperature
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100
100
90
80
70
60
50
90
80
70
60
50
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D022
D021
A.
A.
VCC = 3.3V
RON = 0Ω
VDD=18V
CL = 100pF
VCC = 3.3V
RON = 0Ω
VDD=18V
CL = 100pF
ROFF = 0Ω
ROFF = 0Ω
Figure 6-15. Propagation Delay tPDHL vs.
Temperature
Figure 6-14. Propagation Delay tPDLH vs.
Temperature
60
60
50
40
30
20
10
50
40
30
20
10
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D023
D024
A.
A.
VCC = 3.3V
RON = 0Ω
VDD=18V
CL = 10nF
VCC = 3.3V
RON = 0Ω
VDD=18V
CL = 10nF
ROFF = 0Ω
ROFF = 0Ω
Figure 6-16. tr Rise Time vs. Temperature
Figure 6-17. tf Fall Time vs. Temperature
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2.5
2.25
2
3
2.75
2.5
1.75
1.5
1.25
1
2.25
2
1.75
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D025
1.5
Figure 6-19.
VCLP-OUT(H) Short Circuit Clamping Voltage vs. Temperature
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D008
Figure 6-18. VOUTPD Output Active Pulldown
Voltage vs. Temperature
2
1.75
1.5
3
2.75
2.5
1.25
1
2.25
2
0.75
0.5
0.25
-60 -40 -20
1.75
1.5
0
20 40 60 80 100 120 140 160
Temperature (èC)
D026
Figure 6-20.
50
70
90
110
130
150 160
VCLP-OUT(L) Short Circuit Clamping Voltage vs. Temperature
Temperature (èC)
D009
Figure 6-21. VCLMPTH Miller Clamp Threshold
Voltage vs. Temperature
400
350
300
250
200
150
100
70
60
50
40
30
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D010
D011
Figure 6-22. ICLMPEL Miller Clamp Sink Current vs.
Temperature
Figure 6-23. tDCLMPE Miller Clamp ON Delay Time
vs. Temperature
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330
320
310
300
290
280
270
260
250
240
230
1
0.8
0.6
0.4
0.2
VCC = 3.3V
VCC = 5V
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D020
D003
Figure 6-24. tOCOFF OC Propagation Delay vs.
Temperature
Figure 6-25. VOCTH OC Detection Threshold vs.
Temperature
700
650
600
550
500
450
400
10
9.5
9
8.5
8
7.5
7
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D004
D005
Figure 6-26. tOCFLT OC to FLT Low Delay Time vs.
Temperature
Figure 6-27. V2LOFF 2-Level Turn Off Voltage
Threshold vs. Temperature
900
800
700
600
500
-60 -40 -20
0
20 40 60 80 100 120 140 160
Temperature (èC)
D006
Figure 6-28. t2LOFF 2-Level Turn Off Time vs. Temperature
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7 Parameter Measurement Information
7.1 Propagation Delay
7.1.1 Regular Turn-OFF
Figure 7-1 shows the propagation delay measurement for non-inverting configurations. Figure 7-2 shows the
propagation delay measurement with the inverting configurations.
50%
50%
IN+
INÅ
tPDLH
tPDHL
90%
10%
OUT
Figure 7-1. Non-inverting Logic Propagation Delay Measurement
IN+
INÅ
50%
50%
tPDLH
tPDHL
90%
OUT
10%
Figure 7-2. Inverting Logic Propagation Delay Measurement
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7.2 Input Deglitch Filter
In order to increase the robustness of gate driver over noise transient and accidental small pulses on the input
pins, i.e. IN+, IN–, RST/EN, a 40ns deglitch filter is designed to filter out the transients and make sure there is no
faulty output responses or accidental driver malfunctions. When the IN+ or IN– PWM pulse is smaller than the
input deglitch filter width, TINFIL, there will be no responses on OUT drive signal. Figure 7-3 and Figure 7-4
shows the IN+ pin ON and OFF pulse deglitch filter effect. Figure 7-5 and Figure 7-6 shows the IN– pin ON and
OFF pulse deglitch filter effect.
IN+
tPWM < TINFIL
tPWM < TINFIL
IN+
INÅ
INÅ
OUT
OUT
Figure 7-3. IN+ ON Deglitch Filter
Figure 7-4. IN+ OFF Deglitch Filter
IN+
IN+
INÅ
tPWM < TINFIL
tPWM < TINFIL
INÅ
OUT
OUT
Figure 7-5. IN– ON Deglitch Filter
Figure 7-6. IN– OFF Deglitch Filter
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7.3 Active Miller Clamp
7.3.1 External Active Miller Clamp
For gate driver application with unipolar bias supply or bipolar supply with small negative turn-off voltage, active
miller clamp can help add an additional low impedance path to bypass the miller current and prevent the high
dV/dt introduced unintentional turn-on through the miller capacitance. Different from the internal active miller
clamp, external active miller clamp function is used for applications where the gate driver may not be close to the
power device or power module due to system layout considerations. External active miller clamp function provide
a 5V gate drive signal to turn-on the external miller clamp FET when the gate driver voltage is less than miller
clamp threshold, VCLMPTH. Figure 7-7 shows the timing diagram for external active miller clamp function.
(”IN+‘ Å ”INÅ‘)
IN
VDD
tDCLMPE
OUT
VCLMPTH
COM
VEE
HIGH
90%
tCLMPER
LOW
10%
CLMPE
Figure 7-7. Timing Diagram for External Active Miller Clamp Function
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7.4 Under Voltage Lockout (UVLO)
UVLO is one of the key protection features designed to protect the system in case of bias supply failures on VCC
— primary side power supply, and VDD — secondary side power supply.
7.4.1 VCC UVLO
The VCC UVLO protection details are discussed in this section. Figure 7-8 shows the timing diagram illustrating
the definition of UVLO ON/OFF threshold, deglitch filter, response time, RDY and AIN–APWM.
IN
(”IN+‘ Å ”INÅ‘)
tVCCFIL
tVCCÅ to OUT
VVCC_ON
VCC
VVCC_OFF
VDD
COM
VEE
tVCC+ to OUT
90%
VCLMPTH
OUT
10%
tVCC+ to RDY
tRDYHLD
tVCCÅ to RDY
Hi-Z
RDY
VCC
APWM
Figure 7-8. VCC UVLO Protection Timing Diagram
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7.4.2 VDD UVLO
The VDD UVLO protection details are discussed in this section. Figure 7-9 shows the timing diagram illustrating
the definition of UVLO ON/OFF threshold, deglitch filter, response time, RDY and AIN–APWM.
IN
(”IN+‘ Å ”INÅ‘)
tVDDFIL
VDD
tVDDÅ to OUT
VVDD_ON
VVDD_OFF
COM
VEE
VCC
tVDD+ to OUT
VCLMPTH
OUT
90%
tRDYHLD
10%
tVDD+ to RDY
tVDDÅ to RDY
RDY
Hi-Z
VCC
APWM
Figure 7-9. VDD UVLO Protection Timing Diagram
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7.5 OC (Over Current) Protection
7.5.1 OC Protection with 2-Level Turn-OFF
OC Protection is used to sense the current of SiC-MOSFETs and IGBTs under over current or shoot-through
condition. Figure 7-10 shows the timing diagram of OC operation with 2-level turn-off.
IN
(”IN+‘ Å ”INÅ‘)
tOCFIL
VOCTH
OC
tOCOFF
90%
V2LOFF
t2LOFF
GATE
VCLMPTH
tOCFLT
tFLTMUTE
Hi-Z
FLT
tRSTFIL
tRSTFIL
RST/EN
HIGH
Hi-Z
OUTH
OUTL
LOW
Hi-Z
LOW
Figure 7-10. OC Protection with 2-Level Turn-OFF
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8 Detailed Description
8.1 Overview
The UCC21732 device is an advanced isolated gate driver with state-of-art protection and sensing features for
SiC MOSFETs and IGBTs. The device can support up to 2121V DC operating voltage based on SiC MOSFETs
and IGBTs, and can be used to above 10kW applications such as HEV/EV traction inverter, motor drive, on-
board and off-board battery charger, solar inverter, etc. The galvanic isolation is implemented by the capacitive
isolation technology, which can realize a reliable reinforced isolation between the low voltage DSP/MCU and
high voltage side.
The ±10A peak sink and source current of UCC21732 can drive the SiC MOSFET modules and IGBT modules
directly without an extra buffer. The driver can also be used to drive higher power modules or parallel modules
with external buffer stage. The input side is isolated with the output side with a reinforced isolation barrier based
on capacitive isolation technology. The device can support up to 1.5-kVRMS working voltage, 12.8-kVPK surge
immunity with longer than 40 years isolation barrier life. The strong drive strength helps to switch the device fast
and reduce the switching loss. While the 150V/ns minimum CMTI guarantees the reliability of the system with
fast switching speed. The small propagation delay and part-to-part skew can minimize the deadtime setting, so
the conduction loss can be reduced.
The device includes extensive protection and monitor features to increase the reliability and robustness of the
SiC MOSFET and IGBT based systems. The 12V output side power supply UVLO is suitable for switches with
gate voltage ≥ 15V. The active miller clamp feature prevents the false turn on causing by miller capacitance
during fast switching. External miller clamp FET can be used, providing more versatility to the system design.
The device has the state-of-art overcurrent and short circuit detection time, and fault reporting function to the low
voltage side DSP/MCU. The 2-level turn-off with soft turn off is triggered when the overcurrent or short circuit
fault is detected, minimizing the short circuit energy while reducing the overshoot voltage on the switches.
The isolated analog to PWM sensor can be used as switch temperature sensing, DC bus voltage sensing,
auxiliary power supply sensing, etc. The PWM signal can be fed directly to DSP/MCU or through a low-pass-filter
as an analog signal.
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8.2 Functional Block Diagram
CLMPE
OUTH
7
4
6
10
11
15
9
IN+
INt
55kQ
55kQ
PWM Inputs
MOD
DEMOD
Output Stage
t
ON/OFF Control
STO
VCC
OUTL
VDD
VCC
UVLO
VCC Supply
5
GND
RDY
UVLO
LDO[s for VEE,
COM and channel
3
8
COM
VEE
OC
12
13
14
16
Fault Decode
FLT
OCP
2
Fault Encode
RST/EN
50kQ
Analog 2 PWM
PWM Driver
AIN
1
APWM
DEMOD
MOD
8.3 Feature Description
8.3.1 Power Supply
The input side power supply VCC can support a wide voltage range from 3V to 5.5V. The device supports both
unipolar and bipolar power supply on the output side, with a wide range from 13V to 33V from VDD to VEE. The
negative power supply with respect to switch source or emitter is usually adopted to avoid false turn on when the
other switch in the phase leg is turned on. The negative voltage is especially important for SiC MOSFET due to
its fast switching speed.
8.3.2 Driver Stage
UCC21732 has ±10A peak drive strength and is suitable for high power applications. The high drive strength can
drive a SiC MOSFET module, IGBT module or paralleled discrete devices directly without extra buffer stage.
UCC21732 can also be used to drive higher power modules or parallel modules with extra buffer stage.
Regardless of the values of VDD, the peak sink and source current can be kept at 10A. The driver features an
important safety function wherein, when the input pins are in floating condition, the OUTH/OUTL is held in LOW
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state. The split output of the driver stage is depicted in . The driver has rail-to-rail output by implementing a
hybrid pull-up structure with a P-Channel MOSFET in parallel with an N-Channel MOSFET, and an N-Channel
MOSFET to pulldown. The pull-up NMOS is the same as the pull down NMOS, so the on resistance RNMOS is
the same as ROL. The hybrid pull-up structure delivers the highest peak-source current when it is most needed,
during the miller plateau region of the power semiconductor turn-on transient. The ROH in represents the on-
resistance of the pull-up P-Channel MOSFET. However, the effective pull-up resistance is much smaller than
ROH. Since the pull-up N-Channel MOSFET has much smaller on-resistance than the P-Channel MOSFET, the
pull-up N-Channel MOSFET dominates most of the turn-on transient, until the voltage on OUTH pin is about 3V
below VDD voltage. The effective resistance of the hybrid pull-up structure during this period is about 2 x ROL
Then the P-Channel MOSFET pulls up the OUTH voltage to VDD rail. The low pull-up impedance results in
.
strong drive strength during the turn-on transient, which shortens the charging time of the input capacitance of
the power semiconductor and reduces the turn on switching loss.
The pull-down structure of the driver stage is implemented solely by a pull-down N-Channel MOSFET. The on-
resistance of the N-Channel MOSFET ROL can be found in the . This MOSFET can ensure the OUTL voltage be
pulled down to VEE rail. The low pull-down impedance not only results in high sink current to reduce the turn-off
time, but also helps to increase the noise immunity considering the miller effect.
VDD
ROH
RNMOS
OUTH
Input
Signal
Anti Shoot-
through
Circuitry
OUTL
ROL
Figure 8-1. Gate Driver Output Stage
8.3.3 VCC and VDD Undervoltage Lockout (UVLO)
UCC21732 implements the internal UVLO protection feature for both input and output power supplies VCC and
VDD. When the supply voltage is lower than the threshold voltage, the driver output is held as LOW. The output
only goes HIGH when both VCC and VDD are out of the UVLO status. The UVLO protection feature not only
reduces the power consumption of the driver itself during low power supply voltage condition, but also increases
the efficiency of the power stage. For SiC MOSFET and IGBT, the on-resistance reduces while the gate-source
voltage or gate-emitter voltage increases. If the power semiconductor is turned on with a low VDD value, the
conduction loss increases significantly and can lead to a thermal issue and efficiency reduction of the power
stage. UCC21732 implements 12V threshold voltage of VDD UVLO, with 800mV hysteresis. This threshold
voltage is suitable for both SiC MOSFET and IGBT.
The UVLO protection block features with hysteresis and deglitch filter, which help to improve the noise immunity
of the power supply. During the turn on and turn off switching transient, the driver sources and sinks a peak
transient current from the power supply, which can result in sudden voltage drop of the power supply. With
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hysteresis and UVLO deglitch filter, the internal UVLO protection block will ignore small noises during the normal
switching transients.
The timing diagrams of the UVLO feature of VCC and VDD are shown in Figure 7-8, and Figure 7-9. The RDY
pin on the input side is used to indicate the power good condition. The RDY pin is open drain. During UVLO
condition, the RDY pin is held in low status and connected to GND. Normally the pin is pulled up externally to
VCC to indicate the power good. The AIN-APWM function stops working during the UVLO status. The APWM
pin on the input side will be held LOW.
8.3.4 Active Pulldown
UCC21732 implements an active pulldown feature to ensure the OUTH/OUTL pin clamping to VEE when the
VDD is open. The OUTH/OUTL pin is in high-impedance status when VDD is open, the active pulldown feature
can prevent the output be false turned on before the device is back to control.
VDD
OUTL
Ra
Control
Circuit
VEE
COM
Figure 8-2. Active Pulldown
8.3.5 Short Circuit Clamping
During short circuit condition, the miller capacitance can cause a current sinking to the OUTH/OUTL pin due to
the high dV/dt and boost the OUTH/OUTL voltage. The short circuit clamping feature of UCC21732 can clamp
the OUTH/OUTL pin voltage to be slightly higher than VDD, which can protect the power semiconductors from a
gate-source and gate-emitter overvoltage breakdown. This feature is realized by an internal diode from the
OUTH/OUTL to VDD.
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VDD
D1 D2
OUTH
OUTL
Control
Circuitry
Figure 8-3. Short Circuit Clamping
8.3.6 External Active Miller Clamp
Active miller clamp feature is important to prevent the false turn-on while the driver is in OFF state. In
applications which the device can be in synchronous rectifier mode, the body diode conducts the current during
the deadtime while the device is in OFF state, the drain-source or collector-emitter voltage remains the same
and the dV/dt happens when the other power semiconductor of the phase leg turns on. The low internal pull-
down impedance of UCC21732 can provide a strong pulldown to hold the OUTL to VEE. However, external gate
resistance is usually adopted to limit the dV/dt. The miller effect during the turn on transient of the other power
semiconductor can cause a voltage drop on the external gate resistor, which boost the gate-source or gate-
emitter voltage. If the voltage on VGS or VGE is higher than the threshold voltage of the power semiconductor, a
shoot through can happen and cause catastrophic damage. The active miller clamp feature of UCC21732 drives
an external MOSFET, which connects to the device gate. The external MOSFET is triggered when the gate
voltage is lower than VCLMPTH, which is 2V above VEE, and creates a low impedance path to avoid the false turn
on issue.
VCLMPTH
VCC
OUTH
+
3V to 5.5V
IN+
œ
CLMPE
OUTL
Control
Circuitry
µC
MOD
DEMOD
IN-
VEE
COM
VCC
Figure 8-4. Active Miller Clamp
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8.3.7 Overcurrent and Short Circuit Protection
The UCC21732 implements a fast overcurrent and short circuit protection feature to protect the SiC MOSFET or
IGBT from catastrophic breakdown during fault. The OC pin of the device has a typical 0.7V threshold with
respect to COM, source or emitter of the power semiconductor. When the input is in floating condition, or the
output is held in low state, the OC pin is pulled down by an internal MOSFET and held in LOW state, which
prevents the overcurrent and short circuit fault from false triggering. The OC pin is in high-impedance state when
the output is in high state, which means the overcurrent and short circuit protection feature only works when the
power semiconductor is in on state. The internal pulldown MOSFET helps to discharge the voltage of OC pin
when the power semiconductor is turned off.
The overcurrent and short circuit protection feature can be used to SiC MOSFET module or IGBT module with
SenseFET, traditional desaturation circuit and shunt resistor in series with the power loop for lower power
applications. For SiC MOSFET module or IGBT module with SenseFET, the SenseFET integrated in the module
can scale down the drain current or collector current. With an external high precision sense resistor, the drain
current or collector current can be accurately measured. If the voltage of the sensed resistor higher than the
overcurrent threshold VOCTH is detected, the 2-Level turn-off is initiated. A fault will be reported to the input side
FLT pin to DSP/MCU. The output is held to LOW after the fault is detected, and can only be reset by the RST/EN
pin. The state-of-art overcurrent and short circuit detection time helps to ensure a short shutdown time for SiC
MOSFET and IGBT.
The overcurrent and short circuit protection feature can also be paired with desaturation circuit and shunt
resistors. The DESAT threshold can be programmable in this case, which increases the versatility of the device.
Detailed application diagrams of desaturation circuit and shunt resistor will be given in Overcurrent and Short
Circuit Protection.
•
High current and high dI/dt during the overcurrent and short circuit fault can cause a voltage bounce on shunt
resistor’s parasitic inductance and board layout parasitic, which results in false trigger of OC pin. High
precision, low ESL and small value resistor must be used in this approach.
•
Shunt resistor approach is not recommended for high power applications and short circuit protection of the
low power applications.
The detailed applications of the overcurrent and short circuit feature will be discussed in the Application and
Implementation section.
OUTL
ROFF
OC
RFLT
+
FLT
DEMOD
MOD
+
RS
CFLT
VOCTH
œ
Control
Logic
GND
COM
VEE
Figure 8-5. Overcurrent and Short Circuit Protection
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8.3.8 2-Level Turn-off
UCC21732 initiates a fast 2-level turn-off when the overcurrent and short circuit protection is triggered. When the
overcurrent and short circuit fault happens, the power power semiconductor transits from the linear region to the
saturation region very fast. The channel current is controlled by the gate voltage. By pulling down the gate
voltage to a mid-voltage level V2LOFF and stay for a fixed time t2LOFF, the channel current can be limited to a
much lower level, which significantly reduces the energy dissipation during the fault event. After t2LOFF, the
driver continues to pull down the gate voltage by the soft turn off current ITL3 until it reaches VEE. With dI/dt of
the channel current is controlled by the gate voltage and decreasing in a soft manner, thus the overshoot of the
power semiconductor is limited and prevents the overvoltage breakdown. The timing diagram of 2-level turn-off
shows in Figure 7-10.
OUTL
2-Level
Turn Off
ROFF
OC
RFLT
+
FLT
DEMOD
MOD
+
VOCTH
RS
CFLT
œ
Control
Logic
GND
COM
VEE
Figure 8-6. 2-Level Turn-off
8.3.9 Fault ( FLT, Reset and Enable ( RST/EN)
The FLT pin of UCC21732 is open drain and can report a fault signal to the DSP/MCU when the overcurrent and
short circuit fault is detected through OC pin. The FLT pin is pulled down to GND, and is held in low state unless
a reset signal is received from RST/EN. The device has a fault mute time tFLTMUTE, within which the device
ignores any reset signal.
The RST/EN is pulled down internally. The device is disabled by default if the RST/EN pin is floating. The pin has
two purposes:
•
Resets the overcurrent and short circuit fault signaled on FLT pin. The RST/EN pin is active low, if the pin is
set and held in low state for more than tRSTFIL, the fault signal is reset and FLT is reset back to the high
impedance status at the rising edge of RST/EN pin.
•
Enable and shutdown the device. If the RST/EN pin is pulled low, the driver is disabled and shut down by the
regular turn off. The pin must be pulled up externally to enable the part, otherwise the device is disabled by
default.
8.3.10 Isolated Analog to PWM Signal Function
The UCC21732 features an isolated analog to PWM signal function from AIN to APWM pin, which allows the
isolated temperature sensing, high voltage dc bus voltage sensing, etc. An internal current source IAIN in AIN pin
is implemented in the device to bias an external thermal diode or temperature sensing resistor. The UCC21732
encodes the voltage signal VAIN to a PWM signal, passing through the reinforced isolation barrier, and output to
APWM pin on the input side. The PWM signal can either be transferred directly to DSP/MCU to calculate the
duty cycle, or filtered by a simple RC filter as an analog signal. The AIN voltage input range is from 0.6V to 4.5V,
and the corresponding duty cycle of the APWM output ranges from 88% to 10%. The duty cycle increases
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linearly from 10% to 88% while the AIN voltage decreases from 4.5V to 0.6V. This corresponds to the
temperature coefficient of the negative temperature coefficient (NTC) resistor and thermal diode. When AIN is
floating, the AIN voltage is 5V and the APWM operates at 400kHz with approximately 10% duty cycle. The
accuracy of the duty cycle is ±5% across temperature without one time calibration. The accuracy can be
improved to ±2% with calibration. The accuracy of the internal current source IAIN is 3% across temperature.
The isolated analog to PWM signal feature can also support other analog signal sensing, such as the high
voltage dc bus voltage, etc. The internal current source IAIN should be taken into account when designing the
potential divider if sensing a high voltage.
UCC21732-Q1
In Module or
Discrete
VCC
VDD
13V to
33V
+
+
3V to 5.5V
APWM
œ
œ
AIN
+
DEMOD
MOD
µC
Rfilt
Cfilt
OSC
GND
COM
Thermal
Diode
NTC or
PTC
Figure 8-7. Isolated Analog to PWM Signal
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8.4 Device Functional Modes
Table 8-1 lists the device function.
Table 8-1. Function Table
Input
Output
OUTH/
OUTL
VCC
VDD
VEE
IN+
IN-
RST/EN
AIN
RDY
FLT
CLMPE
APWM
PU
PD
PU
PU
PU
PU
PU
PU
PD
PU
PU
PU
X
X
X
X
X
X
X
X
X
X
X
X
X
X
Low
HiZ
HiZ
Low
Low
HiZ
HiZ
HiZ
HiZ
HiZ
HiZ
HiZ
HiZ
HiZ
HiZ
HiZ
Low
Low
Low
HiZ
Low
High
High
HiZ
Low
Low
Low
HiZ
Low
P*
PU
PU
X
X
Low
X
Open
PU
PU
X
X
Open
PU
X
X
X
Low
Low
Low
Low
High
High
High
High
PU
Low
X
X
High
High
High
PU
PU
High
High
P*
PU
PU
High
P*
PU: Power Up (VCC ≥ 2.85V, VDD ≥ 13.1V, VEE ≤ 0V); PD: Power Down (VCC ≤ 2.35V, VDD ≤ 9.9V); X:
Irrelevant; P*: PWM Pulse; HiZ: High Impedance
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9 Applications and Implementation
Note
Information in the following applications sections is not part of the TI component specification, and TI
does not warrant its accuracy or completeness. TI’s customers are responsible for determining
suitability of components for their purposes. Customers should validate and test their design
implementation to confirm system functionality.
9.1 Application Information
The UCC21732 device is very versatile because of the strong drive strength, wide range of output power supply,
high isolation ratings, high CMTI and superior protection and sensing features. The 1.5-kVRMS working voltage
and 12.8-kVPK surge immunity can support up both SiC MOSFET and IGBT modules with DC bus voltage up to
2121V. The device can be used in both low power and high power applications such as the traction inverter in
HEV/EV, on-board charger and charging pile, motor driver, solar inverter, industrial power supplies and etc. The
device can drive the high power SiC MOSFET module, IGBT module or paralleled discrete device directly
without external buffer drive circuit based on NPN/PNP bipolar transistor in totem-pole structure, which allows
the driver to have more control to the power semiconductor and saves the cost and space of the board design.
UCC21732 can also be used to drive very high power modules or paralleled modules with external buffer stage.
The input side can support power supply and microcontroller signal from 3.3V to 5V, and the device level shifts
the signal to output side through reinforced isolation barrier. The device has wide output power supply range
from 13V to 33V and support wide range of negative power supply. This allows the driver to be used in SiC
MOSFET applications, IGBT application and many others. The 12V UVLO benefits the power semiconductor
with lower conduction loss and improves the system efficiency. As a reinforced isolated single channel driver, the
device can be used to drive either a low-side or high-side driver.
UCC21732 device features extensive protection and monitoring features, which can monitor, report and protect
the system from various fault conditions.
•
Fast detection and protection for the overcurrent and short circuit fault. The feature is preferable in a split
source SiC MOSFET module or a split emitter IGBT module. For the modules with no integrated current
mirror or paralleled discrete semiconductors, the traditional desaturation circuit can be modified to implement
short circuit protection. The semiconductor is shutdown when the fault is detected and FLTb pin is pulled
down to indicate the fault detection. The device is latched unless reset signal is received from the RST/EN
pin.
•
•
2-level turn-off feature to protect the power semiconductor from catastrophic breakdown during overcurrent
and short circuit fault. The shutdown energy can be controlled while the overshoot of the power
semiconductor is limited.
UVLO detection to protect the semiconductor from excessive conduction loss. Once the device is detected to
be in UVLO mode, the output is pulled down and RDY pin indicates the power supply is lost. The device is
back to normal operation mode once the power supply is out of the UVLO status. The power good status can
be monitored from the RDY pin.
•
•
Analog signal seensing with isolated analog to PWM signal feature. This feature allows the device to sense
the temperature of the semiconductor from the thermal diode or temperature sensing resistor, or dc bus
voltage with resistor divider. A PWM signal is generated on the low voltage side with reinforced isolated from
the high voltage side. The signal can be fed back to the microcontroller for the temperature monitoring,
voltage monitoring and etc.
The active miller clamp feature protects the power semiconductor from false turn on by driving an external
MOSFET. This feature allows the flexibility of the board layout design and the pulldown strength of miller
clamp FET.
•
•
•
Enable and disable function through the RSTb/EN pin.
Short circuit clamping.
Active pulldown.
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9.2 Typical Application
Figure 9-1 shows the typical application of a half bridge using two UCC21732 isolated gate drivers. The half
bridge is a basic element in various power electronics applications such as traction inverter in HEV/EV to convert
the DC current of the electric vehicle’s battery to the AC current to drive the electric motor in the propulsion
system. The topology can also be used in motor drive applications to control the operating speed and torque of
the AC motors.
UCC
UCC
UCC
UCC
21732
21732
21732
21732
1
2
3
4
5
6
PWM
3-Phase
Input
1
2
3
4
5
6
µC
M
APWM
FLT
UCC
UCC
21732
21732
Figure 9-1. Typical Application Schematic
9.2.1 Design Requirements
The design of the power system for end equipment should consider some design requirements to ensure the
reliable operation of UCC1732 through the load range. The design considerations include the peak source and
sink current, power dissipation, overcurrent and short circuit protection, AIN-APWM function for analog signal
sensing and etc.
A design example for a half bridge based on IGBT is given in this subsection. The design parameters are show
in Table 9-1.
Table 9-1. Design Parameters
Parameter
Input Supply Voltage
IN-OUT Configuration
Positive Output Voltage VDD
Negative Output Voltage VEE
DC Bus Voltage
Value
5V
Non-inverting
15V
-5V
800V
Peak Drain Current
Switching Frequency
Switch Type
300A
50kHz
IGBT Module
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9.2.2 Detailed Design Procedure
9.2.2.1 Input filters for IN+, IN- and RST/EN
In the applications of traction inverter or motor drive, the power semiconductors are in hard switching mode. With
the strong drive strength of UCC21732, the dV/dt can be high, especially for SiC MOSFET. Noise can not only
be coupled to the gate voltage due to the parasitic inductance, but also to the input side as the non-ideal PCB
layout and coupled capacitance.
UCC21732 features a 40ns internal deglitch filter to IN+, IN- and RST/EN pin. Any signal less than 40ns can be
filtered out from the input pins. For noisy systems, external low pass filter can be added externally to the input
pins. Adding low pass filters to IN+, IN- and RST/EN pins can effectively increase the noise immunity and
increase the signal integrity. When not in use, the IN+, IN- and RST/EN pins should not be floating. IN- should be
tied to GND if only IN+ is used for non-inverting input to output configuration. The purpose of the low pass filter is
to filter out the high frequency noise generated by the layout parasitics. While choosing the low pass filter
resistors and capacitors, both the noise immunity effect and delay time should be considered according to the
system requirements.
9.2.2.2 PWM Interlock of IN+ and IN-
UCC21732 features the PWM interlock for IN+ and IN- pins, which can be used to prevent the phase leg shoot
through issue. As shown in Table 8-1, the output is logic low while both IN+ and IN- are logic high. When only IN
+ is used, IN- can be tied to GND. To utilize the PWM interlock function, the PWM signal of the other switch in
the phase leg can be sent to the IN- pin. As shown in Figure 9-2, the PWM_T is the PWM signal to top side
switch, the PWM_B is the PWM signal to bottom side switch. For the top side gate driver, the PWM_T signal is
given to the IN+ pin, while the PWM_B signal is given to the IN- pin; for the bottom side gate driver, the PWM_B
signal is given to the IN+ pin, while PWM_T signal is given to the IN- pin. When both PWM_T and PWM_B
signals are high, the outputs of both gate drivers are logic low to prevent the shoot through condition.
IN+
IN-
RON
OUTH
OUTL
ROFF
PWM_T
PWM_B
RON
IN+
IN-
OUTH
OUTL
ROFF
Figure 9-2. PWM Interlock for a Half Bridge
9.2.2.3 FLT, RDY and RST/EN Pin Circuitry
Both FLT and RDY pin are open-drain output. The RST/EN pin has 50kΩ internal pulldown resistor, so the driver
is in OFF status if the RST/EN pin is not pulled up externally. A 5kΩ resistor can be used as pullup resistor for
the FLT, RDY and RST/EN pins.
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To improve the noise immunity due to the parasitic coupling and common mode noise, low pass filters can be
added between the FLT, RDY and RST/EN pins and the microcontroller. A filter capacitor between 100pF to
300pF can be added.
3.3V to 5V
VCC
15
1µF
0.1µF
GND
IN+
9
10
INt
11
5kQ
5kQ 5kQ
FLT
12
13
100pF
RDY
100pF
RST/EN
14
16
100pF
APWM
Figure 9-3. FLT, RDY and RST/EN Pins Circuitry
9.2.2.4 RST/EN Pin Control
RST/EN pin has two functions. It can be used to enable and shutdown the outputs of the driver, and reset the
fault signaled on the FLT pin. RST/EN pin needs to be pulled up to enable the device; when the pin is pulled
down, the device is in disabled status. With a 50kΩ pulldown resistor existing, the driver is disabled by default.
When the driver is latched after overcurrent or short circuit fault is detected, the FLT pin and output are latched
low and need to be reset by RST/EN pin. RST/EN pin is active low. The microcntroller needs to send a signal to
RST/EN pin after the fault mute time tFLTMUTE to reset the driver. This pin can also be used to automatically reset
the driver. The continuous input signal IN+ or IN- can be applied to RST/EN pin, so the microcontroller does not
need to generate another control signal to reset the driver. If non-inverting input IN+ is used, then IN+ can be
tied to RST/EN pin. If inverting input IN- is used, then a NOT logic is needed between the inverting PWM signal
from the microcontroller and the RST/EN pin. In this case, the driver can be reset in every switching cycle
without an extra control signal from microcontroller to RST/EN pin.
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3.3V to 5V
0.1µF
3.3V to 5V
0.1µF
VCC
VCC
15
15
1µF
1µF
GND
IN+
GND
9
9
IN+
10
10
INt
INt
5kQ
11
5kQ
5kQ
11
5kQ
FLT
FLT
12
13
12
13
100pF
100pF
100pF
100pF
RDY
RDY
RST/EN
APWM
RST/EN
14
14
16
APWM
16
Figure 9-4. Automatic Reset Control
9.2.2.5 Turn on and turn off gate resistors
UCC21732 features split outputs OUTH and OUTL, which enables the independent control of the turn on and
turn off switching speed. The turn on and turn off resistance determine the peak source and sink current, which
controls the switching speed in turn. Meanwhile, the power dissipation in the gate driver should be considered to
ensure the device is in the thermal limit. At first, the peak source and sink current are calculated as:
VDD - VEE
ROH_EFF +RON +RG _Int
Isource _ pk = min(10A,
)
VDD - VEE
ROL +ROFF +RG _Int
Isink _ pk = min(10A,
)
(1)
Where
•
ROH_EFF is the effective internal pull up resistance of the hybrid pull-up structure, which is approximately 2 x
ROL, about 0.7 Ω
•
•
•
•
ROL is the internal pulldown resistance, about 0.3 Ω
RON is the external turn on gate resistance
ROFF is the external turn off gate resistance
RG_Int is the internal resistance of the SiC MOSFET or IGBT module
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VDD
Cies=Cgc+Cge
+
Cgc
VDD
ROH_EFF
t
OUTH
OUTL
RON
RG_Int
ROFF
Cge
+
VEE
ROL
t
VEE
COM
Figure 9-5. Output Model for Calculating Peak Gate Current
For example, for an IGBT module based system with the following parameters:
•
•
•
Qg = 3300 nC
RG_Int = 1.7 Ω
RON=ROFF= 1 Ω
The peak source and sink current in this case are:
VDD - VEE
ROH_EFF +RON +RG _Int
Isource _ pk = min(10A,
) ö 5.9A
VDD - VEE
ROL +ROFF +RG _Int
Isink _ pk = min(10A,
) ö 6.7A
(2)
Thus by using 1Ω external gate resistance, the peak source current is 5.9A, the peak sink current is 6.7A. The
collector-to-emitter dV/dt during the turn on switching transient is dominated by the gate current at the miller
plateau voltage. The hybrid pullup structure ensures the peak source current at the miller plateau voltage, unless
the turn on gate resistor is too high. The faster the collector-to-emitter, Vce, voltage rises to VDC, the smaller the
turn on switching loss is. The dV/dt can be estimated as Qgc/Isource_pk. For the turn off switching transient, the
drain-to-source dV/dt is dominated by the load current, unless the turn off gate resistor is too high. After Vce
reaches the dc bus voltage, the power semiconductor is in saturation mode and the channel current is controlled
by Vge. The peak sink current determines the dI/dt, which dominates the Vce voltage overshoot accordingly. If
using relatively large turn off gate resistance, the Vce overshoot can be limited. The overshoot can be estimated
by:
DV = Lstray ∂Iload / ((ROFF +ROL +RG_Int )∂Cies ∂ln(Vplat / V ))
ce
th
(3)
Where
•
•
•
•
•
Lstray is the stray inductance in power switching loop, as shown in Figure 9-6
Iload is the load current, which is the turn off current of the power semiconductor
Cies is the input capacitance of the power semiconductor
Vplat is the plateau voltage of the power semiconductor
Vth is the threshold voltage of the power semiconductor
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LDC
Lc1
Lstray=LDC+Le1+Lc1+Le1+Lc1
RG
Lload
t
+
Le1
+
VDC
t
Lc2
VDD
Cgc
Cies=Cgc+Cge
RG
OUTH
OUTL
COM
Cge
Le2
Figure 9-6. Stray Parasitic Inductance of IGBTs in a Half-Bridge Configuration
The power dissipation should be taken into account to maintain the gate driver within the thermal limit. The
power loss of the gate driver includes the quiescent loss and the switching loss, which can be calculated as:
P
= PQ +P
DR
SW
(4)
PQ is the quiescent power loss for the driver, which is Iq x (VDD-VEE) = 5mA x 20V = 0.100W. The quiescent
power loss is the power consumed by the internal circuits such as the input stage, reference voltage, logic
circuits, protection circuits when the driver is swithing when the driver is biased with VDD and VEE, and also the
charging and discharing current of the internal circuit when the driver is switching. The power dissipation when
the driver is switching can be calculated as:
ROH_EFF
2 ROH_EFF +RON +RG _Int ROL +ROFF +RG _Int
ROL
1
P
=
∂(
+
)∂(VDD - VEE)∂ fsw ∂Qg
SW
(5)
Where
•
•
Qg is the gate charge required at the operation point to fully charge the gate voltage from VEE to VDD
fsw is the switching frequency
In this example, the PSW can be calculated as:
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ROH_EFF
ROL
1
P
=
∂(
+
)∂(VDD - VEE)∂ fsw ∂Qg = 0.505W
SW
2 ROH_EFF + RON + RG _Int ROL +ROFF +RG _Int
(6)
Thus, the total power loss is:
P =P +P = 0.10W +0.505W = 0.605W
DR
Q
SW
(7)
When the board temperature is 125°C, the junction temperature can be estimated as:
Tj = T + yjb ∂P ö 150oC
b
DR
(8)
Therefore, for the application in this example, with 125°C board temperature, the maximum switching frequency
is ~50kHz to keep the gate driver in the thermal limit. By using a lower switching frequency, or increasing
external gate resistance, the gate driver can be operated at a higher switching frequency.
9.2.2.6 External Active Miller Clamp
External active miller clamp feature allows the gate driver to stay at the low status when the gate voltage is
detected below VCLMPTH. When the other switch of the phase leg turns on, the dV/dt can cause a current through
the parasitic miller capacitance of the switch and sink in the gate driver. The sinking current causes a negative
voltage drop on the turn off gate resistance, and bumps up the gate voltage to cause a false turn on. The
external active miller clamp features allows flexibility of board layout and active miller clamp pulldown strength.
Limited by the board layout, if the driver cannot be placed close enough to the switch, external active miller
clamp MOSFET can be placed close to the switch and the MOSFET can be chosen according to the peak
current needed. Caution must be exercised when the driver is place far from the power semiconductor. Since the
device has high peak sink and source current, the high dI/dt in the gate loop can cause a ground bounce on the
board parasitics. The ground bounce can cause a positive voltage bump on CLMPE pin during the turn off
transient, and results in the external active miller clamp MOSFET to turn on shortly and add extra drive strength
to the sink current. To reduce the ground bounce, a 2Ω resistance is recommended to the gate of the external
active clamp MOSFET.
When the VOUTH is detected to be lower than VCLMPTH above VEE, the CLMPE pin outputs a 5V voltage with
respect to VEE, the external clamp FET is in linear region and the pulldown current is determined by the peak
drain current, unless the on-resistance of the external clamp FET is large.
VDS
ICLMPE _PK = min(ID _PK
,
)
RDS _ ON
(9)
Where
•
•
•
ID_PK is the peak drain current of the external clamp FET
VDS is the drain-to-source voltage of the clamp FET when the CLMPE is activated
RDS_ON is the on-resistance of the external clamp FET
The total delay time of the active miller clamp circuit from the gate voltage detection threshold VCLMPTH can be
calculated as tDCLMPE+tCLMPER. tCLMPER depends on the parameter of the external active miller clamp MOSFET.
As long as the total delay time is longer than the deadtime of high side and low side switches, the driver can
effectively protect the switch from false turn on issue caused by miller effect.
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VCLMPTH
VCC
OUTH
+
3V to 5.5V
IN+
œ
CLMPE
OUTL
Control
Circuitry
µC
MOD
DEMOD
IN-
VEE
COM
VCC
Figure 9-7. External Active Miller Clamp Configuration
9.2.2.7 Overcurrent and Short Circuit Protection
Fast and reliable overcurrent and short circuit protection is important to protect the catastrophic break down of
the SiC MOSFET and IGBT modules, and improve the system reliability. The UCC21732 features a state-of-art
overcurrent and short circuit protection, which can be applied to both SiC MOSFET and IGBT modules with
various detection circuits.
9.2.2.7.1 Protection Based on Power Modules with Integrated SenseFET
The overcurrent and short circuit protection function is suitable for the SiC MOSFET and IGBT modules with
integrated SenseFET. The SenseFET scales down the main power loop current and outputs the current with a
dedicated pin of the power module. With external high precision sensing resistor, the scaled down current can be
measured and the main power loop current can be calculated. The value of the sensing resistor RS sets the
protection threshold of the main current. For example, with a ratio of 1:N = 1:50000 of the integrated current
mirror, by using the RS as 20Ω, the threshold protection current is:
VOCTH
IOC _ TH
=
∂N = 1750A
RS
(10)
The overcurrent and short circuit protection based on integrated SenseFET has high precision, as it is sensing
the current directly. The accuracy of the method is related to two factors: the scaling down ratio of the main
power loop current and the SenseFET, and the precision of the sensing resistor. Since the current is sensed from
the SenseFET, which is isolated from the main power loop, and the current is scaled down significantly with
much less dI/dt, the sensing loop has good noise immunity. To further improve the noise immunity, a low pass
filter can be added. A 100pF to 10nF filter capacitor can be added. The delay time caused by the low pass filter
should also be considered for the protection circuitry design.
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OUTL
ROFF
SenseFET
Kelvin
Connection
OC
+
FLT
DEMOD
MOD
RFLT
+
VOCTH
œ
RS
CFLT
Control
Logic
GND
COM
VEE
Figure 9-8. Overcurrent and Short Circuit Protection Based on IGBT Module with SenseFET
9.2.2.7.2 Protection Based on Desaturation Circuit
For SiC MOSFET and IGBT modules without SenseFET, desaturation (DESAT) circuit is the most popular circuit
which is adopted for overcurrent and short circuit protection. The circuit consists of a current source, a resistor, a
blanking capacitor and a diode. Normally the current source is provided from the gate driver, when the device
turns on, a current source charges the blanking capacitor and the diode forward biased. During normal
operation, the capacitor voltage is clamped by the switch VCE voltage. When short circuit happens, the capacitor
voltage is quickly charged to the threshold voltage which triggers the device shutdown. For the UCC21732, the
OC pin does not feature an internal current source. The current source should be generated externally from the
output power supply. When UCC21732 is in OFF state, the OC pin is pulled down by an internal MOSFET, which
creates an offset voltage on OC pin. By choosing R1 and R2 significantly higher than the pulldown resistance of
the internal MOSFET, the offset can be ignored. When UCC21732 is in ON state, the OC pin is high impedance.
The current source is generated by the output power supply VDD and the external resistor divider R1, R2 and
R3. The overcurrent detection threshold voltage of the IGBT is:
R2 + R3
R3
VDET =VOCTH
∂
-VF
(11)
(12)
The blanking time of the detection circuit is:
R1 + R2
R1 + R2 + R3
R1 + R2 + R3 VOCTH
tBLK = -
∂R3 ∂CBLK ∂ln(1-
∂
)
R3
VDD
Where:
•
•
•
•
VOCTH is the detection threshold voltage of the gate driver
R1, R2 and R3 are the resistance of the voltage divider
CBLK is the blanking capacitor
VF is the forward voltage of the high voltage diode DHV
The modified desaturation circuit has all the benefits of the conventional desaturation circuit. The circuit has
negligible power loss, and is easy to implement. The detection threshold voltage of IGBT and blanking time can
be programmed by external components. Different with the conventional desaturation circuit, the overcurrent
detection threshold voltage of the IGBT can be modified to any voltage level, either higher or lower than the
detection threshold voltage of the driver. A parallel schottky diode can be connected between OC and COM pins
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to prevent the negative voltage on the OC pin in noisy system. Since the desaturation circuit measures the VCE
of the IGBT or VDS of the SiC MOSFET, not directly the current, the accuracy of the protection is not as high as
the SenseFET based protection method. The current threshold cannot be accurately controlled in the protection.
ROFF
RDESAT
DHV
VDD
R1
OC
R2
+
FLT
DEMOD
MOD
+
R3
CBLK
VOCTH
œ
Control
Logic
GND
COM
VEE
Figure 9-9. Overcurrent and Short Circuit Protection Based on Desaturation Circuit
9.2.2.7.3 Protection Based on Shunt Resistor in Power Loop
In lower power applications, to simplify the circuit and reduce the cost, a shunt resistor can be used in series in
the power loop and measure the current directly. Since the resistor is in series in the power loop, it directly
measures the current and can have high accuracy by using a high precision resistor. The resistance needs to be
small to reduce the power loss, and should have large enough voltage resolution for the protection. Since the
sensing resistor is also in series in the gate driver loop, the voltage drop on the sensing resistor can cause the
voltage drop on the gate voltage of the IGBT or SiC MOSFET modules. The parasitic inductance of the sensing
resistor and the PCB trace of the sensing loop will also cause a noise voltage source during switching transient,
which makes the gate voltage oscillate. Thus, this method is not recommended for high power application, or
when dI/dt is high. To use it in low power application, the shunt resistor loop should be designed to have the
optimal voltage drop and minimum noise injection to the gate loop.
ROFF
OC
RFLT
+
FLT
DEMOD
MOD
+
RS
CFLT
VOCTH
œ
Control
Logic
GND
COM
VEE
Figure 9-10. Overcurrent and Short Circuit Protection Based on Shunt Resistor
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9.2.2.8 Isolated Analog Signal Sensing
The isolated analog signal sensing feature provides a simple isolated channel for the isolated temperature
detection, voltage sensing and etc. One typical application of this function is the temperature monitor of the
power semiconductor. Thermal diodes or temperature sensing resistors are integrated in the SiC MOSFET or
IGBT module close to the dies to monitor the junction temperature. UCC21732 has an internal 200uA current
source with 3% accuracy across temperature, which can forward bias the thermal diodes or create a voltage
drop on the temperature sensing resistors. The sensed voltage from the AIN pin is passed through the isolation
barrier to the input side and transformed to a PWM signal. The duty cycle of the PWM changes linearly from
10% to 88% when the AIN voltage changes from 4.5V to 0.6V and can be represented using Equation 13.
DAPWM(%) = -20 * VAIN +100
(13)
9.2.2.8.1 Isolated Temperature Sensing
A typical application circuit is shown in Figure 9-11. To sense temperature, the AIN pin is connected to the
thermal diode or thermistor which can be discrete or integrated within the power module. A low pass filter is
recommended for the AIN input. Since the temperature signal does not have a high bandwidth, the low pass
filter is mainly used for filtering the noise introduced by the switching of the power device, which does not require
stringent control for propagation delay. The filter capacitance for Cfilt can be chosen between 1nF to 100nF and
the filter resistance Rfilt between 1Ω to 10Ω according to the noise level.
The output of APWM is directly connected to the microcontroller to measure the duty cycle dependent on the
voltage input at AIN, using Equation 13.
UCC21732-Q1
In Module or
Discrete
VCC
VDD
13V to
33V
+
+
3V to 5.5V
APWM
œ
œ
AIN
+
DEMOD
MOD
µC
Rfilt
Cfilt
OSC
GND
COM
Thermal
Diode
NTC or
PTC
Figure 9-11. Thermal Diode or Thermistor Temperature Sensing Configuration
When a high-precision voltage supply for VCC is used on the primary side of UCC21732 the duty cycle output of
APWM may also be filtered and the voltage measured using the microcontroller's ADC input pin, as shown in
Figure 9-12. The frequency of APWM is 400kHz, so the value for Rfilt_2 and Cfilt_2 should be such that the cutoff
frequency is below 400kHz. Temperature does not change rapidly, thus the rise time due to the RC constant of
the filter is not under a strict requirement.
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VDD
AIN
In Module or
Discrete
VCC
13V to
33V
+
+
œ
3V to 5.5V
APWM
œ
+
DEMOD
MOD
µC
Rfilt_1
Rfilt_2
Cfilt_2
GND
OSC
Cfilt_1
COM
Thermal
Diode
NTC or
PTC
Figure 9-12. APWM Channel with Filtered Output
The example below shows the results using a 4.7kΩ NTC, NTCS0805E3472FMT, in series with a 3kΩ resistor
and also the thermal diode using four diode-connected MMBT3904 NPN transistors. The sensed voltage of the 4
MMBT3904 thermal diodes connected in series ranges from about 2.5V to 1.6V from 25°C to 135°C,
corresponding to 50% to 68% duty cycle. The sensed voltage of the NTC thermistor connected in series with the
3kΩ resistor ranges from about 1.5V to 0.6V from 25°C to 135°C, corresponding to 70% to 88% duty cycle. The
voltage at VAIN of both sensors and the corresponding measured duty cycle at APWM is shown in Figure 9-13.
2.7
2.4
2.1
1.8
1.5
1.2
0.9
0.6
90
84
78
72
66
60
54
Thermal Diode VAIN
NTC VAIN
Thermal Diode APWM
NTC APWM
48
20
40
60
80
Temperature (èC)
100
120
140
VAIN
Figure 9-13. Thermal diode and NTC VAIN and Corresponding Duty Cycle at APWM
The duty cycle output has an accuracy of ±3% throughout temperature without any calibration, as shown in
Figure 9-14 but with single-point calibration at 25°C, the duty accuracy can be improved to ±1%, as shown in
Figure 9-15.
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1.5
1.25
1
Thermal Diode APWM Duty Error
NTC APWM Duty Error
0.75
0.5
0.25
0
-0.25
20
40
60
80
Temperature (èC)
100
120
140
APWM
Figure 9-14. APWM Duty Error Without Calibration
0.8
0.6
0.4
0.2
0
Thermal Diode APWM Duty Error
NTC APWM Duty Error
-0.2
20
40
60
80
Temperature (èC)
100
120
140
APWM
Figure 9-15. APWM Duty Error with Single-Point Calibration
9.2.2.8.2 Isolated DC Bus Voltage Sensing
The AIN to APWM channel may be used for other applications such as the DC-link voltage sensing, as shown in
Figure 9-16. The same filtering requirements as given above may be used in this case, as well. The number of
attenuation resistors, Ratten_1 through Ratten_n, is dependent on the voltage level and power rating of the resistor.
The voltage is finally measured across RLV_DC to monitor the stepped-down voltage of the HV DC-link which
must fall within the voltage range of AIN from 0.6V to 4.5V. The driver should be referenced to the same point as
the measurement reference, thus in the case shown below the UCC21732 is driving the lower IGBT in the half-
bridge and the DC-link voltage measurement is referenced to COM. The internal current source IAIN should be
taken into account when designing the resistor divider. The AIN pin voltage is:
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RLV _DC
n
VAIN
=
∂ VDC +RLV _DC ∂IAIN
RLV _DC
+
R
atten _ i
ƒ
i=1
(14)
Ratten_1
Ratten_2
VDD
VCC
Ratten_n
13V to
33V
+
+
3V to 5.5V
APWM
œ
œ
CDC
+
AIN
DEMOD
MOD
µC
Rfilt
Cfilt
Rfilt_2
Cfilt_2
GND
RLV_DC
OSC
COM
Figure 9-16. DC-link Voltage Sensing Configuration
9.2.2.9 Higher Output Current Using an External Current Buffer
To increase the IGBT gate drive current, a non-inverting current buffer (such as the NPN/PNP buffer shown in
Figure 9-17) can be used. Inverting types are not compatible with the desaturation fault protection circuitry and
must be avoided. The MJD44H11/MJD45H11 pair is appropriate for peak currents up to 15 A, the D44VH10/
D45VH10 pair is up to 20 A peak.
In the case of a over-current detection, the soft turn off (STO) is activated. External components must be added
to implement STO instead of normal turn off speed when an external buffer is used. CSTO sets the timing for soft
turn off and RSTO limits the inrush current to below the current rating of the internal FET (10A). RSTO should be at
least (VDD-VEE)/10. The soft turn off timing is determined by the internal current source of 400mA and the
capacitor CSTO. CSTO is calculated using .
ISTO ∂ tSTO
VDD VEE
CSTO
=
(15)
•
•
ISTO is the the internal STO current source, 400mA
tSTO is the desired STO timing
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VDD
VDD
ROH
Cies=Cgc+Cge
OUTH
OUTL
RNMOS
Cgc
Cgc
RG_2
RG_1
RG_Int
RG_Int
Cge
Cge
ROL
CSTO
COM
VEE
RSTO
Figure 9-17. Current Buffer for Increased Drive Strength
10 Power Supply Recommendations
During the turn on and turn off switching transient, the peak source and sink current is provided by the VDD and
VEE power supply. The large peak current is possible to drain the VDD and VEE voltage level and cause a
voltage droop on the power supplies. To stabilize the power supply and ensure a reliable operation, a set of
decoupling capacitors are recommended at the power supplies. Considering UCC21732 has ±10A peak drive
strength and can generate high dV/dt, a 10µF bypass cap is recommended between VDD and COM, VEE and
COM. A 1µF bypass cap is recommended between VCC and GND due to less current comparing with output
side power supplies. A 0.1µF decoupling cap is also recommended for each power supply to filter out high
frequency noise. The decoupling capacitors must be low ESR and ESL to avoid high frequency noise, and
should be placed as close as possible to the VCC, VDD and VEE pins to prevent noise coupling from the system
parasitics of PCB layout.
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11 Layout
11.1 Layout Guidelines
Due to the strong drive strength of UCC21732, careful considerations must be taken in PCB design. Below are
some key points:
•
The driver should be placed as close as possible to the power semiconductor to reduce the parasitic
inductance of the gate loop on the PCB traces
•
The decoupling capacitors of the input and output power supplies should be placed as close as possible to
the power supply pins. The peak current generated at each switching transient can cause high dI/dt and
voltage spike on the parasitic inductance of PCB traces
•
The driver COM pin should be connected to the Kelvin connection of SiC MOSFET source or IGBT emitter. If
the power device does not have a split Kelvin source or emitter, the COM pin should be connected as close
as possible to the source or emitter terminal of the power device package to separate the gate loop from the
high power switching loop
•
•
Use a ground plane on the input side to shield the input signals. The input signals can be distorted by the
high frequency noise generated by the output side switching transients. The ground plane provides a low-
inductance filter for the return current flow
If the gate driver is used for the low side switch which the COM pin connected to the dc bus negative, use the
ground plane on the output side to shield the output signals from the noise generated by the switch node; if
the gate driver is used for the high side switch, which the COM pin is connected to the switch node, ground
plane is not recommended
•
•
If ground plane is not used on the output side, separate the return path of the OC and AIN ground loop from
the gate loop ground which has large peak source and sink current
No PCB trace or copper is allowed under the gate driver. A PCB cutout is recommended to avoid any noise
coupling between the input and output side which can contaminate the isolation barrier
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11.2 Layout Example
Figure 11-1. Layout Example
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12 Device and Documentation Support
12.1 Documentation Support
12.1.1 Related Documentation
For related documentation see the following:
•
Isolation Glossary
12.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
12.3 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
12.4 Trademarks
TI E2E™ is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
12.6 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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20-Aug-2020
PACKAGING INFORMATION
Orderable Device
Status Package Type Package Pins Package
Eco Plan
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
Samples
Drawing
Qty
(1)
(2)
(3)
(4/5)
(6)
UCC21732DW
PREVIEW
SOIC
SOIC
DW
16
16
40
Green (RoHS
& no Sb/Br)
NIPDAU
Level-2-260C-1 YEAR
Level-3-260C-168 HR
-40 to 125
-40 to 125
UCC21732
UCC21732
UCC21732DWR
PREVIEW
DW
2000
Green (RoHS
& no Sb/Br)
NIPDAU
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6)
Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
20-Aug-2020
Addendum-Page 2
IMPORTANT NOTICE AND DISCLAIMER
TI PROVIDES TECHNICAL AND RELIABILITY DATA (INCLUDING DATASHEETS), DESIGN RESOURCES (INCLUDING REFERENCE
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IMPLIED WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE OR NON-INFRINGEMENT OF THIRD
PARTY INTELLECTUAL PROPERTY RIGHTS.
These resources are intended for skilled developers designing with TI products. You are solely responsible for (1) selecting the appropriate
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Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2020, Texas Instruments Incorporated
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TI
UCC21736QDWRQ1
适用于 IGBT/SiC MOSFET 且具有主动短路的汽车类 5.7kVrms ±10A 单通道隔离式栅极驱动器 | DW | 16 | -40 to 125Warning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
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TI
UCC21737-Q1
适用于 SiC/IGBT、具有主动短路保护功能的汽车类 10A 隔离式单通道栅极驱动器Warning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
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TI
UCC21737QDWRQ1
适用于 SiC/IGBT、具有主动短路保护功能的汽车类 10A 隔离式单通道栅极驱动器 | DW | 16 | -40 to 125Warning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
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TI
UCC21739-Q1
适用于 IGBT/SiC MOSFET 且具有隔离式模拟检测的汽车类 3kVrms ±10A 单通道隔离式栅极驱动器Warning: Undefined variable $rtag in /www/wwwroot/website_ic37/www.icpdf.com/pdf/pdf/index.php on line 217
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TI
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