UCC2817NG4 [TI]
BiCMOS POWER FACTOR PREREGULATOR; BiCMOS功率因数前置稳压器
| 型号： | UCC2817NG4 |
| 厂家： | 
TEXAS INSTRUMENTS |
| 描述： | BiCMOS POWER FACTOR PREREGULATOR |
| 文件： | 总32页 (文件大小：948K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
ꢀꢁꢁ ꢂ ꢃꢄ ꢅ ꢆ ꢀꢁꢁ ꢂ ꢃꢄ ꢃ ꢆ ꢀꢁꢁ ꢇ ꢃꢄ ꢅ ꢆ ꢀ ꢁꢁ ꢇꢃ ꢄꢃ
ꢈꢉ ꢁꢊꢋ ꢌ ꢍꢋ ꢎ ꢏꢐ ꢑꢒꢁꢓꢋ ꢐ ꢍꢐ ꢏꢐꢏꢔ ꢀꢕ ꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
D
Controls Boost Preregulator to Near-Unity
Power Factor
D, DW, N, and PW PACKAGES
(TOP VIEW)
D
D
D
D
D
D
D
D
D
D
D
D
Limits Line Distortion
GND
PKLMT
CAOUT
CAI
DRVOUT
VCC
CT
World Wide Line Operation
Over-Voltage Protection
1
2
3
4
5
6
7
8
16
15
14
13
12
Accurate Power Limiting
SS
Average Current Mode Control
Improved Noise Immunity
Improved Feed-Forward Line Regulation
Leading Edge Modulation
MOUT
IAC
RT
11 VSENSE
10 OVP/EN
VAOUT
VFF
9
VREF
150-µA Typical Start-Up Current
Low-Power BiCMOS Operation
12-V to 17-V Operation
Frequency Range 6 kHz to 220 kHz
description
The UCCx817/18 family provides all the functions necessary for active power factor corrected preregulators.
The controller achieves near unity power factor by shaping the ac input line current waveform to correspond
to that of the ac input line voltage. Average current mode control maintains stable, low distortion sinusoidal line
current.
Designed in Texas Instrument’s BiCMOS process, the UCC2817/UCC2818 offers new features such as lower
start-up current, lower power dissipation, overvoltage protection, a shunt UVLO detect circuitry, a leading-edge
modulation technique to reduce ripple current in the bulk capacitor and an improved, low-offset ( 2 mV) current
amplifier to reduce distortion at light load conditions.
block diagram
VCC
15
OVP/EN 10
SS 13
16 V (FOR UCC2817 ONLY)
7.5 V
REFERENCE
9
VREF
UVLO
1.9 V
ENABLE
−
+
VAOUT
7
16 V/10 V (UCC2817)
10.5 V/10 V (UCC2818)
ZERO POWER
0.33 V
VCC
−
+
VOLTAGE
ERROR AMP
VSENSE 11
7.5 V
−
+
CURRENT
AMP
8.0 V
PWM
OVP
Q
+
−
X
÷
MULT
−
+
16 DRVOUT
−
+
X
S
2
VFF
8
X
PWM
LATCH
OSC
CLK
R
R
MIRROR
2:1
1
2
GND
CLK
OSCILLATOR
IAC
6
5
PKLMT
−
+
MOUT
4
3
12
14
UDG-98182
CAI
CAOUT RT
CT
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
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Copyright 2006 − 2009, Texas Instruments Incorporated
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ꢠ ꢤ ꢡ ꢠꢉ ꢚꢮ ꢜꢛ ꢟ ꢧꢧ ꢥꢟ ꢝ ꢟ ꢞ ꢤ ꢠ ꢤ ꢝ ꢡ ꢩ
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1
www.ti.com
ꢀ ꢁꢁꢂ ꢃ ꢄ ꢅ ꢆ ꢀ ꢁꢁ ꢂ ꢃꢄ ꢃ ꢆ ꢀꢁ ꢁꢇ ꢃ ꢄ ꢅ ꢆ ꢀꢁ ꢁꢇ ꢃ ꢄ ꢃ
ꢈ ꢉꢁꢊ ꢋꢌ ꢍꢋꢎ ꢏꢐ ꢑꢒꢁ ꢓꢋ ꢐ ꢍ ꢐꢏ ꢐꢏ ꢔꢀ ꢕꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
description (continued)
UCC2817 offers an on-chip shunt regulator with low start-up current, suitable for applications utilizing a
bootstrap supply. UCC2818 is intended for applications with a fixed supply (VCC).
Available in the 16-pin D, DW, N and PW packages.
†
absolute maximum ratings over operating free-air temperature (unless otherwise noted)
Supply voltage VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V
Supply current ICC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 mA
Gate drive current, continuous . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.2 A
Gate drive current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.2 A
Input voltage, CAI, MOUT, SS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 V
Input voltage, PKLMT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 V
Input voltage, VSENSE, OVP/EN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 V
Input current, RT, IAC, PKLMT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 mA
Input current, VCC (no switching) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 mA
Maximum negative voltage, DRVOUT, PKLMT, MOUT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.5 V
Power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 W
Junction temperature, T . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −55°C to 150°C
J
stg
Storage temperature, T
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 150°C
Lead temperature, T (soldering, 10 seconds) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300°C
Power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 W
sol
†
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied.
Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
AVAILABLE OPTIONS
PACKAGE DEVICES
SOIC (D) PACKAGE
SOIC (DW) PACKAGE
PDIP (N) PACKAGE
TSSOP (PW) PACKAGE
T
A
= T
J
Turn-on
Threshold
16 V
Turn-on
Threshold
10.2 V
Turn-on
Threshold
16 V
Turn-on
Threshold
10.2 V
Turn-on
Threshold
16 V
Turn-on
Threshold
10.2 V
Turn-on
Threshold
16 V
Turn-on
Threshold
10.2 V
−40°C to 85°C
0°C to 70°C
UCC2817D
UCC3817D
UCC2818D UCC2817DW UCC2818DW
UCC3818D UCC3817DW UCC3818DW
UCC2817N
UCC3817N
UCC2818N UCC2817PW UCC2818PW
UCC3818N UCC3817PW UCC3818PW
THERMAL RESISTANCE TABLE
PACKAGE
θjc(°C/W)
θja(°C/W)
(1)
(1)
(1)
SOIC−16 (D)
SOIC−16 (DW)
PDIP−16 (N)
22
26
12
40 to 70
89 to 102
25 to 50
(2)
(2)
TSSOP−16 (PW)
14
123 to 147
2
NOTES: (1) Specifiedθja (junction to ambient) is for devices mounted to 5-inch FR4 PC board with one ounce copper
2
where noted. When resistance range is given, lower values are for 5 inch aluminum PC board. Test PWB
was 0.062 inch thick and typically used 0.635-mm trace widths for power packages and 1.3-mm trace
widths for non-power packages with a 100-mil x 100-mil probe land area at the end of each trace.
(2). Modeled data. If value range given for θja, lower value is for 3x3 inch. 1 oz internal copper ground plane,
higher value is for 1x1-inch. ground plane. All model data assumes only one trace for each non-fused
lead.
2
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ꢀꢁꢁ ꢂ ꢃꢄ ꢅ ꢆ ꢀꢁꢁ ꢂ ꢃꢄ ꢃ ꢆ ꢀꢁꢁ ꢇ ꢃꢄ ꢅ ꢆ ꢀ ꢁꢁ ꢇꢃ ꢄꢃ
ꢈꢉ ꢁꢊꢋ ꢌ ꢍꢋ ꢎ ꢏꢐ ꢑꢒꢁꢓꢋ ꢐ ꢍꢐ ꢏꢐꢏꢔ ꢀꢕ ꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
electrical characteristics, T = 0°C to 70°C for the UCC3817 and T = −40°C to 85°C for the UCC2817, T
A
A
T
A
= T VCC = 12 V, R = 22 kΩ, C = 270 pF, (unless otherwise noted)
J,
T
supply current section
PARAMETER
Supply current, off
TEST CONDITIONS
VCC = (VCC turn-on threshold −0.3 V)
VCC = 12 V, No load on DRVOUT
MIN
TYP
150
4
MAX UNITS
300
6
µA
Supply current, on
2
mA
UVLO section
PARAMETER
TEST CONDITIONS
MIN
15.4
9.4
TYP
16
MAX UNITS
VCC turn-on threshold (UCCx817)
VCC turn-off threshold (UCCx817)
UVLO hysteresis (UCCx817)
16.6
V
V
V
V
V
V
V
9.7
6.3
17
5.8
Maximum shunt voltage (UCCx817)
VCC turn-on threshold (UCCx818)
VCC turn-off threshold (UCCx818)
UVLO hysteresis (UCCx818)
I
= 10 mA
15.4
9.7
17.5
10.8
VCC
10.2
9.7
0.5
9.4
0.3
voltage amplifier section
PARAMETER
TEST CONDITIONS
MIN
7.387
7.369
TYP
MAX UNITS
T
= 0°C to 70°C
7.5 7.613
7.5 7.631
V
V
A
Input voltage
T
A
= −40°C to 85°C
V
bias current
V
= V
,
VAOUT = 2.5 V
50
90
200
nA
dB
V
SENSE
SENSE
VAOUT = 2 V to 5 V
REF
Open loop gain
50
5.3
0
High-level output voltage
Low-level output voltage
I
= −150 µA
= 150 µA
5.5
50
5.6
L
L
I
150
mV
over voltage protection and enable section
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNITS
VREF VREF VREF
+0.48 +0.50 +0.52
Over voltage reference
V
Hysteresis
300
1.7
0.1
500
1.9
0.2
600
2.1
0.3
mV
V
Enable threshold
Enable hysteresis
V
current amplifier section
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNITS
Input offset voltage
V
V
V
V
V
= 0 V,
V
V
V
V
V
= 3 V
= 3 V
= 3 V
−3.5
0
2.5
mV
nA
nA
dB
dB
V
CM
CM
CM
CM
CM
CAOUT
CAOUT
CAOUT
CAOUT
CAOUT
Input bias current
= 0 V,
−50 −100
Input offset current
= 0 V,
25
100
Open loop gain
= 0 V,
= 2 V to 5 V
= 3 V
90
60
Common-mode rejection ratio
High-level output voltage
Low-level output voltage
Gain bandwidth product
= 0 V to 1.5 V,
80
6.5
0.2
2.5
I
I
= −120 µA
5.6
0.1
6.8
0.5
L
= 1 mA
V
L
See Note 1
MHz
NOTES: 1. Ensured by design, not production tested.
3
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ꢀ ꢁꢁꢂ ꢃ ꢄ ꢅ ꢆ ꢀ ꢁꢁ ꢂ ꢃꢄ ꢃ ꢆ ꢀꢁ ꢁꢇ ꢃ ꢄ ꢅ ꢆ ꢀꢁ ꢁꢇ ꢃ ꢄ ꢃ
ꢈ ꢉꢁꢊ ꢋꢌ ꢍꢋꢎ ꢏꢐ ꢑꢒꢁ ꢓꢋ ꢐ ꢍ ꢐꢏ ꢐꢏ ꢔꢀ ꢕꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
electrical characteristics, T = 0°C to 70°C for the UCC3817 and T = −40°C to 85°C for the UCC2817, T
A
T
A
A
= T VCC = 12 V, R = 22 kΩ, C = 270 pF, (unless otherwise noted)
J,
T
voltage reference section
PARAMETER
TEST CONDITIONS
MIN
7.387
7.369
0
TYP
MAX UNITS
T
= 0°C to 70°C
7.5 7.613
V
A
Input voltage
T
A
= −40°C to 85°C
7.5 7.631
V
Load regulation
I
= 1 mA to 2 mA
10
10
mV
mV
mA
REF
VCC = 10.8 V to 15 V,
= 0 V
Line regulation
See Note 2
0
Short-circuit current
V
−20
−25
−50
REF
< 10.8 V is shown in Figure 8.
NOTES: 2. Reference variation for V
CC
oscillator section
PARAMETER
Initial accuracy
TEST CONDITIONS
MIN
85
TYP
MAX UNITS
T
A
= 25°C
100
115
1
kHz
%
Voltage stability
VCC = 10.8 V to 15 V
Line, temp
−1
Total variation
80
120
5.5
kHz
V
Ramp peak voltage
4.5
5
4
Ramp amplitude voltage
(peak to peak)
3.5
4.5
V
peak current limit section
PARAMETER
PKLMT reference voltage
PKLMT propagation delay
TEST CONDITIONS
MIN
−15
150
TYP
MAX UNITS
15
mV
ns
350
500
multiplier section
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX UNITS
I
, high line, low power output
MOUT
I
I
I
I
I
= 500 µA,
= 500 µA,
= 500 µA,
= 150 µA,
= 150 µA,
V
V
V
V
V
= 4.7 V,
= 4.7 V,
= 4.7 V,
= 1.4 V,
= 1.4 V,
VAOUT = 1.25 V
VAOUT = 1.25 V
VAOUT = 5 V
0
−6
−20
−23
µA
µA
µA
µA
µA
AC
AC
AC
AC
AC
FF
FF
FF
FF
FF
current, (0°C to 85°C)
I
, high line, low power output
MOUT
current, (−40°C to 85°C)
0
−70
−10
−6
I
, high line, high power output
, low line, low power output
, low line, high power output
, IAC limited output current
MOUT
current
−90 −105
−19 −50
I
MOUT
current
VAOUT = 1.25 V
VAOUT = 5 V
I
MOUT
current
−268 −300 −345
−250 −300 −400
I
I
I
I
I
I
I
I
= 150 µA,
= 300 µA,
= 150 µA,
= 500 µA,
= 500 µA,
= 500 µA,
= 150 µA,
V
FF
V
FF
V
FF
V
FF
V
FF
V
FF
V
FF
= 1.3 V,
= 3 V,
VAOUT = 5 V
µA
1/V
µA
µA
µA
µA
µW
MOUT
Gain constant (K)
AC
AC
AC
AC
AC
AC
AC
VAOUT = 2.5 V
VAOUT = 0.25 V
VAOUT = 0.25 V
VAOUT = 0.5 V
VAOUT = 0.5 V
VAOUT = 5 V
0.5
1
0
0
0
0
1.5
−2
= 1.4 V,
= 4.7 V,
= 4.7 V,
= 4.7 V,
= 1.4 V,
I
, zero current
MOUT
−2
I
I
, zero current, (0°C to 85°C)
, zero current, (−40°C to 85°C)
−3
MOUT
−3.5
MOUT
Power limit (I
MOUT
x V )
FF
−375 −420 −485
4
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ꢀꢁꢁ ꢂ ꢃꢄ ꢅ ꢆ ꢀꢁꢁ ꢂ ꢃꢄ ꢃ ꢆ ꢀꢁꢁ ꢇ ꢃꢄ ꢅ ꢆ ꢀ ꢁꢁ ꢇꢃ ꢄꢃ
ꢈꢉ ꢁꢊꢋ ꢌ ꢍꢋ ꢎ ꢏꢐ ꢑꢒꢁꢓꢋ ꢐ ꢍꢐ ꢏꢐꢏꢔ ꢀꢕ ꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
electrical characteristics, T = 0°C to 70°C for the UCC3817 and T = −40°C to 85°C for the UCC2817, T
A
T
A
A
= T VCC = 12 V, R = 22 kΩ, C = 270 pF, (unless otherwise noted)
J,
T
feed-forward section
PARAMETER
TEST CONDITIONS
TEST CONDITIONS
TEST CONDITIONS
MIN
TYP
MAX UNITS
µA
VFF output current
I
= 300 µA
−140 −150 −160
AC
soft start section
PARAMETER
MIN
TYP
MAX UNITS
−16 µA
SS charge current
−6
−10
gate driver section
PARAMETER
Pullup resistance
MIN
TYP
5
MAX UNITS
I
I
= –100 mA to −200 mA
= 100 mA
12
10
50
50
99
2
Ω
Ω
O
Pulldown resistance
Output rise time
2
O
C
C
= 1 nF,
= 1 nF,
R
R
= 10 Ω,
= 10 Ω,
V
V
= 0.7 V to 9.0 V
= 9.0 V to 0.7 V
25
10
95
ns
ns
%
%
L
L
L
L
DRVOUT
Output fall time
DRVOUT
Maximum duty cycle
Minimum controlled duty cycle
93
At 100 kHz
zero power section
PARAMETER
TEST CONDITIONS
Measured on VAOUT
MIN
TYP
MAX UNITS
0.50
Zero power comparator threshold
0.20
0.33
V
pin descriptions
CAI: (current amplifier noninverting input) Place a resistor between this pin and the GND side of current sense
resistor. This input and the inverting input (MOUT) remain functional down to and below GND.
CAOUT: (current amplifier output) This is the output of a wide bandwidth operational amplifier that senses line
current and commands the PFC pulse-width modulator (PWM) to force the correct duty cycle. Compensation
components are placed between CAOUT and MOUT.
CT: (oscillator timing capacitor) A capacitor from CT to GND sets the PWM oscillator frequency according to:
0.6
RT CT
f [ ǒ
Ǔ
The lead from the oscillator timing capacitor to GND should be as short and direct as possible.
DRVOUT: (gate drive) The output drive for the boost switch is a totem-pole MOSFET gate driver on DRVOUT.
Use a series gate resistor to prevent interaction between the gate impedance and the output driver that might
cause the DRVOUT to overshoot excessively. See characteristic curve (Figure 13) to determine minimum
required gate resister value. Some overshoot of the DRVOUT output is always expected when driving a
capacitive load.
GND: (ground) All voltages measured with respect to ground. VCC and REF should be bypassed directly to
GND with a 0.1-µF or larger ceramic capacitor.
5
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SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
pin descriptions (continued)
IAC: (current proportional to input voltage) This input to the analog multiplier is a current proportional to
instantaneous line voltage. The multiplier is tailored for very low distortion from this current input (I ) to
IAC
multiplier output. The recommended maximum I
is 500 µA.
IAC
MOUT: (multiplier output and current amplifier inverting input) The output of the analog multiplier and the
inverting input of the current amplifier are connected together at MOUT. As the multiplier output is a current, this
is a high-impedance input so the amplifier can be configured as a differential amplifier. This configuration
improves noise immunity and allows for the leading-edge modulation operation. The multiplier output current
is limited to ǒ2 I Ǔ. The multiplier output current is given by the equation:
IAC
I
(V
* 1)
IAC
VAOUT
I
+
MOUT
2
V
K
VFF
1
V
where K + is the multiplier gain constant.
OVP/EN: (over-voltage/enable) A window comparator input that disables the output driver if the boost output
voltage is a programmed level above the nominal or disables both the PFC output driver and resets SS if pulled
below 1.9 V (typ).
PKLMT: (PFC peak current limit) The threshold for peak limit is 0 V. Use a resistor divider from the negative side
of the current sense resistor to VREF to level shift this signal to a voltage level defined by the value of the sense
resistor and the peak current limit. Peak current limit is reached when PKLMT voltage falls below 0 V.
RT: (oscillator charging current) A resistor from RT to GND is used to program oscillator charging current. A
resistor between 10 kΩ and 100 kΩ is recommended. Nominal voltage on this pin is 3 V.
SS: (soft-start) V is discharged for V
with a current source. This voltage is used as the voltage error signal during start-up, enabling the PWM duty
low conditions. When enabled, SS charges an external capacitor
SS
VCC
cycle to increase slowly. In the event of a V
discharges to disable the PWM.
dropout, the OVP/EN is forced below 1.9 V (typ), SS quickly
VCC
Note: In an open-loop test circuit, grounding the SS pin does not ensure 0% duty cycle. Please see the
application section for details.
VAOUT: (voltage amplifier output) This is the output of the operational amplifier that regulates output voltage.
The voltage amplifier output is internally limited to approximately 5.5 V to prevent overshoot.
VCC: (positive supply voltage) Connect to a stable source of at least 20 mA between 10 V and 17 V for normal
operation. Bypass VCC directly to GND to absorb supply current spikes required to charge external MOSFET
gate capacitances. To prevent inadequate gate drive signals, the output devices are inhibited unless V
exceeds the upper under-voltage lockout voltage threshold and remains above the lower threshold.
VCC
VFF: (feed-forward voltage) The RMS voltage signal generated at this pin by mirroring 1/2 of the I
pole external filter. At low line, the VFF voltage should be 1.4 V.
into a single
IAC
VSENSE: (voltage amplifier inverting input) This is normally connected to a compensation network and to the
boost converter output through a divider network.
VREF: (voltage reference output) VREF is the output of an accurate 7.5-V voltage reference. This output is
capable of delivering 20 mA to peripheral circuitry and is internally short-circuit current limited. VREF is disabled
and remains at 0 V when V
is below the UVLO threshold. Bypass VREF to GND with a 0.1-µF or larger
VCC
ceramic capacitor for best stability. Please refer to Figures 8 and 9 for VREF line and load regulation
characteristics.
6
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SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
The UCC3817 is a BiCMOS average current mode boost controller for high power factor, high efficiency
preregulator power supplies. Figure 1 shows the UCC3817 in a 250-W PFC preregulator circuit. Off-line
switching power converters normally have an input current that is not sinusoidal. The input current waveform
has a high harmonic content because current is drawn in pulses at the peaks of the input voltage waveform.
An active power factor correction circuit programs the input current to follow the line voltage, forcing the
converter to look like a resistive load to the line. A resistive load has 0° phase displacement between the current
and voltage waveforms. Power factor can be defined in terms of the phase angle between two sinusoidal
waveforms of the same frequency:
PF + cosQ
Therefore, a purely resistive load would have a power factor of 1. In practice, power factors of 0.999 with THD
(total harmonic distortion) of less than 3% are possible with a well-designed circuit. Following guidelines are
provided to design PFC boost converters using the UCC3817.
NOTE: Schottky diodes, D5 and D6, are required to protect the PFC controller from electrical over stress during
system power up.
C10
1 µ F
C11
1 µF
R16
100Ω
VCC
R15
24k
D7
D8
R21
R13
383k 383k
L1
IAC
1mH
R18
24k
V
D1
8A, 600V
O
F1
AC2
+
D2
6A, 600V
C14
C13
V
F
F
µ
µ
1.5
400V
0.47
600V
LINE
85−270 V
AC
V
Q1
IRFP450
OUT
D3
C12
220µF
450V
385V−DC
AC1
R14
0.25
Ω
3W
6A 600V
−
R17
20Ω
UCC3817
R9
4.02k
R10
4.02k
R12
2k
1
2
GND
DRVOUT 16
D4
VCC
C3
PKLIMIT
1µF CER
D5
VCC 15
R11
10k
3
4
5
CAOUT
CAI
C2
100
µ F AI EI
C1
560pF
V
REF
MOUT
CT 14
SS 13
C9 1.2nF
R8 12k
µ
C4 0.01 F
6
IAC
C8 270pF
R1 12k
RT 12
C7 150nF
D6
µ
R7 100k C15 2.2
F
VSENSE 11
V
R2
499k
R19
499k
O
R3 20k
7
8
VAOUT
VFF
C6 2.2µF
R4
249k
R20 274k
OVP/EN 10
R5
10k
R6 30k
µF
C5 1
VREF
9
V
UDG-98183
REF
Figure 1. Typical Application Circuit
7
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SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
power stage
L
: The boost inductor value is determined by:
BOOST
ǒVIN(min) DǓ
+
L
BOOST
(
)
DI fs
where D is the duty cycle, ∆I is the inductor ripple current and f is the switching frequency. For the example
S
circuit a switching frequency of 100 kHz, a ripple current of 875 mA, a maximum duty cycle of 0.688 and a
minimum input voltage of 85 V
equation are at the peak of low line, where the inductor current and its ripple are at a maximum.
gives us a boost inductor value of about 1 mH. The values used in this
RMS
C
: Two main criteria, the capacitance and the voltage rating, dictate the selection of the output capacitor.
OUT
The value of capacitance is determined by the holdup time required for supporting the load after input ac voltage
is removed. Holdup is the amount of time that the output stays in regulation after the input has been removed.
For this circuit, the desired holdup time is approximately 16 ms. Expressing the capacitor value in terms of output
power, output voltage, and holdup time gives the equation:
ǒ2 P DtǓ
OUT
C
+ ǒV
OUT(min) Ǔ
OUT
2
2
* V
OUT
In practice, the calculated minimum capacitor value may be inadequate because output ripple voltage
specifications limit the amount of allowable output capacitor ESR. Attaining a sufficiently low value of ESR often
necessitates the use of a much larger capacitor value than calculated. The amount of output capacitor ESR
allowed can be determined by dividing the maximum specified output ripple voltage by the inductor ripple
current. In this design holdup time was the dominant determining factor and a 220-µF, 450-V capacitor was
chosen for the output voltage level of 385 VDC at 250 W.
Power switch selection: As in any power supply design, tradeoffs between performance, cost and size have
to be made. When selecting a power switch, it can be useful to calculate the total power dissipation in the switch
for several different devices at the switching frequencies being considered for the converter. Total power
dissipation in the switch is the sum of switching loss and conduction loss. Switching losses are the combination
of the gate charge loss, C
loss and turnon and turnoff losses:
OSS
P
P
P
+ Q
V
fs
GATE
COSS
GATE
GATE
1
2
2
+
C
V
fs
OSS
OFF
1
2
I ǒtON
Ǔ
) P
+
V
) t
fs
ON
OFF
OFF
L
OFF
where Q
is the total gate charge, V
is the gate drive voltage, f is the clock frequency, C
is the drain
OSS
GATE
GATE
L
S
source capacitance of the MOSFET, I is the peak inductor current, t
(estimated using device parameters R
off time, in this case V
and t
are the switching times
ON
OFF
, Q
and V ) and V
is the voltage across the switch during the
GATE GD
TH
OFF
= V
.
OFF
OUT
8
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SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
Conduction loss is calculated as the product of the R
and the square of RMS current:
of the switch (at the worst case junction temperature)
DS(on)
2
P
+ R
K I
COND
DS(on)
RMS
where K is the temperature factor found in the manufacturer’s R
vs. junction temperature curves.
DS(on)
Calculating these losses and plotting against frequency gives a curve that enables the designer to determine
either which manufacturer’s device has the best performance at the desired switching frequency, or which
switching frequency has the least total loss for a particular power switch. For this design example an IRFP450
HEXFET from International Rectifier was chosen because of its low R
and its V
rating. The IRFP450’s
DS(on)
DSS
R
of 0.4 Ω and the maximum V
of 500 V made it an ideal choice. An excellent review of this procedure
DS(on)
DSS
can be found in the Unitrode Power Supply Design Seminar SEM1200, Topic 6, Design Review: 140 W, [Multiple
Output High Density DC/DC Converter].
softstart
The softstart circuitry is used to prevent overshoot of the output voltage during start up. This is accomplished
by bringing up the voltage amplifier’s output (V ) slowly which allows for the PWM duty cycle to increase
VAOUT
slowly. Please use the following equation to select a capacitor for the softstart pin.
In this example t is equal to 7.5 ms, which would yield a C of 10 nF.
DELAY
SS
10 mA t
DELAY
7.5 V
C
+
SS
In an open-loop test circuit, shorting the softstart pin to ground does not ensure 0% duty cycle. This is due to
the current amplifiers input offset voltage, which could force the current amplifier output high or low depending
on the polarity of the offset voltage. However, in the typical application there is sufficient amount of inrush and
bias current to overcome the current amplifier’s offset voltage.
multiplier
The output of the multiplier of the UCC3817 is a signal representing the desired input line current. It is an input
to the current amplifier, which programs the current loop to control the input current to give high power factor
operation. As such, the proper functioning of the multiplier is key to the success of the design. The inputs to the
multiplier are VAOUT, the voltage amplifier error signal, I
, a representation of the input rectified ac line
IAC
voltage, and an input voltage feedforward signal, V
as:
. The output of the multiplier, I
, can be expressed
VFF
MOUT
ǒVVAOUT * 1Ǔ
I
+ I
MOUT
IAC
2
K V
VFF
1
V
where K is a constant typically equal to
.
The electrical characteristics table covers all the required operating conditions for designing with the
multiplier. Additionally, curves in figures 10, 11, and 12 provide typical multiplier characteristics over its entire
operating range.
9
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ꢈ ꢉꢁꢊ ꢋꢌ ꢍꢋꢎ ꢏꢐ ꢑꢒꢁ ꢓꢋ ꢐ ꢍ ꢐꢏ ꢐꢏ ꢔꢀ ꢕꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
multiplier (continued)
The I
signal is obtained through a high-value resistor connected between the rectified ac line and the IAC
IAC
pin of the UCC3817/18. This resistor (R
UCC3817/18 the maximum I
) is sized to give the maximum I
current at high line. For the
IAC
IAC
current is about 500 µA. A higher current than this can drive the multiplier out
IAC
of its linear range. A smaller current level is functional, but noise can become an issue, especially at low input
line. Assuming a universal line operation of 85 V to 265 V gives a R value of 750 kΩ. Because of
RMS
RMS
IAC
voltage rating constraints of standard 1/4-W resistor, use a combination of lower value resistors connected in
series to give the required resistance and distribute the high voltage amongst the resistors. For this design
example two 383-kΩ resistors were used in series.
The current into the IAC pin is mirrored internally to the VFF pin where it is filtered to produce a voltage feed
forward signal proportional to line voltage. The VFF voltage is used to keep the power stage gain constant; and
to provid input power limiting. Please refer to Texas Instruments application note SLUA196 for detailed
explanation on how the VFF pin provides power limiting. The following equation can be used to size the VFF
resistor (R
) to provide power limiting where V
is the minimum RMS input voltage and R
is the total
VFF
IN(min)
IAC
resistance connected between the IAC pin and the rectified line voltage.
1.4 V
R
+
[ 30 kW
VFF
V
0.9
IN(min)
2 R
IAC
Because the VFF voltage is generated from line voltage it needs to be adequately filtered to reduce total
harmonic distortion caused by the 120 Hz rectified line voltage. Refer to Unitrode Power Supply Design
Seminar, SEM−700 Topic 7, [Optimizing the Design of a High Power Factor Preregulator.] A single pole filter
was adequate for this design. Assuming that an allocation of 1.5% total harmonic distortion from this input is
allowed, and that the second harmonic ripple is 66% of the input ac line voltage, the amount of attenuation
required by this filter is:
1.5 %
+ 0.022
66 %
With a ripple frequency (f ) of 120 Hz and an attenuation of 0.022 requires that the pole of the filter (f ) be placed
R
P
at:
f
+ 120 Hz 0.022 [ 2.6 Hz
P
The following equation can be used to select the filter capacitor (C
filter.
) required to produce the desired low pass
VFF
1
C
+
[ 2.2 mF
VFF
2 p R
f
VFF
P
The R
resistor is sized to match the maximum current through the sense resistor to the maximum multiplier
MOUT
current. The maximum multiplier current, or I
, can be determined by the equation:
MOUT(max)
ǒVVAOUT(max) * 1VǓ
I
@V
IAC
IN(min)
I
+
MOUT(max)
2
K V
VFF
(min)
10
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ꢈꢉ ꢁꢊꢋ ꢌ ꢍꢋ ꢎ ꢏꢐ ꢑꢒꢁꢓꢋ ꢐ ꢍꢐ ꢏꢐꢏꢔ ꢀꢕ ꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
multiplier (continued)
I
for this design is approximately 315 µA. The R
resistor can then be determined by:
MOUT(max)
MOUT
V
RSENSE
+
R
MOUT
I
MOUT(max)
In this example V
was selected to give a dynamic operating range of 1.25 V, which gives an R
of
RSENSE
MOUT
roughly 3.91 kΩ.
voltage loop
The second major source of harmonic distortion is the ripple on the output capacitor at the second harmonic
of the line frequency. This ripple is fed back through the error amplifier and appears as a 3rd harmonic ripple
at the input to the multiplier. The voltage loop must be compensated not just for stability but also to attenuate
the contribution of this ripple to the total harmonic distortion of the system. (refer to Figure 2).
C
f
V
OUT
C
R
Z
f
R
IN
−
+
R
D
V
REF
Figure 2. Voltage Amplifier Configuration
The gain of the voltage amplifier, G , can be determined by first calculating the amount of ripple present on
VA
the output capacitor. The peak value of the second harmonic voltage is given by the equation:
P
IN
V
+ ǒ2 p f
OUTǓ
OPK
C
V
R
OUT
In this example V
is equal to 3.91 V. Assuming an allowable contribution of 0.75% (1.5% peak to peak) from
OPK
the voltage loop to the total harmonic distortion budget we set the gain equal to:
ǒDV
Ǔ(
)
0.015
VAOUT
2 V
G
+
VA
OPK
where ∆V
is the effective output voltage range of the error amplifier (5 V for the UCC3817). The network
VAOUT
needed to realize this filter is comprised of an input resistor, R , and feedback components C , C , and R . The
IN
f
Z
f
value of R is already determined because of its function as one half of a resistor divider from V
feeding
IN
OUT
back to the voltage amplifier for output voltage regulation. In this case the value was chosen to be 1 MΩ. This
high value was chosen to reduce power dissipation in the resistor. In practice, the resistor value would be
realized by the use of two 500-kΩ resistors in series because of the voltage rating constraints of most standard
1/4-W resistors. The value of C is determined by the equation:
f
11
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ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
voltage loop (continued)
1
C +
f
ǒ
INǓ
2 p f G
R
R
VA
In this example C equals 150 nF. Resistor R sets the dc gain of the error amplifier and thus determines the
f
f
frequency of the pole of the error amplifier. The location of the pole can be found by setting the gain of the loop
equation to one and solving for the crossover frequency. The frequency, expressed in terms of input power, can
be calculated by the equation:
P
2
IN
f
+
VI
2
ǒ 2 p
(
)
CfǓ
DV
V
R C
IN
OUT
VAOUT
OUT
f
for this converter is 10 Hz. A derivation of this equation can be found in the Unitrode Power Supply Design
VI
Seminar SEM1000, Topic 1, [A 250-kHz, 500-W Power Factor Correction Circuit Employing Zero Voltage
Transitions].
Solving for R becomes:
f
1
R +
f
ǒ
CfǓ
2 p f
VI
or R equals 100 kΩ.
f
Due to the low output impedance of the voltage amplifier, capacitor C was added in series with R to reduce
Z
F
loading on the voltage divider. To ensure the voltage loop crossed over at f , C was selected to add a zero
VI
Z
at a 10th of f . For this design a 2.2-µF capacitor was chosen for C . The following equation can be used to
VI
Z
calculate C .
Z
1
C
+
Z
f
VI
2 p
R
f
10
current loop
The gain of the power stage is:
ǒVOUT SENSEǓ
R
G
(s) +
ID
ǒ
PǓ
s L
V
BOOST
R
has been chosen to give the desired differential voltage for the current sense amplifier at the desired
SENSE
current limit point. In this example, a current limit of 4 A and a reasonable differential voltage to the current amp
of 1 V gives a R value of 0.25 Ω. V in this equation is the voltage swing of the oscillator ramp, 4 V for
SENSE
P
the UCC3817. Setting the crossover frequency of the system to 1/10th of the switching frequency, or 10 kHz,
requires a power stage gain at that frequency of 0.383. In order for the system to have a gain of 1 at the crossover
frequency, the current amplifier needs to have a gain of 1/G at that frequency. G , the current amplifier gain
ID
EA
is then:
1
1
G
+
+
+ 2.611
EA
0.383
G
ID
12
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SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
current loop (continued)
R is the R
resistor, previously calculated to be 3.9 kΩ. (refer to Figure 3). The gain of the current amplifier
I
MOUT
is R /R , so multiplying R by G gives the value of R , in this case approximately 12 kΩ. Setting a zero at the
f
I
I
EA
f
crossover frequency and a pole at half the switching frequency completes the current loop compensation.
1
C
C
+
+
Z
P
2 p R f
f
C
1
f
s
2 p R
f
2
C
P
C
R
Z
f
R
I
−
+
CAOUT
Figure 3. Current Loop Compensation
The UCC3817 current amplifier has the input from the multiplier applied to the inverting input. This change in
architecture from previous Texas Instruments PFC controllers improves noise immunity in the current amplifier.
It also adds a phase inversion into the control loop. The UCC3817 takes advantage of this phase inversion to
implement leading-edge duty cycle modulation. Synchronizing a boost PFC controller to a downstream dc-to-dc
controller reduces the ripple current seen by the bulk capacitor between stages, reducing capacitor size and
cost and reducing EMI. This is explained in greater detail in a following section. The UCC3817 current amplifier
configuration is shown in Figure 4.
L
BOOST
V
OUT
Q
−
+
BOOST
R
SENSE
Z
f
PWM
MULT
CA
COMPARATOR
−
−
+
+
Figure 4. UCC3817 Current Amplifier Configuration
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ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
start up
The UCC3818 version of the device is intended to have VCC connected to a 12-V supply voltage. The UCC3817
has an internal shunt regulator enabling the device to be powered from bootstrap circuitry as shown in the typical
application circuit of Figure 1. The current drawn by the UCC3817 during undervoltage lockout, or start-up
current, is typically 150 µA. Once VCC is above the UVLO threshold, the device is enabled and draws 4 mA
typically. A resistor connected between the rectified ac line voltage and the VCC pin provides current to the shunt
regulator during power up. Once the circuit is operational, the bootstrap winding of the inductor provides the
VCC voltage. Sizing of the start-up resistor is determined by the start-up time requirement of the system design.
DV
Dt
I
+ C
C
(
)
V
0.9
RMS
I
R +
C
Where I is the charge current, C is the total capacitance at the VCC pin, ∆V is the UVLO threshold and ∆t is
C
the allowed start-up time.
Assuming a 1 second allowed start-up time, a 16-V UVLO threshold, and a total VCC capacitance of 100 µF,
a resistor value of 51 kΩ is required at a low line input voltage of 85 V
small as to be ignored in sizing the start-up resistor.
. The IC start-up current is sufficiently
RMS
capacitor ripple reduction
For a power system where the PFC boost converter is followed by a dc-to-dc converter stage, there are benefits
to synchronizing the two converters. In addition to the usual advantages such as noise reduction and stability,
proper synchronization can significantly reduce the ripple currents in the boost circuit’s output capacitor.
Figure 5 helps illustrate the impact of proper synchronization by showing a PFC boost converter together with
the simplified input stage of a forward converter. The capacitor current during a single switching cycle depends
on the status of the switches Q1 and Q2 and is shown in Figure 6. It can be seen that with a synchronization
scheme that maintains conventional trailing-edge modulation on both converters, the capacitor current ripple
is highest. The greatest ripple current cancellation is attained when the overlap of Q1 offtime and Q2 ontime
is maximized. One method of achieving this is to synchronize the turnon of the boost diode (D1) with the turnon
of Q2. This approach implies that the boost converter’s leading edge is pulse width modulated while the forward
converter is modulated with traditional trailing edge PWM. The UCC3817 is designed as a leading edge
modulator with easy synchronization to the downstream converter to facilitate this advantage. Table 1 compares
the I
for D1/Q2 synchronization as offered by UCC3817 vs. the I
for the other extreme of
of 385 V.
CB(rms)
CB(rms)
synchronizing the turnon of Q1 and Q2 for a 200-W power system with a V
BST
UDG-97130-1
Figure 5. Simplified Representation of a 2-Stage PFC Power Supply
14
www.ti.com
ꢀꢁꢁ ꢂ ꢃꢄ ꢅ ꢆ ꢀꢁꢁ ꢂ ꢃꢄ ꢃ ꢆ ꢀꢁꢁ ꢇ ꢃꢄ ꢅ ꢆ ꢀ ꢁꢁ ꢇꢃ ꢄꢃ
ꢈꢉ ꢁꢊꢋ ꢌ ꢍꢋ ꢎ ꢏꢐ ꢑꢒꢁꢓꢋ ꢐ ꢍꢐ ꢏꢐꢏꢔ ꢀꢕ ꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
capacitor ripple reduction (continued)
UDG-97131
Figure 6. Timing Waveforms for Synchronization Scheme
Table 1. Effects of Synchronization on Boost Capacitor Current
V
IN
= 85 V
V
IN
= 120 V
V
IN
= 240 V
D(Q2)
0.35
Q1/Q2
D1/Q2
Q1/Q2
D1/Q2
Q1/Q2
D1/Q2
1.491 A
1.432 A
0.835 A
0.93 A
1.341 A
1.276 A
0.663 A
0.664 A
1.024 A
0.897 A
0.731 A
0.614 A
0.45
Table 1 illustrates that the boost capacitor ripple current can be reduced by about 50% at nominal line and about
30% at high line with the synchronization scheme facilitated by the UCC3817. Figure 7 shows the suggested
technique for synchronizing the UCC3817 to the downstream converter. With this technique, maximum ripple
reduction as shown in Figure 6 is achievable. The output capacitance value can be significantly reduced if its
choice is dictated by ripple current or the capacitor life can be increased as a result. In cost sensitive designs
where holdup time is not critical, this is a significant advantage.
An alternative method of synchronization to achieve the same ripple reduction is possible. In this method, the
turnon of Q1 is synchronized to the turnoff of Q2. While this method yields almost identical ripple reduction and
maintains trailing edge modulation on both converters, the synchronization is much more difficult to achieve and
the circuit can become susceptible to noise as the synchronizing edge itself is being modulated.
15
www.ti.com
ꢀ ꢁꢁꢂ ꢃ ꢄ ꢅ ꢆ ꢀ ꢁꢁ ꢂ ꢃꢄ ꢃ ꢆ ꢀꢁ ꢁꢇ ꢃ ꢄ ꢅ ꢆ ꢀꢁ ꢁꢇ ꢃ ꢄ ꢃ
ꢈ ꢉꢁꢊ ꢋꢌ ꢍꢋꢎ ꢏꢐ ꢑꢒꢁ ꢓꢋ ꢐ ꢍ ꢐꢏ ꢐꢏ ꢔꢀ ꢕꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
capacitor ripple reduction (continued)
Gate Drive
From Down
Stream PWM
C1
D1
UCC3817
CT
D2
C
T
R
T
RT
Figure 7. Synchronizing the UCC3817 to a Down-Stream Converter
REFERENCE VOLTAGE
vs
REFERENCE CURRENT
REFERENCE VOLTAGE
vs
SUPPLY VOLTAGE
7.60
7.55
7.50
7.45
7.40
7.510
7.505
7.500
7.495
7.490
0
5
I
10
15
20
25
9
10
11
12
13
14
− Reference Current − mA
VCC − Supply Voltage − V
VREF
Figure 8
Figure 9
16
www.ti.com
ꢀꢁꢁ ꢂ ꢃꢄ ꢅ ꢆ ꢀꢁꢁ ꢂ ꢃꢄ ꢃ ꢆ ꢀꢁꢁ ꢇ ꢃꢄ ꢅ ꢆ ꢀ ꢁꢁ ꢇꢃ ꢄꢃ
ꢈꢉ ꢁꢊꢋ ꢌ ꢍꢋ ꢎ ꢏꢐ ꢑꢒꢁꢓꢋ ꢐ ꢍꢐ ꢏꢐꢏꢔ ꢀꢕ ꢒꢓꢋ ꢐ
ꢖ
SLUS395J - FEBRUARY 2000 - REVISED MARCH 2009
APPLICATION INFORMATION
MULTIPLIER OUTPUT CURRENT
MULTIPLIER GAIN
vs
vs
VOLTAGE ERROR AMPLIFIER OUTPUT
VOLTAGE ERROR AMPLIFIER OUTPUT
1.5
1.3
1.1
350
300
250
IAC = 150 µA
FF
V
= 1.4 V
IAC = 150
A
µ
200
150
IAC = 300 µA
= 3.0 V
V
FF
0.9
0.7
0.5
IAC = 300
A
µ
IAC = 500
A
µ
100
50
0
IAC = 500 µA
FF
V
= 4.7 V
1.0
2.0
3.0
4.0
5.0
0.0
1.0
2.0
3.0
4.0
5.0
VAOUT − Voltage Error Amplifier Output − V
VAOUT − Voltage Error Amplifier Output − V
Figure 10
Figure 11
RECOMMENDED MINIMUM GATE RESISTANCE
vs
MULTIPLIER CONSTANT POWER PERFORMANCE
500
SUPPLY VOLTAGE
17
16
15
14
13
12
11
10
9
400
VAOUT = 5 V
300
VAOUT = 4 V
200
VAOUT = 3 V
100
VAOUT = 2 V
8
0
10
12
14
16
18
20
0.0
1.0
2.0
3.0
4.0
5.0
VCC − Supply Voltage − V
VFF − Feedforward Voltage − V
Figure 12
Figure 13
17
www.ti.com
PACKAGE OPTION ADDENDUM
www.ti.com
30-Jul-2011
PACKAGING INFORMATION
Status (1)
Eco Plan (2)
MSL Peak Temp (3)
Samples
Orderable Device
Package Type Package
Drawing
Pins
Package Qty
Lead/
Ball Finish
(Requires Login)
UCC2817D
UCC2817DG4
UCC2817DTR
UCC2817DTRG4
UCC2817DW
UCC2817DWG4
UCC2817N
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
PDIP
PDIP
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
PDIP
D
D
16
16
16
16
16
16
16
16
16
16
16
16
16
16
16
16
16
40
40
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-1-260C-UNLIM
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU N / A for Pkg Type
CU NIPDAU N / A for Pkg Type
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU N / A for Pkg Type
D
2500
2500
40
Green (RoHS
& no Sb/Br)
D
Green (RoHS
& no Sb/Br)
DW
DW
N
Green (RoHS
& no Sb/Br)
40
Green (RoHS
& no Sb/Br)
25
Green (RoHS
& no Sb/Br)
UCC2817NG4
UCC2818D
N
25
Green (RoHS
& no Sb/Br)
D
40
Green (RoHS
& no Sb/Br)
UCC2818DG4
UCC2818DTR
UCC2818DTRG4
UCC2818DW
UCC2818DWG4
UCC2818DWTR
UCC2818DWTRG4
UCC2818N
D
40
Green (RoHS
& no Sb/Br)
D
2500
2500
40
Green (RoHS
& no Sb/Br)
D
Green (RoHS
& no Sb/Br)
DW
DW
DW
DW
N
Green (RoHS
& no Sb/Br)
40
Green (RoHS
& no Sb/Br)
2000
2000
25
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
30-Jul-2011
Status (1)
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
ACTIVE
Eco Plan (2)
MSL Peak Temp (3)
Samples
Orderable Device
Package Type Package
Drawing
Pins
Package Qty
Lead/
Ball Finish
(Requires Login)
UCC2818NG4
UCC2818PW
PDIP
TSSOP
TSSOP
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
PDIP
PDIP
SOIC
SOIC
SOIC
SOIC
SOIC
N
PW
PW
D
16
16
16
16
16
16
16
16
16
16
16
16
16
16
16
16
16
16
25
90
Green (RoHS
& no Sb/Br)
CU NIPDAU N / A for Pkg Type
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU N / A for Pkg Type
CU NIPDAU N / A for Pkg Type
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-1-260C-UNLIM
CU NIPDAU Level-2-260C-1 YEAR
UCC2818PWG4
UCC3817D
90
Green (RoHS
& no Sb/Br)
40
Green (RoHS
& no Sb/Br)
UCC3817DG4
UCC3817DTR
UCC3817DTRG4
UCC3817DW
UCC3817DWG4
UCC3817DWTR
UCC3817DWTRG4
UCC3817N
D
40
Green (RoHS
& no Sb/Br)
D
2500
2500
40
Green (RoHS
& no Sb/Br)
D
Green (RoHS
& no Sb/Br)
DW
DW
DW
DW
N
Green (RoHS
& no Sb/Br)
40
Green (RoHS
& no Sb/Br)
2000
2000
25
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
UCC3817NG4
UCC3818D
N
25
Green (RoHS
& no Sb/Br)
D
40
Green (RoHS
& no Sb/Br)
UCC3818DG4
UCC3818DTR
UCC3818DTRG4
UCC3818DW
D
40
Green (RoHS
& no Sb/Br)
D
2500
2500
40
Green (RoHS
& no Sb/Br)
D
Green (RoHS
& no Sb/Br)
DW
Green (RoHS
& no Sb/Br)
Addendum-Page 2
PACKAGE OPTION ADDENDUM
www.ti.com
30-Jul-2011
Status (1)
ACTIVE
ACTIVE
ACTIVE
ACTIVE
Eco Plan (2)
MSL Peak Temp (3)
Samples
Orderable Device
Package Type Package
Drawing
Pins
Package Qty
Lead/
Ball Finish
(Requires Login)
UCC3818DWG4
UCC3818DWTR
UCC3818DWTRG4
UCC3818N
SOIC
SOIC
SOIC
PDIP
DW
DW
DW
N
16
16
16
16
40
2000
2000
25
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-2-260C-1 YEAR
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU N / A for Pkg Type
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
UCC3818N/81511
UCC3818NG4
OBSOLETE
ACTIVE
PDIP
PDIP
N
N
16
16
TBD
Call TI
Call TI
25
90
Green (RoHS
& no Sb/Br)
CU NIPDAU N / A for Pkg Type
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
CU NIPDAU Level-2-260C-1 YEAR
UCC3818PW
UCC3818PWG4
UCC3818PWTR
UCC3818PWTRG4
ACTIVE
ACTIVE
ACTIVE
ACTIVE
TSSOP
TSSOP
TSSOP
TSSOP
PW
PW
PW
PW
16
16
16
16
Green (RoHS
& no Sb/Br)
90
Green (RoHS
& no Sb/Br)
2000
2000
Green (RoHS
& no Sb/Br)
Green (RoHS
& no Sb/Br)
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Addendum-Page 3
PACKAGE OPTION ADDENDUM
www.ti.com
30-Jul-2011
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF UCC2818 :
Automotive: UCC2818-Q1
•
Enhanced Product: UCC2818-EP
•
NOTE: Qualified Version Definitions:
Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
•
Enhanced Product - Supports Defense, Aerospace and Medical Applications
•
Addendum-Page 4
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
B0
K0
P1
W
Pin1
Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
UCC2817DTR
UCC2818DTR
UCC2818DWTR
UCC3817DTR
UCC3817DWTR
UCC3818DTR
UCC3818DWTR
UCC3818PWTR
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
TSSOP
D
D
16
16
16
16
16
16
16
16
2500
2500
2000
2500
2000
2500
2000
2000
330.0
330.0
330.0
330.0
330.0
330.0
330.0
330.0
16.4
16.4
16.4
16.4
16.4
16.4
16.4
12.4
6.5
6.5
10.3
10.3
2.1
2.1
2.7
2.1
2.7
2.1
2.7
1.6
8.0
8.0
16.0
16.0
16.0
16.0
16.0
16.0
16.0
12.0
Q1
Q1
Q1
Q1
Q1
Q1
Q1
Q1
DW
D
10.75 10.7
6.5 10.3
10.75 10.7
6.5 10.3
10.75 10.7
6.9 5.6
12.0
8.0
DW
D
12.0
8.0
DW
PW
12.0
8.0
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
14-Jul-2012
*All dimensions are nominal
Device
Package Type Package Drawing Pins
SPQ
Length (mm) Width (mm) Height (mm)
UCC2817DTR
UCC2818DTR
UCC2818DWTR
UCC3817DTR
UCC3817DWTR
UCC3818DTR
UCC3818DWTR
UCC3818PWTR
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
SOIC
TSSOP
D
D
16
16
16
16
16
16
16
16
2500
2500
2000
2500
2000
2500
2000
2000
333.2
333.2
367.0
333.2
367.0
333.2
367.0
367.0
345.9
345.9
367.0
345.9
367.0
345.9
367.0
367.0
28.6
28.6
38.0
28.6
38.0
28.6
38.0
35.0
DW
D
DW
D
DW
PW
Pack Materials-Page 2
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UCC3818DWG4 UCC2817D UCC2817DG4 UCC2817DTR UCC2817DTRG4 UCC2817DW UCC2817DWG4
UCC2817N UCC2817NG4 UCC2818D UCC2818DG4 UCC2818DTR UCC2818DTRG4 UCC2818DW
UCC2818DWG4 UCC2818DWTR UCC2818DWTRG4 UCC2818N UCC2818NG4 UCC2818PW UCC2818PWG4
UCC3817D UCC3817DG4 UCC3817DTR UCC3817DTRG4 UCC3817DW UCC3817DWG4 UCC3817DWTR
UCC3817N UCC3818D UCC3818DTR UCC3818DTRG4 UCC3818DW UCC3818DWTR UCC3818DWTRG4
UCC3818N UCC3818NG4 UCC3818PW UCC3818PWG4 UCC3818PWTR UCC3818PWTRG4 UCC3817NG4
UCC3817DWTRG4 UCC3818DG4
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