UCC27222PWPR中文资料_数据手册_规格书

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

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FEATURES

APPLICATIONS

ꢀ

Maximizes Efficiency by Minimizing

Body-Diode Conduction and Reverse

Recovery Losses

ꢀ

Non-Isolated Single or Multi-phased

DC-to-DC Converters for Processor Power,

General Computer, Telecom and Datacom

Applications

ꢀ

Transparent Synchronous Buck Gate Drive

Operation From the Single Ended PWM Input

Signal

DESCRIPTION

ꢀ

12-V or 5-V Input Operation

The UCC27221 and UCC27222 are high-speed

synchronous buck drivers for today’s

high-efficiency, lower-output voltage designs.

Using Predictive Gate Driveꢁ (PGD) control

technology, these drivers reduce diode

conduction and reverse recovery losses in the

3.3-V Input Operation With Availability of

12-V Bus Bias

ꢀ

On-Board 6.5-V Gate Drive Regulator

3.3-A TrueDrive Gate Drives for High

Current Delivery at MOSFET Miller

Thresholds

synchronous

rectifier

MOSFET(s).

The

UCC27221 has an inverted PWM input while the

UCC27222 has a non-inverting PWM input.

ꢀ

Automatically Adjusts for Changing

Operating Conditions

Predictive Gate Driveꢁ technology uses control

loops which are stabilized internally and are

therefore transparent to the user. These loops use

no external components, so no additional design

is needed to take advantage of the higher

efficiency of these drivers.

Thermally Enhanced 14-Pin PowerPAD

HTSSOP Package Minimizes Board Area and

Junction Temperature Rise

FUNCTIONAL APPLICATION DIAGRAM

This closed loop feedback system detects

body-diode conduction, and adjusts deadtime

delays to minimize the conduction time interval.

This virtually eliminates body-diode conduction

while adjusting for temperature, load- dependent

delays, and for different MOSFETs. Precise gate

timing at the nanosecond level reduces the

reverse recovery time of the synchronous rectifier

MOSFET body-diode, reducing reverse recovery

losses seen in the main (high-side) MOSFET. The

lower junction temperature in the low-side

MOSFET increases product reliability. Since the

power dissipation is minimized, a higher switching

frequency can also be used, allowing for smaller

component sizes.

V

IN

UCC27222

IN VHI

6,8 GND G1 13

7

14

PWM

IN

V

OUT

3

VDD

SW 11,12

G2 9,10

4,5 VLO

The UCC27221 and UCC27222 are offered in the

GND

thermally enhanced 14-pin PowerPADꢁ package

GND

OUT

IN

Note: 12-V input system shown. For 5-V input only systems, see Figure 6.

with 2°C/W θ .

jc

Predictive Gate Driveꢁ and PowerPADꢁ are trademarks of Texas Instruments Incorporated.

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Copyright  2002, Texas Instruments Incorporated

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ꢞ ꢢ ꢟ ꢞꢗ ꢘꢬ ꢚꢙ ꢝ ꢥꢥ ꢣꢝ ꢛ ꢝ ꢜ ꢢ ꢞ ꢢ ꢛ ꢟ ꢧ

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1

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

PWP PACKAGE

(TOP VIEW)

1

2

3

4

5

6

7

14

13

12

11

10

9

N/C

VDD

VLO

PVLO

AGND

IN

VHI

G1

SW

SWS

G2S

G2

8

PGND

N/C − No internal connection

AVAILABLE OPTIONS

PACKAGED DEVICES

PowerPAD

HTSSOP−14 (PWP)

UCC27221PWP

UCC27222PWP

PWM INPUT

(IN)

T

ꢁ

A

INVERTING

−40ꢂ C to 105ꢂ C

NON-INVERTING

ꢃThe PWP package is available taped and reeled. Add R suffix to device type

(e.g. UCC27221PWPR) to order quantities of 2,000 devices per reel and 90

units per tube.

†}

absolute maximum ratings over operating free-air temperature (unless otherwise noted)

Supply voltage range, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.3 to 20 V

Input voltage, VHI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 V

SW, SWS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 V

Supply current, I

including gate drive current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .100 mA

DD,

Sink current (peak) pulsed, G1/G2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.0 A

Source current (peak) pulsed, G1/G2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −4.0 A

Analog input, IN . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −3.0 V to V

+ 0.3 V, not to exceed 15 V

DD

Power Dissipation at T = 25°C (PWP package) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 W

A

Operating junction temperature range, T . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −55°C to 115°C

J

Storage temperature range, T . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 150°C

stg

Lead temperature soldering 1.6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . 300°C

†

‡

Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and

functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not

implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

All voltages are with respect to AGND and PGND. Currents are positive into, negative out of the specified terminal.

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

ELECTRICAL CHARACTERISTICS

V

DD

= 12-V, 1-µF capacitor from VDD to GND, 1-µF capacitor from VHI to SW, 0.1-µF and 2.2-µF capacitor from PVLO to PGND, PVLO tied to

VLO, T = −40ꢂ C to 105ꢂ C for the UCC2722x, T = T (unless otherwise noted)

A

J

VLO regulator

PARAMETER

TEST CONDITIONS

MIN

6.2

6.1

TYP

6.5

2

MAX

6.8

6.9

10

UNIT

V

= 12 V,

I

= 0 mA

DD

VLO

= 20 V,

= 0 mA

Regulator output voltage

V

DD

= 8.5 V,

= 100 mA

DD

Line Regulation

Load Regulation

= 12 V to 20 V

= 0 mA to 100 mA

= 8.5 V

DD

mV

I

15

40

VLO

(1)

Short-circuit current

V

DD

220

7.8

mA

V

Dropout voltage, (VDD at 5% VLO drop)

VLO = 6.175 V,

I

= 100 mA

7.1

8.5

VLO

undervoltage lockout

PARAMETER

TEST CONDITIONS

Measured at VLO

MIN

3.30

3.15

0.07

TYP

3.82

3.70

0.12

MAX

4.40

4.25

0.20

UNIT

Start threshold voltage

Minimum operating voltage after start

Hysteresis

V

bias currents

PARAMETER

TEST CONDITIONS

MIN

3.6

5.5

TYP

4.7

7.1

10

MAX

5.8

8.5

20

UNIT

V

bias current at VLO (ON), 5 V applications only

VLO = 4.5 V,

VDD = 8.5 V

VDD = no connect

LO

mA

bias current

DD

f

IN

= 500 kHz,

No load on G1/G2

input command (IN)

PARAMETER

TEST CONDITIONS

MIN

3.3

TYP

3.6

MAX

3.9

2.8

1

UNIT

High-level input voltage

Low-level input voltage

Input bias current

10 V < VDD < 20 V

V

2.2

2.5

V

DD

= 15 V

µA

input (SWS)

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNIT

f

= 500 kHz,

t

maximum,

minimum,

minimum

IN

G2S = 0.0 V

ON, G2

ON, G1

High-level input threshold voltage

1.4

2.0

2.6

V

f

= 500 kHz,

IN

G2S = 0.0 V

0.7

1.0

1.3

Low-level input threshold voltage

Input bias current

f

= 500 kHz,

−100 −300 −500

mV

mA

IN

SWS = 0.0 V

−0.9

−1.2

−1.5

input (G2S)

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNIT

f

= 500 kHz,

t

maximum,

minimum,

IN

SWS = 0.0 V

ON, G2

High-level input voltage

Low-level input voltage

1.4

2.0

2.6

V

f

IN

= 500 kHz,

t

ON, G2

0.7

1.0

1.3

SWS = 0.0 V

Input bias current

G2S = 0 V

−370 −470 −570

µA

NOTE 1: Ensured by design. Not production tested.

3

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

ELECTRICAL CHARACTERISTICS

V

DD

= 12-V, 1-µF capacitor from VDD to GND, 1-µF capacitor from VHI to SW, 0.1-µF and 2.2-µF capacitor from PVLO to PGND, PVLO tied to

VLO, T = −40ꢂ C to 105ꢂ C for the UCC2722x, T = T (unless otherwise noted)

A

J

G1 main output

PARAMETER

TEST CONDITIONS

MIN

0.3

10

−3

3

TYP

0.9

25

MAX

1.5

UNIT

Sink resistance

SW = 0 V,

VHI = 6 V, IN = 0 V,

G1 = 0.5 V

Ω

(2)

Source resistance

VHI = 6 V, IN = 6.5 V, G1 = 5.5 V

VHI = 6 V, IN = 6.5 V, G1 = 3.0 V

45

(1)(2)

Source current

−3.3

3.3

17

A

(1)(2)

Sink current

Rise time

Fall time

VHI = 6 V, IN = 0 V,

G1 = 3.0 V

C = 2.2 nF from G1 to SW,

V

= 20 V

25

DD

ns

V

17

G2 SR output

PARAMETER

(2)

TEST CONDITIONS

PVLO = 6.5 V, IN = 6.5 V, G1 = 0.25 V

MIN

5

TYP

15

MAX

30

UNIT

Sink resistance

Source resistance

(1)(2)

Ω

(2)

PVLO = 6.5 V, IN = 0 V,

PVLO = 6.5 V, IN = 0 V

PVLO = 6.5 V, IN = 6.5 V

G2 = 6.0 V

G2 = 3.25 V

10

−3

3

20

35

Source current

(1)(2)

−3.3

3.3

17

A

Sink current

(2)

Rise time

C = 2.2 nF from G2 to PGND V

= 20 V

25

35

DD

ns

Fall time

20

DD

deadtime delay

PARAMETER

TEST CONDITIONS

MIN

60

TYP

80

MAX

100

110

4.7

UNIT

t

, IN to G2 falling

, IN to G1 falling

OFF, G2

55

80

OFF, G1

Delay Step Resolution

3.5

4.1

−15

48

t

minimum

maximum

minimum

maximum

ns

ON, G1

ON, G2

−21

38

,

ON G2

NOTE 1: Ensured by design. Not production tested.

2: The pullup / pulldown circuits of the drivers are bipolar and MOSFET transistors in parallel. The peak output current rating is the

combined current from the bipolar and MOSFET transistors. The output resistance is the RDS(ON) of the MOSFET transistor when the

voltage on the driver output is less than the saturation voltage of the bipolar transistor.

t

OFF,G1

3.25 V

UCC27222

IN

t

ON,G2

OFF,G2

ON,G1

90%

10%

G1

G2

90%

10%

UDG−01042

Figure 1. Predictive Gate Drive Timing Diagram

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

TERMINAL FUNCTIONS

TERMINAL

NAME NO.

DESCRIPTION

I/O

Analog ground for all internal logic circuitry. AGND and PGND should be tied to the PCB ground plane

with vias.

AGND

6

−

G1

G2

13

9

O

High-side gate driver output that swings between SW and VHI.

Low-side gate driver output that swings between PGND and PVLO.

Used by the predictive deadtime controller for sensing the SR MOSFET gate voltage to set the

appropriate deadtime.

G2S

IN

10

7

I

Digital input command pin. A logic high forces on the main switch and forces off the synchronous

rectifier.

PGND

PVLO

SW

8

5

−

I

Ground return for the G2 driver. Connect PGND to PCB ground plane with several vias.

PVLO supplies the G2 driver. Connect PVLO to VLO and bypass on the PCB.

G1 driver return connection.

12

−

Used by the predictive controller to sense SR body-diode conduction. Connect to SR MOSFET drain

close to the MOSFET package.

SWS

VDD

VHI

11

3

I

Input to the internal VLO regulator. Nominal VDD range is from 8.5 V to 20 V. Bypass with at least

0.1 µF of capacitance.

Floating G1 driver supply pin. VHI is fed by an external Schottky diode during the SR MOSFET on-time.

Bypass VHI to SW with an external capacitor.

14

4

I

Output of the VLO regulator and supply input for the logic and control circuitry. Connect VLO to PVLO and

bypass on the PCB.

VLO

O

SIMPLIFIED BLOCK DIAGRAM

N/C

1

2

14 VHI

13 G1

12 SW

VLO

REGULATOR

VDD

VLO

3

4

11 SWS

10 G2S

PREDICTIVE

DELAY

CONTROLLER

+

3.82 V/ 3.7 V

PVLO

UVLO

PVLO

AGND

5

6

PVLO

UCC27221

UCC27222

9

8

G2

IN

7

PGND

UDG−01030

5

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

predictive gate drive technique

The Predictive Gate Driveꢁ technology utilizes a digital feedback system to detect body-diode conduction, and

then adjusts the deadtime delays to minimize it. This system virtually eliminates the body-diode conduction time

intervals for the synchronous MOSFET, while adjusting for different MOSFETs characteristics, propagation and

load dependent delays. Maximum power stage efficiency is the end result.

Two internal feedback loops in the predictive delay controller continuously adjusts the turn on delays for the two

MOSFET gate drives G1 and G2. As shown in Figure 2, t

body-diode conduction in the synchronous rectifier MOSFET Q . The turn-off delay for both G1 and G2, t

and t

are varied to provide minimum

ON,G1

ON,G2

2

OFF,G1

and t

are fixed by propagation delays internal to the device.

OFF,G2

The predictive delay controller is implemented using a digital control technique, and the time delays are

therefore discrete. The turn-on delays, t and t , are changed by a single step (typically 3 ns) every

ON, G1

ON, G2

switching cycle. The minimum and maximum turn-on delays for G1 and G2 are specified in the electrical

characteristics table.

UCC27221

3.25 V

t

OFF,G1

IN

3.25 V

UCC37222

t

ON,G2

OFF,G2

ON,G1

90%

10%

G1

G2

90%

10%

Figure 2. Predictive Gate Drive Timing Diagram

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

A typical application circuit for systems with 8.5-V to 20-V input is shown in Figure 3.

VIN

D1

R1

UCC27222

N/C

VHI

G1

Q1

L1

C1

SW

SWS

G2S

G2

VDD

C

V

IN

OUT

VLO

PVLO

AGND

Q2

C2

Cout

PWM

Input

PGND

IN

GND

Figure 3. System Application: 8.5-V to 20-V Input

selection of VHI series resistor R1 (dV/dt Considerations):

The series resistor R1 may be needed to slowdown the turn-on of the main forward switch to limit the dV/dt which

can inadvertently turn on the synchronous rectifier switch. In nominal 12-V input designs, a R1 value of 4-Ω to

10-Ω can be used depending on the type of MOSFET used and the high-side/low-side MOSFET ratio. In 5-V

or lower input applications however, R1 is not needed.

When the drain-source voltage of a MOSFET quickly rises, inadvertent dV/dt induced turn-on of the device is

possible. This can especially be a problem for input voltages of 12 V or greater. As Q1 rapidly turns on, the

drain-to-source voltage of Q2 rises sharply, resulting in a dV/dt voltage spike appearing on the gate signal of

Q2. If the dV/dt induced voltage spike were to exceed the given threshold voltage, the MOSFET may briefly

turn on when it should otherwise be commanded off. Obviously this undesired event would have a negative

impact on overall efficiency.

Minimizing the dV/dt effect on Q2 can be accomplished by proper MOSFET selection and careful layout

techniques. The details of how to select a MOSFET to minimize dV/dt susceptibility are outlined in SEM−1400,

Topic 2, Appendix A, Section A5. Secondly, the switch node connecting Q1, Q2 and L1 should be laid out as

tight as possible, minimizing any parasitic inductance, which might worsen the dV/dt problem.

If the dV/dt induced voltage spike is still present on the gate Q2, a 4W to 10W value of R1 is recommended to

minimize the possibility of inadvertently turning on Q2. The addition of R1 slows the turn-on of Q1, limiting the

dV/dt rate appearing on the drain-to-source of Q2. Slowing down the turn-on of Q1 will result in slightly higher

switching loss for that device only, but the efficiency gained by preventing dV/dt turn-on of Q2 will far outweigh

the negligible effect of adding R1.

When Q2 is optimally selected for dV/dt robustness and careful attention is paid to the PCB layout of the switch

node, R1 may not be needed at all, and can therefore be replaced with a 0-Ω jumper to maintain high efficiency.

The goal of the designer should not be to completely eliminate the dV/dt turn-on spike but to assure that the

maximum amplitude is less than the MOSFET gate-to-source turn-on threshold voltage under all operating

conditions.

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

selection of bypass capacitor C1

Bypass capacitors should be selected based upon allowable ripple voltage, usually expressed as a percent of

the regulated power supply rail to be bypassed. In all of the UCC27222 application circuits shown herein, C1

provides the bypass for the main (high-side) gate driver. Every time Q1 is switched on, a packet of charge is

removed from C1 to charge Q1’s gate to approximately 6.0 V. The charge delivered to the gate of Q1 can be

found in the manufacturer’s datasheet curves. An example of a gate charge curve is shown in Figure 4.

GATE-TO-SOURCE VOLTAGE

vs

TOTAL GATE CHARGE

8

6

4

2

31 nC

0

10

20

30

40

Q6 − Total Gate Charge − nC

Figure 4.

As shown in Figure 4, 31 nC of gate charge is required in order for Q1’s gate to be charged to 6.0 V, relative

to its source. The minimum bypass capacitor value can be found using the following calculation:

Q

G

C1

+

MIN

_kǒ_{VHI * V}_SWǓ

(1)

where k is the percent ripple on C1, Q is the total gate charge required to drive the gate of Q1 from zero to

G

the final value of (VHI−VSW). In this example gate charge curve, the value of the quantity (VHI−VSW) is taken

to be 6.0 V. This value represents the nominal VLO regulator output voltage minus the forward voltage drop of

the external Schottky diode, D1. For the MOSFET with the gate charge described in Figure 4, the minimum

capacitance required to maintain a 3% peak-to-peak ripple voltage can be calculated to be 172 nF, so a 180-nF

or a 220-nF capacitor could be used. The maximum peak-to-peak C1 ripple must be kept below 0.4 V for proper

operation.

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APPLICATION INFORMATION

selection of MOSFETs

The peak current rating of a driver imposes a limit on the maximum gate charge of the external power MOSFET

driven by it. The limit is based on the amount of time needed to deliver or remove the required charge to achieve

the desired switching speed during turn-on and turn-off of the external transistor. Hence, there are the families

of gate driver circuits with different current ratings.

To demonstrate this, assume a constant time interval for the switching transition and a fixed gate drive

amplitude. A larger MOSFET with more gate charge will require higher current capability from the driver to

turn-on or turn-off the device in the same amount of time. Accordingly, there is a practical upper limit on gate

charge which can be driven by the UCC27222 family of drivers. Considering the current capability of the

TrueDrive output stage and the available dynamic range (delay adjust range) of the Predictive Gate Drive

circuitry, this limit is approximately 120 nC of gate charge.

Some higher current applications require several MOSFETs to be connected parallel and driven by the same

gate drive signal. If their combined gate charge exceeds 120 nC, the rise and fall times of the gate drive signals

will extend and limit the delay adjust range of the PGD circuit in the UCC27222. This may limit the benefits of

the PGD technology under certain operating conditions.

Note that there are additional considerations in the gate drive circuit design which influence the maximum gate

charge of the external MOSFETs. The most significant of these is the operating frequency which, together with

the amount of gate charge, will define the power dissipation in the driver. The allowable power dissipation is a

function of the maximum junction and operating temperatures, thermal and reliability considerations.

selection of bypass capacitor C2

C2 supplies the peak current required to turn on the Q2 synchronous rectifier MOSFET, as well as the peak

current to charge the C1 capacitor through the bootstrap diode. Since the synchronous MOSFET is turned on

with 0 V across its drain-to-source, there is no Miller, or gate-to-drain charge. Therefore the synchronous

MOSFET gate can be modeled as a simple linear capacitance. The value of this capacitance can be found from

the datasheet’s gate charge curve. Referring to Figure 5, the slope of the curve past the Miller plateau indicates

the equivalent gate capacitance. Because the Y-axis is described in volts, the capacitance is actually the inverse

of the slope of the curve. For example, the curve in Figure 4 has a slope of approximately 2 V / 12 nC over the

gate charge range of 10 nC to 40 nC. The equivalent capacitance is 12 nC / 2 V = 6 nF. With the equivalent

capacitance, the minimum bypass capacitor value can be calculated as:

C

EQ

C2

+

MIN

k

(2)

where

ꢀ

C

is the equivalent gate capacitance,

EQ

k is the voltage ripple on C2, expressed as a percentage

For a peak-to-peak ripple of 3%, the minimum C2 capacitor value is calculated to be 200 nF. A 220-nF capacitor

would be used in this case.

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

regulator current and power dissipation

The regulator current can be calculated from the dc or average current required by the two gate drivers. This

current can be expressed as:

ǒ_C

_GǓ

I

+ F

VLO ) Q

EQ

REG

SW

(3)

Assuming all the power dissipation is internal to the device, and the internal bias current is negligible, the power

dissipated by the device is:

ǒ_C

Ǔ

P

+ F

VLO ) Q VDD

DIS

SW

EQ G

(4)

For a 500-kHz design, using MOSFETs with the gate charge characteristics shown in Figure 4 for both Q1 and

Q2, the average regulator current would be 35 mA, and, when operated from a 12-V input rail, the resulting

power dissipation is calculated to be 420 mW.

systems using 3.3-V or 5-V power input and 12-V gate drive

Figure 5 shows a schematic for systems where the power bus input is 5 V and 12 V is available for powering

the gate drives. This system provides the 6.5-V gate drive to both MOSFETs, while the power stage operates

off the 3.3-V or 5-V bus.

+3.3 V

or +5V

D1

R1

UCC27222

N/C

VHI

G1

C

Q1

IN

L1

C1

+12 V

VDD

VLO

SW

SWS

G2S

G2

V

OUT

C3

PVLO

Q2

C2

AGND

IN

Cout

PWM

Input

PGND

GND

Figure 5. System Application: 3.3-V or 5-V Power Input with 12 V Available for Gate Drive

Note that the series resistor R1 may be needed to slowdown the turn-on of the main forward switch to limit the

dV/dt which can inadvertently turn on the synchronous rectifier switch. The dV/dt considerations and the

selection of R1 are discussed in the previous section.

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

systems with 5-V input only

The circuit pictured in Figure 6 starts up from a 5-V input bus and provides a 6.5-V gate drive to the power

MOSFETs. This circuit uses a charge pump consisting of D , D and C to effectively double the input voltage

3

4

3

and apply this to the input of the linear regulator. The regulator then regulates the doubled input voltage to the

6.5-V nominal for VLO.

D3

D4

+5V

D1

C3

D2

C2

UCC27222

N/C

VHI

G1

C

Q1

IN

L1

C1

VDD

VLO

SW

SWS

G2S

G2

V

OUT

C4

PVLO

AGND

Q2

Cout

PWM

Input

IN

PGND

GND

Figure 6. System Application: 5-V-Only Power Input with 6.5-V Gate Drive Using Charge Pump Circuit

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

selecting D , D , and D

4

2

3

Selection of suitable diodes is based upon the conducted peak and average currents. D2 simply provides a path

to charge C2 at converter power-up. Virtually any one of the common BAT54 series of Schottky diodes can be

used. To select D3 and D4, the peak currents of these two diodes need to be taken into account. First, the

average current flowing in both D3 and D4 is the same as the regulator current described in equation (3). The

peak currents in D3 and D4 are described as:

I

REG

I

+

D3PK

1 * D

(5)

(6)

I

REG

D4PK

D

For most UCC27222 applications, the duty cycle is much less than 50%, and the peak current in D3 is quite

reasonable. However, the peak current in D4 is quite high. This high peak current requires using a diode with

a higher current rating for D4.

To maintain a reasonable charge pump efficiency, BAT54-type diodes can be used for applications where the

peak currents are below approximately 40 mA. For applications where the peak current is greater than 40 mA,

a 350-mA or 500-mA diode should be used. A typical 350-mA diode is SD103CW, SOD−123 package,

manufactured by Diodes Inc. A typical 500-mA diode is the ZHCS500, SOT−23 package, available from Zetex

Inc.

selection of the flying capacitor C3

The flying capacitor is subjected to large peak currents, and to keep the peak-to-peak ripple voltage low, this

capacitor has to be larger than C1 and C2. Selection of C3 should be done based on allowable peak-to-peak

ripple on C3:

I

REG

C3

+

MIN

_kǒ_{VIN * V}_FD3Ǔ

F

SW

(7)

where I

is the regulator output current, F

is the switching frequency, k is the percent ripple on C3, and

REG

SW

V

is the forward drop of D3.

FD2

selection of bypass capacitor C4

The bypass capacitor C4 needs to be sized to take the peak current from the charge pump diode D4. The

capacitor is sized based on allowable ripple voltage:

(

1 * D)

I

REG

C

+

MIN

_kǒ_{2 VIN * V}

_FD4Ǔ

* V

F

FD3

SW

and V

(8)

where V

are the forward voltages of D3 and D4 and k is the percent ripple allowed on C4.

FD4

FD3

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

synchronous rectification and predictive delay

In a normal buck converter, when the main switch turns off, current is flowing to the load in the inductor. This

current cannot be stopped immediately without using infinite voltage. For the current path to flow and maintain

voltage levels at a safe level, a rectifier or catch device is used. This device can be either a conventional diode,

or it can be a controlled active device if a control signal is available to drive it. The UCC27222 provides a signal

to drive an N-channel MOSFET as a rectifier. This control signal is carefully coordinated with the drive signal

for the main switch so that there is minimum delay from the time that the rectifier MOSFET turns off and the main

switch turns on, and minimum delay from when the main switch turns off and the rectifier MOSFET turns on.

This scheme, Predictive Gate Driveꢁ delay, uses information from the current switching cycle to adjust the

delays that are to be used in the next cycle. Figure 7 shows the switch-node voltage waveform for a

synchronously rectified buck converter. Illustrated are the relative effects of a fixed-delay drive scheme

(constant, pre-set delays for the turnoff to turn on intervals), an adaptive delay drive scheme (variable delays

based upon voltages sensed on the current switching cycle) and the predictive delay drive scheme.

Note that the longer the time spent in body-diode conduction during the rectifier conduction period, the lower

the efficiency. Also, not described in Figure 7 is the fact that the predictive delay circuit can prevent the body

diode from becoming forward biased at all while at the same time avoiding cross conduction or shoot through.

This results in a significant power savings when the main MOSFET turns on, and minimizes reverse recovery

loss in the body diode of the rectifier MOSFET.

The power dissipation on the main (forward) MOSFET is reduced as well, although that savings is not as

significant as the savings in the rectifier MOSFET.

During reverse recovery the body diode is still forward biased, thus the reverse recovery current goes through

the forward MOSFET while the drain−source voltage is still high, causing additional switching losses. Without

PGD during this switching transition, Vds = Vin and Ids = Iload + Irr in the main MOSFET. With PGD however,

Vds = Vin and Ids = Iload. The reduction in current accounts for additional power savings in the main MOSFET.

V

IN

0 V

V

D

GND

Channel Conduction

Body Diode Conduction

Fixed Delay

Adaptive Delay

Predictive Delay

UDG−02175

Figure 7. Switch Node Waveforms for Synchronous Buck Converter

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

comparison between predictive and adaptive gate drive techniques

The first synchronous rectifier controllers had a fixed turn-on delay between the two gate drivers. The advantage

of this well-known technique is its simplicity. The drawbacks include the need to make the delay times long

enough to cover the entire application of the device and the temperature and lot-to-lot variation of the time delay.

Since the body-diode of the synchronous rectifier conducts during this deadtime, the efficiency of this technique

varies with different MOSFETs, ambient temperature, and with the lot-to-lot variation of the deadtime delay.

To combat the variability of the internal time delays, second generation controllers used state information from

the power stage to control the turn-on of the two gate drivers. This technique is usually referred to as adaptive

gate drive technique and is pictured in FIgure 8.

V

IN

+

ON ON

OFF

V

OUT

UDG−01031

Figure 8. Adaptive Gate Drive Technique

The main advantage of the adaptive technique is the on-the-fly delay adjustment for different MOSFETs and

temperature-variable time delays. The disadvantages include the body-diode conduction time intervals caused

by delays in the cross-coupling loops and the inability to compensate for the delay to charge the MOSFET gates

to the threshold levels. Additionally, it is difficult to determine whether the synchronous MOSFET channel is off

by solely monitoring the SR MOSFET gate voltage. Some devices actually add a programmable delay between

the turn-off of the synchronous rectifier and the turn-on of the main MOSFET via an external capacitor. This

added delay directly affects the power stage efficiency through additional body-diode conduction losses. Since

these losses are centralized in the synchronous MOSFET, the stress and temperature rise in this component

becomes a major design headache.

The third-generation predictive control technique is different from the adaptive technique in that it uses

information from the previous switching cycle to set the deadtime for the current cycle. The adaptive technique

on the other hand uses the current state information to set the delay times. The inherent feedback loop

propagation delays cause body-diode conduction.

14

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

adaptive vs. predictive waveforms

Figures 9 through 11 illustrate the adaptive (left) vs. predictive (right) switching waveforms. Key comparison

regions are denoted with (A), (B), (C), (D), and (E) for the adaptive control waveforms and (A′), (B′), (C′), (D′),

and (E′) for the predictive control waveforms. Figures 10 and 11 are close-ups of each transition edge.

At (A), the propagation delay from sensing the synchronous rectifier gate going low to the high-side gate going

high results in approximately 60 ns of body-diode conduction shown at (B). With the predictive drive, as soon

as the body-diode conduction of the SR MOSFET (B) is sensed, the high-side turn-on delay is adjusted to

minimize the body-diode conduction time (B′).

At (A′), the high side gate-to-source voltage is increasing while the synchronous rectifier gate-to-source voltage

is decreasing. A natural result of the precise timing of the high-side MOSFET turn-on is shown at (C) and (C′).

The overshoot and ringing for the predictive drive (C′) has much smaller amplitude than the adaptive drive (C)

due a reduction in reverse recovery in the SR MOSFET body diode. This reduction in reverse recovery is only

possible with the extremely precise gate timing used in the predictive drive technique.

At (D), the propagation delay from the synchronous rectifier drain-to-source voltage falling to the gate-to-source

voltage rising causes the body diode of the SR MOSFET to conduct for approximately 60 ns (E). When the

predictive drive is enabled (D′), the inherent delay is eliminated and virtually no body-diode conduction is shown

at (E′).

Complementary

Gate Drive

Waveforms

2 V / div

D

A

A4

D4

C

C4

VDS of SR

MOSFET

Switch

E4

B4

E

B

2 V / div

Predictive Drive

100 ns / div

Adaptive Drive

100 ns / div

Figure 9. Adaptive vs. Predictive Switching Waveforms

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

A4

Complementary

Gate Drive

A

Waveforms

2 V / div

C

C4

VDS of SR

MOSFET

Switch

B4

B

2 V / div

Adaptive Drive

20 ns / div

Predictive Drive

20 ns / div

Figure 10. Close-Up: Turn-Off of Synchronous Rectifier Switch to Turn-On of Main Switch

Complementary

Gate Drive

Waveforms

D4

D

2 V / div

VDS of SR

MOSFET

Switch

E4

E

2 V / div

Adaptive Drive

20 ns / div

Predictive Drive

20 ns / div

Figure 11. Close-Up: Turn-Off of Main Switch to Turn-On of Synchronous Rectifier Switch

efficiency comparison

Figures 12 through 15 show a series of efficiency measurements taken at two output voltages (0.9 V and 1.8

V) and two switching frequencies (250 kHz and 500 kHz) for both predictive and adaptive delay techniques.

The efficiency gain using the predictive technique is 1% for a V

of 250 kHz (Figure 12). Figures 13 and 14 show the efficiency gain approximately doubles when V

level of 1.8 V and at a switching frequency

OUT

is lowered

OUT

by a factor of two (to 0.9 V), or when the switching frequency is doubled to 500 kHz. With both doubled frequency

and one-half of the output voltage, the efficiency gain of predictive technology is about 4% over the adaptive

technology (Figure 15). Therefore, as the switching frequency increases and output voltages are lowered, the

efficiency gains are higher. This results in lower operational temperatures for increased reliability as well as

smaller size designs for increased frequencies.

16

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

APPLICATION INFORMATION

EFFICIENCY

vs

EFFICIENCY

vs

OUTPUT CURRENT

96

94

92

90

88

86

84

82

80

0.5

96

0.5

V

= 5 V

= 0.9 V

= 250 kHz

IN

OUT

94

92

90

88

86

84

82

80

PREDICTIVE

f

SW

0.4

0.3

PREDICTIVE

0.4

0.3

ADAPTIVE

0.2

0.1

0.0

0.2

0.1

0.0

DELTA POWER

DISSIPATION

78

76

78

76

DELTA POWER

DISSIPATION

V

= 5 V

= 1.8 V

= 250 kHz

IN

OUT

f

SW

74

0

5

10

15

20

0

5

10

15

20

I

− Output Current − A

I

− Output Current − A

OUT

Figure 12

Figure 13

EFFICIENCY

vs

EFFICIENCY

vs

OUTPUT CURRENT

96

94

92

90

88

86

84

82

80

1.2

1.0

96

94

92

90

88

86

84

82

80

1.2

1.0

V

= 5 V

IN

= 0.9 V

= 500 kHz

OUT

PREDICTIVE

f

SW

PREDICTIVE

0.8

0.6

0.8

0.6

ADAPTIVE

DELTA POWER

DISSIPATION

ADAPTIVE

0.4

0.2

0.4

0.2

78

76

78

76

V

= 5 V

= 1.8 V

IN

OUT

= 500 kHz

DELTA POWER

DISSIPATION

f

SW

74

0.0

20

74

0.0

20

0

5

10

15

0

5

10

15

I

− Output Current − A

OUT

I

− Output Current − A

OUT

Figure 14

Figure 15

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

LAYOUT CONSIDERATIONS

packaging

The UCC27221/2 are only available in TI’s thermally enhanced 14-pin PowerPad package. This package

offers exceptional thermal impedance with a junction-to-case rating of 2ꢂC/W. Shown as the crosshatched

region in Figure 16, PowerPad includes an exposed leadframe die pad located on the bottom side of the

package. Exposed pad dimensions for the PowerPadꢁ TSSOP 14-pin package are 69 mils x 56 mils (1.8 mm

x 1.4 mm). However, the exposed pad tolerances can be + 41 / − 2 mils (+ 1.05 /− .05 mm) due to position and

mold flow variation. Effectively removing the heat from the PowerPAD package requires a thermal land area,

shown as the shaded gray region in Figure 16, designed into the PCB directly beneath the package. A minimum

thermal land area of 5 mm by 3.4 mm is recommended as illustrated in Figure 16. Any tolerance variances of

the exposed PowerPad falls well within the thermal land area when the recommended minimum land area

is included on the printed circuit board. In addition, a 2-by-3 array of 13-mil thermal vias is required within the

exposed PowerPad area, as shown in Figure 16. If additional heat sinking capability is required, larger 25-mil

vias can be added to the thermal land area.

Required Vias on PowerPad Area

2 x 3 Array

3.4mm

(0.1339”)

0.65mm

0.33mm

(0.0256”)

(13 mil) dia Vias

Exposed

5.0mm

PowerPad

1.8mm (0.069”)

(0.1968”)

1.05mm

0.3mm

(0.0118”)

(0.0413”)

Optional Vias on Thermal Land Area

0.635mm

Exposed

PowerPad

(25 mil) dia Vias

1.4mm (0.056”)

Figure 16. TSSOP−14PWP Package Outline and Minimum PowerPADE PCB Thermal Land

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

REFERENCE DESIGN AND EVALUATION MODULE

A reference design is discussed in, 12 V to 1.8 V, 20-A High Efficiency Synchronous Buck Converter Using the

UCC27222 with Predictive Gate Driveꢁ, TI Literature Number SLUU140 and accompanying evaluation module

(EVM) SLUP192. The design highlights UCC27222 and its Predictive Gate Driveꢁ synchronous buck operation

using a simple single ended PWM controller. The schematic is shown in Figure 17.

Figure 17. Typical Application Diagram

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

TYPICAL CHARACTERISTICS

UVLO THRESHOLD

vs

TEMPERATURE

BIAS CURRENT

vs

TEMPERATURE (NO LOAD)

16

14

4.2

4.1

4.0

3.9

3.8

3.7

3.6

I

: VDD = 12 V, 500 kHz

DD

12

10

UVLO On

8

6

4

2

0

I

: VDD = 8.5 V, Static

UVLO Off

DD

I

: VLO = 12 V, Static, 5 V Only Systems

VLO

3.5

3.4

3.3

3.2

−50

−25

0

25

50

75

100

125

−50

−25

0

25

T − Temperature − °C

A

50

75

100

125

T

− Temperature − °C

A

Figure 18

Figure 19

VLO LINE REGULATION

vs

VLO LOAD REGULATION

vs

TEMPERATURE (NO LOAD, IVLO = 0 mA)

TEMPERATURE

6.8

6.7

6.6

6.7

6.6

V

DD

= 20 V

I

= 0 mA

VLO

6.5

V

DD

= 12 V

I

= 100 mA

VLO

6.4

6.3

6.4

6.3

6.2

−50

−25

0

25

50

75

100

125

−50

−25

0

25

50

75

100

125

T

A

− Temperature − °C

T

A

− Temperature − °C

Figure 20

Figure 21

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

TYPICAL CHARACTERISTICS

DROPOUT VOLTAGE (VDD AT VLO = 6.175 V)

vs

OUTPUT CURENT

TEMPERATURE (I

= 100 mA)

VLO

8.6

8.4

8.2

8.0

7.8

7.6

7.4

8.6

8.4

8.2

8.0

7.8

7.6

7.4

7.2

7.0

7.2

7.0

−50

−25

0

25

50

75

100

125

0

50

100

T

A

− Temperature − °C

IVLO − VLO Output Current − mA

Figure 22

Figure 23

VLO SHORT CIRCUIT CURRENT

INPUT THRESHOLD

vs

TEMPERATURE

TEMPERATURE (VDD = 10 V TO 20 V)

4.0

3.5

3.0

2.5

2.0

1.5

1.0

300

Input Threshold Rising

290

280

270

260

Input Threshold Falling

Input Threshold Hyst.

250

240

230

220

210

200

0.5

0.0

−50

−25

0

25

50

75

100

125

−50

−25

0

25

50

75

100

125

T

A

− Temperature − °C

T

A

− Temperature − °C

Figure 24

Figure 25

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

TYPICAL CHARACTERISTICS

PREDICTIVE DELAY BIT WEIGHT

PROPAGATION DELAYS

vs

TEMPERATURE

4.7

4.5

120

100

t

, G2

OFF

4.3

4.1

80

60

t

, G1

OFF

3.9

3.7

40

20

3.5

0

−50

−25

0

25

50

75

100

125

−50

−25

0

25

50

75

100

125

T

A

− Temperature − °C

T

A

− Temperature − °C

Figure 26

Figure 27

PREDICTIVE DELAY RANGE

vs

TEMPERATURE

60

50

t

, G1 Max

ON

40

30

40

30

t

, G2 Max

ON

20

10

20

10

0

t

, G1 Min

t

, G2 Min

ON

−10

−20

−30

−20

−30

−50

−25

0

25

50

75

100

125

−50

−25

0

25

50

75

100

125

T

A

− Temperature − °C

T

A

− Temperature − °C

Figure 28

Figure 29

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

TYPICAL CHARACTERISTICS

G1 RISE AND FALL TIMES

G2 RISE AND FALL TIMES

vs

TEMPERATURE (2.2 nF)

30

25

30

25

20

Fall Time

20

Rise Time

15

10

15

Rise Time

Fall Time

10

5

0

−50

−25

0

25

50

75

100

125

−50

−25

0

25

50

75

100

125

T

A

− Temperature − °C

T

A

− Temperature − °C

Figure 30

Figure 31

G1 SINK RESISTANCE

vs

G1 SOURCE RESISTANCE

vs

TEMPERATURE (R

)

TEMPERATURE

DS(on)

40

35

30

25

20

15

10

5

1.5

1.4

1.3

1.2

1.1

1.0

0.9

0.8

0.7

0.6

0.5

0

−50

−25

0

25

50

75

100

125

−50

−25

0

25

50

75

100

125

T

A

− Temperature − °C

T

A

− Temperature − °C

Figure 32

Figure 33

23

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SLUS486B − AUGUST 2001 − REVISED JULY 2003

TYPICAL CHARACTERISTICS

G2 SOURCE RESISTANCE

G2 SINK RESISTANCE

vs

TEMPERATURE

TEMPERATURE (R

)

DS(on)

40

35

30

40

35

30

25

20

25

20

15

10

5

15

10

5

0

−50

−25

0

25

50

75

100

125

−50

−25

0

25

50

75

100

125

T

A

− Ambient Temperature − °C

T

A

− Temperature − °C

Figure 34

Figure 35

是否无铅：	不含铅	是否Rohs认证：	符合
生命周期：	Active	零件包装代码：	TSSOP
包装说明：	HTSSOP,	针数：	14
Reach Compliance Code：	compliant	ECCN代码：	EAR99
HTS代码：	8542.39.00.01	风险等级：	5.28
高边驱动器：	YES	接口集成电路类型：	HALF BRIDGE BASED MOSFET DRIVER
JESD-30 代码：	R-PDSO-G14	JESD-609代码：	e4
长度：	5 mm	湿度敏感等级：	2
功能数量：	1	端子数量：	14
最高工作温度：	105 °C	最低工作温度：	-40 °C
最大输出电流：	3 A	标称输出峰值电流：	3.3 A
封装主体材料：	PLASTIC/EPOXY	封装代码：	HTSSOP
封装形状：	RECTANGULAR	封装形式：	SMALL OUTLINE, HEAT SINK/SLUG, THIN PROFILE, SHRINK PITCH
峰值回流温度（摄氏度）：	260	认证状态：	Not Qualified
座面最大高度：	1.2 mm	最大压摆率：	100 mA
最大供电电压：	20 V	最小供电电压：	8.5 V
标称供电电压：	12 V	表面贴装：	YES
温度等级：	INDUSTRIAL	端子面层：	Nickel/Palladium/Gold (Ni/Pd/Au)
端子形式：	GULL WING	端子节距：	0.65 mm
端子位置：	DUAL	处于峰值回流温度下的最长时间：	NOT SPECIFIED
断开时间：	0.11 µs	宽度：	4.4 mm
Base Number Matches：	1

型号	制造商	描述	替代类型	文档
UCC27222PWPRG4	TI	3.3A HALF BRDG BASED MOSFET DRIVER, PDSO14, GREEN, PLASTIC, HTSSOP-14	完全替代
UCC27222PWPG4	TI	高效 Predictive 技术同步降压驱动器 \| PWP \| 14 \| -40 to 1	完全替代
UCC27222PWP	TI	HIGH EFFICIENCY PREDICTIVE SYNCHRONOUS BUCK DRIVER	类似代替

型号	制造商	描述	价格	文档
UCC27222PWPRG4	TI	3.3A HALF BRDG BASED MOSFET DRIVER, PDSO14, GREEN, PLASTIC, HTSSOP-14	获取价格
UCC27223	TI	HIGH-EFFICIENCY PREDICTIVE SYNCHRONOUS BUCK DRIVER WITH ENABLE	获取价格
UCC27223PWP	TI	HIGH-EFFICIENCY PREDICTIVE SYNCHRONOUS BUCK DRIVER WITH ENABLE	获取价格
UCC27223PWPG4	TI	HIGH-EFFICIENCY PREDICTIVE SYNCHRONOUS BUCK DRIVER WITH ENABLE	获取价格
UCC27223PWPR	TI	HIGH-EFFICIENCY PREDICTIVE SYNCHRONOUS BUCK DRIVER WITH ENABLE	获取价格
UCC27223PWPRG4	TI	HIGH-EFFICIENCY PREDICTIVE SYNCHRONOUS BUCK DRIVER WITH ENABLE	获取价格
UCC27282	TI	具有 5V UVLO、互锁和使能功能的 3A、120V 半桥栅极驱动器	获取价格
UCC27282-Q1	TI	具有 5V UVLO、互锁功能和使能功能的汽车类 3A、120V 半桥驱动器	获取价格
UCC27282-Q1-V02	TI	UCC27282-Q1 Automotive 120-V Half-Bridge Driver with Cross Conduction Protection and Low Switching Losses	获取价格
UCC27282-Q1_V03	TI	UCC27282-Q1 Automotive 120-V Half-Bridge Driver with Cross Conduction Protection and Low Switching Losses	获取价格

UCC27222PWPR

UCC27222PWPR 概述

UCC27222PWPR 规格参数

UCC27222PWPR 数据手册

UCC27222PWPR CAD模型

UCC27222PWPR 替代型号

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