EVAL-AD7864-3CB3 [ADI]
4-Channel, Simultaneous Sampling, High Speed, 12-Bit ADC; 4通道，同步采样，高速， 12位ADC
| 型号： | EVAL-AD7864-3CB3 |
| 厂家： | 
ADI |
| 描述： | 4-Channel, Simultaneous Sampling, High Speed, 12-Bit ADC |
| 文件： | 总28页 (文件大小：760K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
4-Channel, Simultaneous Sampling,
High Speed, 12-Bit ADC
AD7864
FEATURES
FUNCTIONAL BLOCK DIAGRAM
DV
V
DRIVE
AV
V
V
GND
REF
DD
DD
REF
High speed (1.65 μs) 12-bit ADC
4 simultaneously sampled inputs
4 track-and-hold amplifiers
0.35 μs track-and-hold acquisition time
1.65 μs conversion time per channel
HW/SW select of channel sequence for conversion
Single-supply operation
TRACK-AND-HOLD
×4
6kΩ
2.5V
REFERENCE
STBY
DGND
AGND
RD
V
SIGNAL
SCALING
IN1A
V
IN1B
AD7864
V
V
IN2A
IN2B
SIGNAL
SCALING
MUX
V
V
IN3A
IN3B
SIGNAL
SCALING
DB11
DB0
OUTPUT
DATA
REGISTERS
12-BIT
ADC
Selection of input ranges
V
V
IN4A
IN4B
SIGNAL
SCALING
10 V, 5 V for AD7864-1
SOFTWARE
LATCH
DB0 TO DB3
CS
2.5 V for AD7864-3 0 V to 2.5 V, 0 V to 5 V for AD7864-2
High speed parallel interface that allows
Interfacing to 3 V processors
FRSTDATA
WR
BUSY
EOC
CONVERSION
CONTROL LOGIC
INT/EXT CLOCK
SELECT
INT
CLOCK
Low power, 90 mW typical
Power saving mode, 20 μW typical
Overvoltage protection on analog inputs
SL1 SL2 SL3 SL4
CLKIN
AGND AGND
CONVST
H/S
SEL
INT/EXT
CLK
Figure 1.
APPLICATIONS
AC motor control
Uninterrupted power supplies
Data acquisition systems
Communications
Data is read from the part by a 12-bit parallel data bus using the
CS RD
standard and
signals. Maximum throughput for a single
channel is 500 kSPS. For all four channels, the maximum throughput
is 130 kSPS for the read-during-conversion sequence operation.
The throughput rate for the read-after-conversion sequence
operation depends on the read cycle time of the processor. See
the Timing and Control section. The AD7864 is available in a
small (0.3 square inch area) 44-lead MQFP.
GENERAL DESCRIPTION
The AD7864 is a high speed, low power, 4-channel, simulta-
neous sampling 12-bit analog-to-digital converter (ADC) that
operates from a single 5 V supply. The part contains a 1.65 μs
successive approximation ADC, four track-and-hold amplifiers,
a 2.5 V reference, an on-chip clock oscillator, signal conditioning
circuitry, and a high speed parallel interface. The input signals
on four channels sample simultaneously preserving the relative
phase information of the signals on the four analog inputs. The
part accepts analog input ranges of 10 V, 5 V (AD7864-1), 0 V
to +2.5 V, 0 V to +5 V (AD7864-2), and 2.5 V (AD7864-3).
PRODUCT HIGHLIGHTS
1. Four track-and-hold amplifiers and a fast (1.65 μs) ADC for
simultaneous sampling and conversion of any subset of the
four channels.
2. A single 5 V supply consuming only 90 mW typical, makes
it ideal for low power and portable applications. See the
Standby Mode Operation section.
3. High speed parallel interface for easy connection to micro-
processors, microcontrollers, and digital signal processors.
4. Available in three versions with different analog input
ranges. The AD7864-1 offers the standard industrial input
ranges of 10 V and 5 V; the AD7864-3 offers the common
signal processing input range of 2.5 V; the AD7864-2 can
be used in unipolar, 0 V to 2.5 V and 0 V to 5 V,
applications.
Any subset of the four channels can be converted to maximize
the throughput rate on the selected sequence. Select the channels to
convert via hardware (channel select input pins) or software (pro-
gramming the channel select register).
CONVST
A single conversion start signal (
) simultaneously places
all the track-and-holds into hold and initiates a conversion se-
EOC
quence for the selected channels. The
signal indicates the end
5. Features very tight aperture delay matching between the
of each individual conversion in the selected conversion sequence.
The BUSY signal indicates the end of the conversion sequence.
four input sample-and-hold amplifiers.
Rev. D
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registeredtrademarks are theproperty of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113 ©1998–2009 Analog Devices, Inc. All rights reserved.
AD7864
TABLE OF CONTENTS
Features .............................................................................................. 1
Standby Mode Operation.......................................................... 18
Accessing the Output Data Registers....................................... 18
Offset and Full-Scale Adjustment ................................................ 20
Positive Full-Scale Adjust .......................................................... 20
Negative Full-Scale Adjust......................................................... 20
Dynamic Specifications ................................................................. 21
Signal-to-Noise Ratio (SNR)..................................................... 21
Effective Number of Bits ........................................................... 21
Intermodulation Distortion...................................................... 21
AC Linearity Plots...................................................................... 22
Measuring Aperture Jitter.......................................................... 22
Microprocessor Interfacing........................................................... 24
AD7864 to ADSP-2100/ADSP-2101/ADSP-2102 Interface . 24
AD7864 to TMS320C5x Interface............................................ 24
AD7864 to MC68HC000 Interface.......................................... 24
Vector Motor Control................................................................ 25
Multiple AD7864s in A System................................................. 26
Outline Dimensions....................................................................... 27
Ordering Guide .......................................................................... 27
Applications....................................................................................... 1
General Description......................................................................... 1
Functional Block Diagram .............................................................. 1
Product Highlights ........................................................................... 1
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Timing Characteristics ................................................................ 5
Absolute Maximum Ratings............................................................ 6
ESD Caution.................................................................................. 6
Pin Configuration and Function Descriptions............................. 7
Terminology ...................................................................................... 9
Theory of Operation ...................................................................... 11
Converter Details........................................................................ 11
Circuit Description......................................................................... 13
Analog Input ............................................................................... 13
Selecting a Conversion Sequence ................................................. 15
Timing and Control ................................................................... 15
Using an External Clock............................................................ 17
REVISION HISTORY
2/09—Rev. C to Rev. D
3/04—Rev. A to Rev. B.
Change to t2 Parameter, Table 2...................................................... 5
Changes to Specifications and to Footnote 4.................................2
Changes to Timing Characteristics Footnote 1.............................4
Addition to Absolute Maximum Ratings .......................................5
Changes to Ordering Guide.............................................................5
Changes to Figure 7.........................................................................11
Changes to Figure 11...................................................................... 13
Updated Outline Dimensions....................................................... 19
Added Revision History ................................................................ 20
Updated Publication Code............................................................ 20
2/09—Rev. B to Rev. C
Updated Format..................................................................Universal
Changes to t5 Timing Parameter, Table 2....................................... 5
Changes to Figure 15...................................................................... 20
Changes to AD7864 to MC68HC000 Interface Section............ 24
Changes to Figure 25...................................................................... 24
Updated Outline Dimensions....................................................... 29
Changes to Ordering Guide .......................................................... 29
Rev. D | Page 2 of 28
AD7864
SPECIFICATIONS
VDD = 5 V 5ꢀ, AGND = DGND = 0 V, VREF = internal, clock = internal; all specifications TMIN to TMAX, unless otherwise noted.
Table 1.
Parameter
A Version1
B Version
Unit
Test Conditions/Comments
SAMPLE AND HOLD
−3 dB Full Power Bandwidth
Aperture Delay
Aperture Jitter
Aperture Delay Matching
3
3
MHz typ
ns max
ps max
ns max
20
50
4
20
50
4
DYNAMIC PERFORMANCE2
Signal-to-(Noise + Distortion) Ratio3
fIN = 100.0 kHz, fS = 500 kSPS
@ 25°C
TMIN to TMAX
70
70
−80
−80
72
70
−80
−80
dB min
dB min
dB max
dB max
Total Harmonic Distortion3
Peak Harmonic or Spurious Noise3
Intermodulation Distortion3
Second-Order Terms
Third-Order Terms
Channel-to-Channel Isolation3
DC ACCURACY
fa = 49 kHz, fb = 50 kHz
−80
−80
−80
−80
−80
−80
dB typ
dB typ
dB max
fIN = 50 kHz sine wave
Any channel
Resolution
12
1
0.9
12
1ꢀ2
0.9
Bits
LSB max
LSB max
Relative Accuracy3
Differential Nonlinearity3
AD7864-1
No missing codes
Positive Gain Error3
Positive Gain Error Match3
Negative Gain Error3
Negative Gain Error Match3
Bipolar Zero Error
3
+3
3
+3
4
3
3
3
3
3
2
LSB max
LSB max
LSB max
LSB max
LSB max
LSB max
Bipolar Zero Error Match
AD7864-3
+2
Positive Gain Error3
Positive Gain Error Match3
Negative Gain Error3
Negative Gain Error Match3
Bipolar Zero Error
3
2
3
LSB max
LSB max
LSB max
LSB max
LSB max
LSB max
2
3
Bipolar Zero Error Match
2
AD7864-2
Positive Gain Error3
Positive Gain Error Match3
Unipolar Offset Error
Unipolar Offset Error Match
ANALOG INPUTS
3
3
3
LSB max
LSB max
LSB max
LSB max
2
AD7864-1
Input Voltage Range
Input Resistance
5, 10
9, 18
5, 10
9, 18
V
kΩ min
AD7864-3
Input Voltage Range
Input Resistance
2.5
4.5
2.5
4.5
V
kΩ min
Rev. D | Page 3 of 28
AD7864
Parameter
A Version1
B Version
Unit
Test Conditions/Comments
AD7864-2
Input Voltage Range
0 to 2.5, 0 to 5 0 to 2.5, 0 to 5
100 100
V
Input Current (0 V to 2.5 V Option)
Input Resistance (0 V to 5 V Option)
REFERENCE INPUTꢀOUTPUT
VREF In Input Voltage Range
VREF In Input Capacitance4
VREF Out Output Voltage
VREF Out Error @ 25°C
VREF Out Error TMIN to TMAX
VREF Out Temperature Coefficient
VREF Out Output Impedance
LOGIC INPUTS
nA max
kΩ min
9
9
2.375ꢀ2.625
2.375ꢀ2.625
VMINꢀVMAX
pF max
V nom
mV max
mV max
ppmꢀ°C typ
kΩ typ
2.5 V 5ꢁ
10
2.5
10
20
25
6
10
2.5
10
20
25
6
See the Reference section
Input High Voltage, VINH
Input Low Voltage, VINL
2.4
0.8
10
2.4
0.8
10
V min
VDD = 5 V 5ꢁ
VDD = 5 V 5ꢁ
V max
μA max
pF max
Input Current, IIN
4
Input Capacitance, CIN
10
10
LOGIC OUTPUTS
Output High Voltage, VOH
Output Low Voltage, VOL
DB11 to DB0
4.0
0.4
4.0
0.4
V min
V max
ISOURCE = 400 μA
ISINK = 1.6 mA
High Impedance
Leakage Current
Capacitance4
10
10
10
10
μA max
pF max
Output Coding
AD7864-1, AD7864-3
AD7864-2
Twos complement
Straight (natural) binary
CONVERSION RATE
Conversion Time
Track-And-Hold Acquisition Time2, 3
Throughput Time
POWER REQUIREMENTS
VDD
1.65
0.35
130
1.65
0.35
130
μs max
μs max
kSPS max
For one channel
For all four channels
5
5
V nom
5ꢁ for specified performance
IDD
5 μA typical, logic inputs = 0 V or VDD
Normal Mode
Standby Mode
24
20
24
20
mA max
μA max
Typically 4 μA
Power Dissipation
Normal Mode
Standby Mode
120
100
120
100
mW max
μW max
Typically 90 mW
Typically 20 μW
1 Temperature ranges are as follows: A, B versions: –40°C to +85°C. The A version is fully specified up to 105°C with a maximum sample rate of 450 kSPS and IDD
maximum (normal mode) of 26 mA.
2 Performance is measured through the full channel (SHA and ADC).
3 See the Terminology section.
4 Sample tested at initial release to ensure compliance.
Rev. D | Page 4 of 28
AD7864
TIMING CHARACTERISTICS
VDRIVE = 5 V 5ꢀ, AGND = DGND = 0 V, VREF = internal, clock = internal; all specifications TMIN to TMAX, unless otherwise noted.1, 2
Table 2.
Parameter
tCONV
A, B Versions
1.65
13
2.6
0.34
Unit
Test Conditions/Comments
Conversion time, internal clock
Conversion time, external clock
CLKIN = 5 MHz
Acquisition time
Selected number of channels multiplied by (tCONV
μs max
Clock cycles
μs max
μs max
μs max
tACQ
tBUSY
No. of channels ×
(tCONV + t9) − t9
+
pulse
EOC
width)—
EOC
pulse width
tWAKE-UP —External VREF
2
μs max
ms max
ns min
ns max
rising edge to
rising edge to
rising edge
rising edge
STBY
CONVST
CONVST
3
tWAKE-UP —Internal VREF
6
STBY
t1
t2
35
70
pulse width
CONVST
CONVST
rising edge to BUSY rising edge
READ OPERATION
t3
t4
t5
0
ns min
ns min
ns min
ns min
ns max
ns max
ns min
ns max
ns min
ns min
ns max
ns max
ns max
ns min
to
CS RD
setup time
hold time
0
to
CS RD
35
40
35
40
5
Read pulse width, VDRIVE = 5 V
Read pulse width, VDRIVE = 3 V
Data access time after falling edge of , VDRIVE = 5 V
RD
4
t6
Data access time after falling edge of , VDRIVE = 3 V
RD
5
t7
Bus relinquish time after rising edge of
RD
30
10
75
180
70
15
0
t8
t9
Time between consecutive reads
pulse width
EOC
t10
t11
t12
rising edge to FRSTDATA edge (rising or falling)
RD
falling edge to FRSTDATA falling delay
EOC
to
EOC RD
delay
WRITE OPERATION
t13
t14
t15
t16
t17
20
0
ns min
ns min
ns min
ns min
ns min
pulse width
WR
to
setup time
CS WR
0
to hold time
WR CS
5
Input data setup time of rising edge of
Input data hold time
WR
5
1 Sample tested at initial release to ensure compliance. All input signals are measured with tr = tf = 1 ns (10ꢁ to 90ꢁ of 5 V) and timed from a voltage level of 1.6 V.
2 See Figure 9, Figure 10,and Figure 11.
3 Refer to the Standby Mode Operation section. The maximum specification of 6 ms is valid when using a 0.1 μF decoupling capacitor on the VREF pin.
4 Measured with the load circuit of Figure 2 and defined as the time required for an output to cross 0.8 V or 2.4 V.
5 These times are derived from the measured time taken by the data outputs to change 0.5 V when loaded with the circuit shown in Figure 2. The measured number is
then extrapolated back to remove the effects of charging or discharging the 50 pF capacitor. This means that the times quoted in the timing characteristics are the
true bus relinquish times of the part, and as such, are independent of external bus loading capacitances.
1.6mA
TO
OUTPUT
1.6V
50pF
400µA
Figure 2. Load Circuit for Access Time and Bus Relinquish Time
Rev. D | Page 5 of 28
AD7864
ABSOLUTE MAXIMUM RATINGS
TA = 25°C, unless otherwise noted.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Table 3.
Parameter
AVDD to AGND
DVDD to DGND
AGND to DGND
AVDD to DVDD
Analog Input Voltage to AGND
AD7864-1 ( 10 V Input Range)
AD7864-1 ( 5 V Input Range)
AD7864-3
Rating
−0.3 V to +7 V
−0.3 V to +7 V
−0.3 V to +0.3 V
−0.3 V to +0.3 V
ESD CAUTION
20 V
−7 V to +20 V
−7 V to +20 V
AD7864-2
−1 V to +20 V
Reference Input Voltage to AGND
Digital Input Voltage to DGND
Digital Output Voltage to DGND
VDRIVE to AGND
−0.3 V to VDD + 0.3 V
−0.3 V to VDD + 0.3 V
−0.3 V to VDD + 0.3 V
−0.3 V to AVDD + 0.3 V
−0.3 V to DVDD + 0.3 V
VDRIVE to DGND
Operating Temperature Range
Commercial (A and B Versions)
Storage Temperature Range
Junction Temperature
MQFP Package, Power Dissipation
θJA Thermal Impedance
Lead Temperature, Soldering
Vapor Phase (60 sec)
Infrared (15 sec)
−40°C to +85°C
−65°C to +150°C
150°C
450 mW
95°CꢀW
215°C
220°C
Rev. D | Page 6 of 28
AD7864
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
44 43 42 41 40 39 38 37 36 35 34
1
33
32
31
30
29
28
27
26
25
24
23
BUSY
DB7
PIN 1
2
FRSTDATA
DB8
3
CONVST
DB9
4
CS
DB10
5
RD
DB11
AD7864
TOP VIEW
(Not to Scale)
6
WR
CLKIN
INT/EXT CLK
AGND
7
SL1
8
SL2
9
AV
DD
SL3
10
V
SL4
REF
11
V
GND
H/S SEL
REF
12 13 14 15 16 17 18 19 20 21 22
Figure 3. Pin Configuration
Table 4. Pin Function Descriptions
Pin No. Mnemonic
Description
1
2
3
BUSY
Busy Output. The busy output is triggered high by the rising edge of
is completed on all selected channels.
First Data Output. FRSTDATA is a logic output which, when high, indicates that the output data register pointer
is addressing Register 1—see the Accessing the Output Data Registers section.
Convert Start Input. Logic input. A low-to-high transition on this input puts all track-and-holds into their hold
mode and starts conversion on the selected channels. In addition, the state of the channel sequence selection is
and remains high until conversion
CONVST
FRSTDATA
CONVST
also latched on the rising edge of
.
CONVST
Chip Select Input. Active low logic input. The device is selected when this input is active.
Read Input. Active low logic input that is used in conjunction with low to enable the data outputs. Ensure the
4
5
CS
RD
CS
pin is at logic high while performing a read operation.
WR
6
WR
Write Input. A rising edge on the
input, with low and
CS
high, latches the logic state on DB0 to DB3 into
RD
WR
the channel select register.
7 to 10
11
SL1 to SL4
Hardware Channel Select. Conversion sequence selection can also be made via the SL1 to SL4 pins if ꢀS SEL is
H
Logic 0. The selection is latched on the rising edge of
. See the Selecting a Conversion Sequence section.
CONVST
ꢀS SEL
H
HardwareꢀSoftware Select Input. When this pin is at Logic 0, the AD7864 conversion sequence selection is
controlled via the SL1 to SL4 input pins. When this pin is at Logic 1, the sequence is controlled via the channel
select register. See the Selecting a Conversion Sequence section.
12
AGND
Analog Ground. General analog ground. Connect this AGND pin to the AGND plane of the system.
Analog Inputs. See the Analog Input section.
Analog Ground. Analog ground reference for the attenuator circuitry. Connect this AGND pin to the AGND plane
of the system.
13 to 16 VIN4x, VIN3x
17 AGND
18 to 21 VIN2x, VIN1x
Analog Inputs. See the Analog Input section.
22
23
24
STBY
Standby Mode Input. TTL-compatible input that is used to put the device into the power save or standby mode.
The
input is high for normal operation and low for standby operation.
STBY
VREFGND
VREF
Reference Ground. This is the ground reference for the on-chip reference buffer of the part. Connect the
VREFGND pin to the AGND plane of the system.
Reference InputꢀOutput. This pin provides access to the internal reference (2.5 V 5ꢁ) and also allows the
internal reference to be overdriven by an external reference source (2.5 V). Connect a 0.1 μF decoupling
capacitor between this pin and AGND.
25
26
AVDD
AGND
Analog Positive Supply Voltage, 5.0 V 5ꢁ.
Analog Ground. Analog ground reference for the DAC circuitry.
Rev. D | Page 7 of 28
AD7864
Pin No. Mnemonic
Description
27
ꢀEXT CLK
INT
InternalꢀExternal Clock Select Input. When this pin is at Logic 0, the AD7864 uses its internally generated master
clock. When this pin is at Logic 1, the master clock is generated externally to the device.
28
CLKIN
Conversion Clock Input. This is an externally applied clock that allows the user to control the conversion rate of
the AD7864. Each conversion needs 14 clock cycles for the conversion to be completed and an
pulse to be
EOC
generated. The clock should have a duty cycle that is no worse than 60ꢀ40. See the Using An External Clock
section.
29 to 34 DB11 to DB6
Data Bit 11 is the MSB, followed by Data Bit 10 to Data Bit 6. Three-state TTL outputs. Output coding is twos
complement for the AD7864-1 and AD7864-3. Output coding is straight (natural) binary for the AD7864-2.
35
36
DVDD
VDRIVE
Positive Supply Voltage for Digital Section, 5.0 V 5ꢁ. Connect a 0.1 μF decoupling capacitor between this pin
and AGND. Both DVDD and AVDD should be externally tied together.
This pin provides the positive supply voltage for the output drivers (DB0 to DB11), BUSY,
, and FRSTDATA. It
EOC
is normally tied to DVDD. Decouple VDRIVE with a 0.1 μF capacitor to improve performance when reading during
the conversion sequence. To facilitate interfacing to 3 V processors and DSPs, the output data drivers can also be
powered by a 3 V 10ꢁ supply.
37
DGND
Digital Ground. This is the ground reference for digital circuitry. Connect this DGND pin to the AGND plane of
the system at the AGND pin.
38, 39
DB5, DB4
Data Bit 5 to Data Bit 4. Three-state TTL outputs.
40 to 43 DB3 to DB0
Data Bit 3 to Data Bit 0. Bidirectional data pins. When a read operation takes place, these pins are three-state TTL
outputs. The channel select register is programmed with the data on the DB0 to DB3 pins with standard and
CS
signals. DB0 represents Channel 1, and DB3 represents Channel 4.
WR
44
EOC
End-of-Conversion. Active low logic output indicating conversion status. The end of each conversion in a
conversion sequence is indicated by a low-going pulse on this line.
Rev. D | Page 8 of 28
AD7864
TERMINOLOGY
Signal-to-(Noise + Distortion) Ratio
Channel-to-Channel Isolation
Channel-to-channel isolation is a measure of the level of
crosstalk between channels. It is measured by applying a full-
scale 50 kHz sine wave signal to all nonselected input channels
and determining how much that signal is attenuated in the
selected channel. The figure given is the worst case across all
four channels.
This is the measured ratio of signal-to-(noise + distortion) at
the output of the ADC. The signal is the rms amplitude of the
fundamental. Noise is the rms sum of all nonfundamental signals
up to half the sampling frequency (fS/2), excluding dc. The ratio
depends on the number of quantization levels in the digitization
process; the more levels, the smaller the quantization noise. The
theoretical signal-to-(noise + distortion) ratio for an ideal N-bit
converter with a sine wave input is given by
Relative Accuracy
Relative accuracy, or endpoint nonlinearity, is the maximum
deviation from a straight line passing through the endpoints of
the ADC transfer function.
Signal-to-(Noise + Distortion) = (6.02 N + 1.76) dB
Thus, for a 12-bit converter, this is 74 dB.
Differential Nonlinearity
This is the difference between the measured and the ideal 1 LSB
change between any two adjacent codes in the ADC.
Total Harmonic Distortion (THD)
THD is the ratio of the rms sum of harmonics to the
fundamental. For the AD7864, it is defined as
Positive Full-Scale Error
This is the deviation of the last code transition (01...110 to 01...111)
from the ideal, 4 × VREF − 3/2 LSB (AD7864-1, 10 V), or 2 ×
2
2
2
2
2
V2 +V3 +V4 +V5 +V6
THD(dB) = 20 log
V1
V
REF − 3/2 LSB (AD7864-1, 5 V range), or VREF − 3/2 LSB
where V1 is the rms amplitude of the fundamental, and V2, V3,
V4, V5, and V6 are the rms amplitudes of the second through the
fifth harmonics.
(AD7864-3, 2.5 V range), after the bipolar offset error has
been adjusted out.
Positive Full-Scale Error (AD7864-2, 0 V to 2.5 V and 0 V to 5 V)
This is the deviation of the last code transition (11...110 to 11...111)
from the ideal 2 × VREF − 3/2 LSB (AD7864-2, 0 V to 5 V range)
or VREF − 3/2 LSB (AD7864-2, 0 V to 2.5 V range), after the
unipolar offset error has been adjusted out.
Peak Harmonic or Spurious Noise
Peak harmonic or spurious noise is defined as the ratio of the
rms value of the next largest component in the ADC output
spectrum (up to fS/2 and excluding dc) to the rms value of the
fundamental. Normally, the value of this specification is deter-
mined by the largest harmonic in the spectrum, but for parts
where the harmonics are buried in the noise floor, it is a noise peak.
Bipolar Zero Error (AD7864-1, 10 V/ 5 V, AD7864-3, 2.5 V)
This is the deviation of the midscale transition (all 0s to all 1s)
from the ideal, AGND − 1/2 LSB.
Intermodulation Distortion
Unipolar Offset Error (AD7864-2, 0 V to 2.5 V and 0 V to 5 V)
This is the deviation of the first code transition (00...000 to
00...001) from the ideal, AGND + 1/2 LSB.
With inputs consisting of sine waves at two frequencies, fa and
fb, any active device with nonlinearities creates distortion products
at sum and difference frequencies of mfa nfb, where m, n = 0,
1, 2, 3, and so on. Intermodulation terms are those for which
neither m nor n are equal to zero. For example, second-order
terms include (fa + fb) and (fa − fb), whereas third-order terms
include (2 fa + fb), (2 fa − fb), (fa + 2 fb), and (fa − 2 fb).
Negative Full-Scale Error (AD7864-1, 10 V/ 5 V, and
AD7864-3, 2.5 V)
This is the deviation of the first code transition (10...000 to
10...001) from the ideal, −4 × VREF + 1/2 LSB (AD7864-1, 10 V),
−2 × VREF + 1/2 LSB (AD7864-1, 5 V range) or −VREF + 1/2 LSB
(AD7864-3, 2.5 V range), after bipolar zero error has been
adjusted out.
The AD7864 is tested using the CCIF standard, where two input
frequencies near the top end of the input bandwidth are used.
In this case, the second- and third-order terms are of different
significance. The second-order terms are usually distanced in
frequency from the original sine waves, whereas the third-order
terms are usually at a frequency close to the input frequencies. As
a result, the second- and third-order terms are specified separately.
The calculation of the intermodulation distortion is as per the
THD specification where it is the ratio of the rms sum of the
individual distortion products to the rms amplitude of the funda-
mental expressed in decibels.
Track-and-Hold Acquisition Time
Track-and-hold acquisition time is the time required for the
output of the track-and-hold amplifier to reach its final value,
within 1/2 LSB, after the end of a conversion (the point at
which the track-and-hold returns to track mode). It also applies
to situations where there is a step input change on the input
voltage applied to the selected VINxA/VINxB input of the AD7864.
Rev. D | Page 9 of 28
AD7864
It means that the user must wait for the duration of the track-
and-hold acquisition time after the end of conversion or after a
step input change to VINxA/VINxB before starting another
conversion to ensure that the part operates to specification.
Rev. D | Page 10 of 28
AD7864
THEORY OF OPERATION
Conversion time for each channel of the AD7864 is 1.65 μs, and
the track-and-hold acquisition time is 0.35 μs. To obtain optimum
performance from the part, the read operation should not occur
during a channel conversion or during the 100 ns prior to the
CONVERTER DETAILS
The AD7864 is a high speed, low power, 4-channel simultaneous
sampling 12-bit ADC that operates from a single 5 V supply.
The part contains a 1.65 μs successive approximation ADC, four
track-and-hold amplifiers, an internal 2.5 V reference, and a
high speed parallel interface. There are four analog inputs that
can be simultaneously sampled, thus preserving the relative
phase information of the signals on all four analog inputs.
Thereafter, conversions are completed on the selected subset of
the four channels. The part accepts an analog input range of
10 V or 5 V (AD7864-1), 2.5 V (AD7864-3), and 0 V to
+2.5 V or 0 V to +5 V (AD7864-2). Overvoltage protection on
the analog inputs of the part allows the input voltage to go to
20 V, (AD7864-1 10 V range), −7 V or +20 V (AD7864-1
5 V range), −1 V to +20 V (AD7864-2), and −7 V to +20 V
(AD7864-3), without causing damage. The AD7864 has two
operating modes: reading-between-conversions and reading-
after-the-conversion sequence. These modes are discussed in
more detail in the Timing and Control section.
CONVST
next
rising edge. This allows the part to operate at
throughput rates up to 130 kHz for all four channels and
achieve data sheet specifications.
Track-and-Hold Amplifiers
The track-and-hold amplifiers on the AD7864 allow the ADCs
to accurately convert an input sine wave of full-scale amplitude
to 12-bit accuracy. The input bandwidth of the track-and-hold
is greater than the Nyquist rate of the ADC even when the ADC
is operated at its maximum throughput rate of 500 kSPS (that is,
the track-and-hold can handle input frequencies in excess of
250 kHz).
The track-and-hold amplifiers acquire input signals to 12-bit
accuracy in less than 350 ns. The operation of the track-and-
holds are essentially transparent to the user. The four track-and-
hold amplifiers sample their respective input channels simulta-
CONVST
, all four on-chip track-
A conversion is initiated on the AD7864 by pulsing the
CONVST
neously, on the rising edge of
. The aperture time for
CONVST
input. On the rising edge of
the track-and-holds (that is, the delay time between the external
CONVST
and-holds are placed into hold simultaneously and the conversion
sequence is started on all the selected channels. Channel selection
signal and the track-and-hold actually going into hold)
is typically 15 ns and, more importantly, is well matched across
the four track-and-holds on one device as well as being well
matched from device to device. This allows the relative phase
information between different input channels to be accurately
preserved. It also allows multiple AD7864s to sample more than
four channels simultaneously. At the end of a conversion sequence,
the part returns to its tracking mode. The acquisition time of
the track-and-hold amplifiers begin at this point.
H
is made via the SL1 to SL4 pins if /S SEL is Logic 0 or via the
H
channel select register if /S SEL is Logic 1—see the Selecting a
Conversion Sequence section. The channel select register is
programmed via the bidirectional data lines (DB0 to DB3) and
a standard write operation. The selected conversion sequence is
CONVST
latched on the rising edge of
, therefore, changing a
selection only takes effect once a new conversion sequence is
initiated. The BUSY output signal is triggered high on the rising
Reference
CONVST
edge of
and remains high for the duration of the conver-
The AD7864 contains a single reference pin, labeled VREF. The
sion sequence. The conversion clock for the part is generated
internally using a laser trimmed, clock oscillator circuit.
V
REF pin provides access to the 2.5 V reference within the part,
or it serves as the reference source for the part by connecting
REF to an external 2.5 V reference. The part is specified with a
There is also the option of using an external clock, by tying the
V
INT
/EXT CLK pin logic high, and applying an external clock to
2.5 V reference voltage. Errors in the reference source result in
gain errors in the transfer function of the AD7864 and adds to
the specified full-scale errors on the part. On the AD7864-1 and
AD7864-3, it also results in an offset error injected in the attenuator
stage; see Figure 4 and Figure 6.
the CLKIN pin. However, the optimum throughput is obtained
by using the internally generated clock—see the Using an
EOC
External Clock section. The
signal indicates the end of
each conversion in the conversion sequence. The BUSY signal
indicates the end of the full conversion sequence, and at this
time, all four track and holds return to tracking mode. The
conversion results can be read either at the end of the full
conversion sequence (indicated by BUSY going low), or as each
The AD7864 contains an on-chip 2.5 V reference. To use this
reference as the reference source for the AD7864, simply con-
nect a 0.1 μF disk ceramic capacitor from the VREF pin to AGND.
The voltage that appears at this pin is internally buffered before
being applied to the ADC. If this reference is used externally to
the AD7864, it should be buffered because the part has a FET
switch in series with the reference output resulting in a 6 kΩ
EOC
result becomes available (indicated by
read from the part via a 12-bit parallel data bus with standard
CS RD
going low). Data is
and
signals—see the Timing and Control section.
Rev. D | Page 11 of 28
AD7864
nominal source impedance for this output. The tolerance on the
internal reference is 10 mV at 25°C with a typical temperature
coefficient of 25 ppm/°C and a maximum error overtemperature
of 20 mV.
has the option of connecting an external reference to this VREF
pin. The external reference effectively overdrives the internal
reference and thus provides the reference source for the ADC.
The reference input is buffered before being applied to the ADC
with the maximum input current of 100 μA. Suitable reference
sources for the AD7864 include the AD680, AD780, REF192,
and REF43 precision 2.5 V references.
If the application requires a reference with a tighter tolerance or
the AD7864 needs to be used with a system reference, the user
Rev. D | Page 12 of 28
AD7864
CIRCUIT DESCRIPTION
Table 5. Ideal Input/Output Code Table for the AD7864-1
ANALOG INPUT
Analog Input1
Digital Output Code Transition
The AD7864 is offered in three models: the AD7864-1, where
each input can be configured for 10 V or a 5 V input voltage
range; the AD7864-3, which handles the input voltage range of
2.5 V; and the AD7864-2, where each input can be configured
to have a 0 V to +2.5 V or 0 V to +5 V input voltage range.
+FSRꢀ2 − 3ꢀ2 LSB2
+FSRꢀ2 − 5ꢀ2 LSB
+FSRꢀ2 − 7ꢀ2 LSB
AGND + 3ꢀ2 LSB
AGND + 1ꢀ2 LSB
AGND − 1ꢀ2 LSB
AGND − 3ꢀ2 LSB
−FSRꢀ2 + 5ꢀ2 LSB
−FSRꢀ2 + 3ꢀ2 LSB
−FSRꢀ2 + 1ꢀ2 LSB
011...110 to 011...111
011...101 to 011...110
011...100 to 011...101
000...001 to 000...010
000...000 to 000...001
111...111 to 000...000
111...110 to 111...111
100...010 to 100...011
100...001 to 100...010
100...000 to 100...001
AD7864-1
Figure 4 shows the analog input section of the AD7864-1. Each
input can be configured for 5 V or 10 V operation on the
AD7864-1. For 5 V (AD7864-1) operation, the VINxA and VINxB
inputs are tied together and the input voltage is applied to both.
For 10 V (AD7864-1) operation, the VINxB input is tied to AGND
and the input voltage is applied to the VINxA input. The VINxA and
1 FSR is full-scale range and is 20 V for the 10 V range and +10 V for the 5 V
range, with VREF = 2.5 V.
2 1 LSB = FSRꢀ4096 = 4.883 mV ( 10 V for the AD7864-1) and 2.441 mV ( 5 V
for the AD7864-1) with VREF = 2.5 V.
VINxB inputs are symmetrical and fully interchangeable. Thus for
ease of printed circuit board (PCB) layout on the 10 V range,
the input voltage may be applied to the VINxB input while the
AD7864-2
VINxA input is tied to AGND.
Figure 5 shows the analog input section of the AD7864-2. Each
input can be configured for 0 V to 5 V operation or 0 V to 2.5 V
operation. For 0 V to 5 V operation, the VINxB input is tied to
AGND and the input voltage is applied to the VINxA input. For
0 V to 2.5 V operation, the VINxA and VINxB inputs are tied together
and the input voltage is applied to both. The VINxA and VINxB
inputs are symmetrical and fully interchangeable. Thus for ease
of PCB layout on the 0 V to 5 V range, the input voltage may be
applied to the VINxB input while the VINxA input is tied to AGND.
AD7864-1
6kΩ
2.5V
REFERENCE
V
REF
TO ADC
REFERENCE
CIRCUITRY
R1
R4
R2
R3
V
IN1A
IN1B
For the AD7864-2, R1 = 6 kΩ and R2 = 6 kΩ. The designed
code transitions occur on successive integer least significant bit
values. Output coding is straight (natural) binary with 1 LSB =
FSR/4096 = 2.5 V/4096 = 0.61 mV, and 5 V/4096 = 1.22 mV, for
the 0 V to 2.5 V and 0 V to 5 V options, respectively.
TO INTERNAL
COMPARATOR
T/H
V
AGND
Figure 4. AD7864-1 Analog Input Structure
Table 6 shows the ideal input and output transfer function for
the AD7864-2.
For the AD7864-1, R1 = 6 kΩ, R2 = 24 kΩ, R3 = 24 kΩ, and
R4 = 12 kΩ. The resistor input stage is followed by the high
input impedance stage of the track-and-hold amplifier.
AD7864-2
The designed code transitions take place midway between
successive integer least significant bit values (that is, 1/2 LSB,
3/2 LSB, 5/2 LSB, and so forth). Least significant bit size is given
by the formula 1 LSB = FSR/4096. For the 5 V range, 1 LSB =
10 V/4096 = 2.44 mV. For the 10 V range, 1 LSB = 20 V/4096 =
4.88 mV. Output coding is twos complement binary with 1 LSB =
FSR/4096. The ideal input/output transfer function for the
AD7864-1 is shown in Table 5.
6kΩ
2.5V
REFERENCE
V
REF
TO ADC
REFERENCE
CIRCUITRY
R1
R2
V
V
IN1A
IN1B
TO INTERNAL
COMPARATOR
T/H
Figure 5. AD7864-2 Analog Input Structure
Rev. D | Page 13 of 28
AD7864
The designed code transitions take place midway between
successive integer least significant bit values (that is, 1/2 LSB,
3/2 LSB, 5/2 LSB, and so on). Least significant bit size is given by
the formula 1 LSB = FSR/4096. Output coding is twos comple-
ment binary with 1 LSB = FSR/4096 = 5 V/4096 = 1.22 mV. The
ideal input/ output transfer function for the AD7864-3 is shown
in Table 7.
Table 6. Ideal Input/Output Code Table for the AD7864-2
Analog Input1
+FSR − 3ꢀ2 LSB2
+FSR − 5ꢀ2 LSB
+FSR − 7ꢀ2 LSB
AGND + 5ꢀ2 LSB
AGND + 3ꢀ2 LSB
AGND + 1ꢀ2 LSB
Digital Output Code Transition
111...110 to 111...111
111...101 to 111...110
111...100 to 111...101
000...010 to 000...011
000...001 to 000...010
000...000 to 000...001
Table 7. Ideal Input/Output Code Table for the AD7864-3
1 FSR is the full-scale range and is 0 V to 2.5 V and 0 V to 5 V for the AD7864-2
with VREF = 2.5 V.
Analog Input1
Digital Output Code Transition
011...110 to 011...111
011...101 to 011...110
011...100 to 011...101
000...001 to 000...010
000...000 to 000...001
111...111 to 000...000
111...110 to 111...111
100...010 to 100...011
100...001 to 100...010
100...000 to 100...001
+FSRꢀ2 − 3ꢀ2 LSB2
+FSRꢀ2 − 5ꢀ2 LSB
+FSRꢀ2 − 7ꢀ2 LSB
AGND + 3ꢀ2 LSB
AGND + 1ꢀ2 LSB
AGND − 1ꢀ2 LSB
AGND − 3ꢀ2 LSB
−FSRꢀ2 + 5ꢀ2 LSB
−FSRꢀ2 + 3ꢀ2 LSB
−FSRꢀ2 + 1ꢀ2 LSB
2 1 LSB = FSRꢀ4096 and is 0.61 mV (0 V to 2.5 V) and 1.22 mV (0 V to 5 V) for the
AD7864-2 with VREF = 2.5 V.
AD7864-3
Figure 6 shows the analog input section of the AD7864-3. The
analog input range is 2.5 V on the VIN1A input. The VIN1B input
can be left unconnected, but if it is connected to a potential,
that potential must be AGND.
AD7864-3
6kΩ
2.5V
REFERENCE
1 FSR is the full-scale range and is 5 V, with VREF = 2.5 V.
2 1 LSB = FSRꢀ4096 = 1.22 mV ( 2.5 V − AD7864-3) with VREF = 2.5 V.
V
REF
TO ADC
REFERENCE
CIRCUITRY
R1
R2
TO INTERNAL
COMPARATOR
V
V
IN1A
IN1B
T/H
Figure 6. AD7864-3 Analog Input Structure
For the AD7864-3, R1 = 6 kΩ and R2 = 6 kΩ. As a result, drive
the VIN1A input from a low impedance source. The resistor input
stage is followed by the high input impedance stage of the track-
and-hold amplifier.
Rev. D | Page 14 of 28
AD7864
SELECTING A CONVERSION SEQUENCE
Any subset of the four channels, VIN1 to VIN4, can be selected for
conversion. The selected channels are converted in ascending
TIMING AND CONTROL
Reading Between Each Conversion in the Conversion
Sequence
order. For example, if the channel selection includes VIN4, VIN1
,
and VIN3, the conversion sequence is VIN1, VIN3, and then VIN4
.
Figure 9 shows the timing and control sequence required to
obtain the optimum throughput rate from the AD7864. To
obtain the optimum throughput from the AD7864, the user
must read the result of each conversion as it becomes available.
The timing diagram in Figure 9 shows a read operation each
The conversion sequence selection can be made either by using
the hardware channel select input pins (SL1 through SL4) or by
programming the channel select register. A logic high on a
hardware channel select pin (or Logic 1 in the channel select
CONVST
register) when
goes logic high marks the associated
EOC
time the
signal goes logic low. The timing in Figure 9
analog input channel for inclusion in the conversion sequence.
shows a conversion on all four analog channels (SL1 to SL4 = 1,
see the Selecting a Conversion Sequence section), thus there are
H
Figure 7 shows the arrangement used. The /S SEL controls a
multiplexer that selects the source of the conversion sequence
information, that is, from the hardware channel select pins (SL1
to SL4) or from the channel selection register. When a conver-
sion begins, the output from the multiplexer is latched until the
end of the conversion sequence. The data bus bits, DB0 to DB3,
(DB0 representing Channel 1 through DB3 representing Channel 4)
are bidirectional and become inputs to the channel select register
EOC
four
pulses and four read operations to access the result of
each of the four conversions.
CONVST
. This
A conversion is initiated on the rising edge of
places all four track-and-holds into hold simultaneously. New
data from this conversion sequence is available for the first
channel selected (VIN1) 1.65 μs later. The conversion on each
subsequent channel is completed at 1.65 μs intervals. The end of
each conversion is indicated by the falling edge of the
signal. The BUSY output signal indicates the end-of-conversion
for all selected channels (four in this case).
RD
CS
WR
when
is logic high and
and
are logic low. The logic
EOC
state on DB0 to DB3 is latched into the channel select register
WR
when
goes logic high.
H/S SEL
Data is read from the part via a 12-bit parallel data bus with
HARDWARE CHANNEL
SELECT PINS
SELECT INDIVIDUAL
TRACK-AND-HOLDS
FOR CONVERSION
CS
RD
CS
RD
standard
and
signals. The
and inputs are
SL1
SL2
SL3
SL4
internally gated to enable the conversion result onto the data
bus. The data lines (DB0 to DB11) leave their high impedance
DATA BUS
D3 D2 D1 D0
SEQUENCER
CS
RD
CS
state when both
permanently tied logic low and the
conversion result. Because each conversion result is latched into
EOC
and
are logic low. Therefore,
can be
CHANNEL SELECT
REGISTER
LATCH
RD
signal used to access the
its output data register prior to
going logic low, another
pins together and use the rising
to latch the conversion result. Although the
WR
TRANSPARENT WHILE WAITING FOR
CONVST. LATCHED ON THE RISING
EDGE OF CONVST AND DURING A
CONVERSION SEQUENCE.
EOC RD
option is to tie the
EOC
and
CS
edge of
WR
AD7864 has some special features that permit reading during a
conversion (such as a separate supply for the output data
drivers, VDRIVE) for optimum performance it is recommended
Figure 7. Channel Select Inputs and Registers
RD
EOC
that the read operation be completed when
is, before the start of the next conversion. Although Figure 10
EOC
is logic low, that
t13
WR
shows the read operation occurring during the
pulse, a
t15
read operation can occur at any time. Figure 10 shows a timing
specification referred to as the quiet time. Quiet time is the
amount of time that should be left after a read operation and
before the next conversion is initiated. The quiet time depends
heavily on data bus capacitance, but 50 ns to 100 ns is typical.
t14
CS
t17
DATA IN
t16
DATA
The signal labeled FRSTDATA (first data-word) indicates to the
user that the pointer associated with the output data registers is
pointing to the first conversion result by going logic high. The
pointer is reset to point to the first data location (that is, the first
conversion result,) at the end of the first conversion (FRSTDATA
Figure 8. Channel Selection via Software Control
Rev. D | Page 15 of 28
AD7864
logic high). The pointer is incremented to point to the next
register (next conversion result) when that conversion result is
available. Thus, FRSTDATA in Figure 9 is shown as going low
is already high when the next conversion sequence initiates. See
the Accessing the Output Data Registers section.
Reading After the Conversion Sequence
EOC
just prior to the second
pulse. Repeated read operations
Figure 10 shows the same conversion sequence as Figure 9. In
this case, however, the results of the four conversions (on VIN1 to
during a conversion continue to access the data at the current
pointer location until the pointer is incremented at the end of
that conversion. Note that FRSTDATA has an indeterminate
logic state after initial power-up. This means that for the first
conversion sequence after power-up, the FRSTDATA logic
output may already be logic high before the end of the first
conversion (this condition is indicated by the dashed line in
Figure 9). Also, the FRSTDATA logic output may already be
high as a result of the previous read sequence, as is the case after
V
IN4) are read after all conversions have finished, that is, when
BUSY goes logic low. The FRSTDATA signal goes logic high at
EOC
the end of the first conversion just prior to
going logic low.
As mentioned previously, FRSTDATA has an indeterminate
state after initial power-up, therefore FRSTDATA may already
be logic high. Unlike the case when reading between each
conversion, the output data register pointer is incremented on
RD
the rising edge of
available. This means FRSTDATA goes logic low after the first
RD
because the next conversion result is
RD
the fourth read in Figure 9. The fourth read (rising edge of
)
resets the pointer to the first data location. Therefore, FRSTDATA
rising edge on
.
tACQ
t1
CONVST
tBUSY
QUIET
TIME
t2
BUSY
tCONV
tCONV
tCONV
tCONV
t8
EOC
t11
t10
FRSTDATA
t12
RD
CS
t4
t3
t5
t7
t6
V
V
V
V
IN4
DATA
IN1
IN2
IN3
100ns
H/S SEL
100ns
SL1 TO SL4
Figure 9. Timing Diagram for Reading During Conversion
t1
CONVST
BUSY
tBUSY
QUIET
TIME
t2
EOC
RD
t8
t3
t4
CS
t7
t6
V
V
V
V
V
DATA
IN1
IN2
IN3
IN4
IN1
t10
t10
FRSTDATA
Figure 10. Timing Diagram, Reading After the Conversion Sequence
Rev. D | Page 16 of 28
AD7864
This means a conversion time of 2.6 μs compared to 1.65 μs
Successive read operations access the remaining conversion
results in an ascending channel order. Each read operation
increments the output data register pointer. The read operation
that accesses the last conversion result causes the output data
register pointer to be reset so that the next read operation accesses
the first conversion result again. This is shown in Figure 10,
wherein the fifth read after BUSY goes low accessing the result
of the conversion on VIN1. Thus, the output data registers act as
a circular buffer in which the conversion results are continually
accessible. The FRSTDATA signal goes high when the first
conversion result is available.
when using the internal clock. In some instances, however, it
may be useful to use an external clock when high throughput
rates are not required. For example, two or more AD7864s can
be synchronized by using the same external clock for all
devices. In this way, there is no latency between output logic
EOC
signals like
due to differences in the frequency of the
internal clock oscillators. Figure 11 shows how the various logic
outputs are synchronized to the CLK signal. Each conversion
requires 14 clocks. The output data register pointer is reset to
point to the first register location on the falling edge of the 12th
clock cycle of the first conversion in the conversion sequence—
see the Accessing the Output Data Registers section. At this
point, the logic output FRSTDATA goes logic high. The result of
the first conversion transfers to the output data registers on the
falling edge of the 13th clock cycle. The FRSTDATA signal is
reset on the falling edge of the 13th clock cycle of the next
conversion, that is, when the result of the second conversion is
transferred to its output data register. As mentioned previously,
CS
Data is enabled onto the data bus (DB0 to DB11) using
and
RD CS RD
. Both
and
have the same functionality as described
in the previous section. There are no restrictions or performance
implications associated with the position of the read operations
after BUSY goes low. The only restriction is that there is minimum
time between read operations. Notice that the quiet time must
be allowed before the start of the next conversion.
RD
the pointer is incremented by the rising edge of the
the result of the next conversion is available. The
signal if
USING AN EXTERNAL CLOCK
EOC
signal
INT
/EXT CLK allows the user to operate the
The logic input
goes logic low on the falling edge of the 13th clock cycle and is
reset high again on the falling edge of the 14th clock cycle.
AD7864 using the internal clock oscillator or an external clock.
To achieve optimum performance on the AD7864, use the internal
clock. The highest external clock frequency allowed is 5 MHz.
1
1
1
9 10 11 12 13 14 2
2
3
4
5
6
7
8
9 10 11 12 13 14
2
3
4
5
6
7
8
13 14
CLK
CONVST
FRSTDATA
EOC
RD
FIRST CONVERSION
COMPLETE
LAST CONVERSION
COMPLETE
BUSY
Figure 11. Using an External Clock
Rev. D | Page 17 of 28
AD7864
STANDBY MODE OPERATION
The AD7864 has a standby mode whereby the device can be
placed in a low current consumption mode (5 μA typical). The
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
STBY
AD7864 is placed in standby by bringing the Logic Input
low. The AD7864 can be powered up again for normal opera-
STBY
+105°C
tion by bringing
logic high. The output data buffers remain
+25°C
operational while the AD7864 is in standby. This means the user
can continue to access the conversion results while the AD7864
is in standby. This feature can be used to reduce the average power
consumption in a system using low throughput rates. To reduce
average power consumption, the AD7864 can be placed in standby
at the end of each conversion sequence, that is, when BUSY
goes low and is taken out of standby again prior to the start of
the next conversion sequence. The time it takes the AD7864 to
come out of standby is referred to as the wake-up time. The
wake-up time limits the maximum throughput rate at which the
AD7864 can be operated when powering down between conver-
sion sequences. The AD7864 wakes up in approximately 2 μs when
using an external reference. The wake-up time is also 2 μs when
the standby time is less than 1 ms while using the internal refer-
ence. Figure 12 shows the wake-up time of the AD7864 for
standby times greater than 1 ms. Note that when the AD7864
is left in standby for periods of time greater than 1 ms, the part
requires more than 2 μs to wake up. For example, after initial
power-up using the internal reference, the AD7864 requires
6 ms to power up. The maximum throughput rate that can be
–40°C
0.0001
0.001
0.01
0.1
1
10
STANDBY TIME (Seconds)
Figure 12. Power-Up Time vs. Standby Time Using the On-Chip Reference
(Decoupled with 0.1 μF Capacitor)
ACCESSING THE OUTPUT DATA REGISTERS
There are four output data registers, one for each of the four
possible conversion results from a conversion sequence. The
result of the first conversion in a conversion sequence is placed
in Register 1, the second result is placed in Register 2, and so
forth. For example, if the conversion sequence VIN1, VIN3, and
V
IN4 is selected (see the Selecting a Conversion Sequence section),
the results of the conversion on VIN1, VIN3, and VIN4 are placed in
Register 1 to Register 3, respectively. The output data register
pointer is reset to point to Register 1 at the end of the first con-
version in the sequence, immediately prior to
At this point, the logic output, FRSTDATA, goes logic high to
indicate that the output data register pointer is addressing Reg-
EOC
going low.
achieved when powering down between conversions is 1/(tBUSY
+
2 μs) = 100 kSPS, approximately. When operating the AD7864 in a
standby mode between conversions, the power savings can be
significant. For example, with a throughput rate of 10 kSPS, the
AD7864 is powered down (IDD = 5 μA) for 90 μs out of every
100 μs (see Figure 13).
CS
RD
ister 1. When
and
are both logic low, the contents of the
addressed register are enabled onto the data bus (DB0 to DB11).
Therefore, the average power consumption drops to
125/10 mW or 12.5 mW approximately.
100µs
tBUSY
CONVST
7µs
tBUSY
BUSY
tWAKE-UP
2µs
I
= 20µA
DD
STBY
Figure 13. Power-Down Between Conversion Sequences
Rev. D | Page 18 of 28
AD7864
When reading the output data registers after a conversion
sequence, that is, when BUSY goes low, the register pointer is
may be used to enable the register contents onto the data bus,
as described in the Reading Between Each Conversion in the
Conversion Sequence subsection within the Selecting a
Conversion Sequence section. The pointer is reset to point
RD
incremented on the rising edge of the
signal, as shown in
Figure 14. However, when reading the conversion results during
the conversion sequence, the pointer is not incremented until a
valid conversion result is in the register to be addressed. In this
case, the pointer is incremented when the conversion has ended
and the result has been transferred to the output data register.
RD
to Register 1 on the rising edge of the
signal when the last
conversion result in the sequence is being read. In the example
shown, this means that the pointer is set to Register 1 when the
contents of Register 3 are read.
EOC
EOC
This happens immediately before
goes low, therefore
FRSTDATA
V
DRIVE
OUTPUT DATA REGISTERS
(V
(V
(V
)
OE NO. 1
OE NO. 2
IN1
IN3
2-BIT
COUNTER
)
OUTPUT
DRIVERS
)
IN4
DB0 TO DB11
OE NO. 3
OE NO. 4
POINTER*
RESET
OE
NOT VALID
AD7864
RD
CS
*THE POINTER IS NOT INCREMENTED BY A RISING EDGE ON RD UNTIL
THE CONVERSION RESULT IS IN THE OUTPUT DATA REGISTER. THE POINTER
IS RESET WHEN THE LAST CONVERSION RESULT IS READ.
Figure 14. Output Data Registers
Rev. D | Page 19 of 28
AD7864
OFFSET AND FULL-SCALE ADJUSTMENT
In most digital signal processing (DSP) applications, offset and
full-scale errors have little or no effect on system performance.
Offset error can always be eliminated in the analog domain by
ac coupling. Full-scale error effect is linear and does not cause
problems as long as the input signal is within the full dynamic
range of the ADC. Invariably, some applications require that the
input signal spans the full analog input dynamic range. In such
applications, offset and full-scale error have to be adjusted to zero.
NEGATIVE FULL-SCALE ADJUST
Apply a voltage of −9.9976 V (−FS + 1/2 LSB) at V1 and adjust
R2 until the ADC output code flickers between 1000 0000 0000
and 1000 0000 0001.
An alternative scheme for adjusting full-scale error in systems
that use an external reference is to adjust the voltage at the VREF
pin until the full-scale error for any of the channels is adjusted
out. Good full-scale matching of the channels ensures small
full-scale errors on the other channels.
Figure 15 shows a circuit that can be used to adjust the offset
and full-scale errors on the AD7864 (VINxA on the AD7864-1
version is shown for example purposes only). Where adjustment
is required, offset error must be adjusted before full-scale error.
This is achieved by trimming the offset of the op amp driving
the analog input of the AD7864 while the input voltage is 1/2
LSB below analog ground. The trim procedure is as follows:
apply a voltage of −2.44 mV (−1/2 LSB) at V1 in Figure 15 and
adjust the op amp offset voltage until the ADC output code
flickers between 1111 1111 1111 and 0000 0000 0000.
INPUT
RANGE = ±10V
V
1
R1
10kΩ
R2
500Ω
V
INxA
R4
AD7864-1*
10kΩ
R5
10kΩ
R3
10kΩ
AGND
Adjust gain error at either the first code transition (ADC
negative full scale) or the last code transition (ADC positive full
scale). The trim procedures for both cases are as follows.
*ADDITIONAL PINS OMITTED FOR CLARITY.
Figure 15. Full-Scale Adjust Circuit
POSITIVE FULL-SCALE ADJUST
Apply a voltage of 9.9927 V (FS − 3/2 LSB) at V1 and adjust R2
until the ADC output code flickers between 0111 1111 1110 and
0111 1111 1111.
Rev. D | Page 20 of 28
AD7864
DYNAMIC SPECIFICATIONS
The AD7864 is specified and 100ꢀ tested for dynamic perfor-
mance specifications as well as traditional dc specifications, such
as integral and differential nonlinearity. These ac specifications are
required for signal processing applications such as phased array
sonar, adaptive filters, and spectrum analysis. These applications
require information on the effect of the ADC on the spectral
content of the input signal. Thus, the parameters for which the
AD7864 is specified include SNR, harmonic distortion, inter-
modulation distortion, and peak harmonics. These terms are
discussed in more detail in the following sections.
graph is 72.6 dB. Note that the harmonics are taken into
account when calculating the SNR.
0
AD7864-1 @ 25°C
5V SUPPLY
–10
SAMPLING AT 499,712Hz
INPUT FREQUENCY OF 99,857Hz
8192 SAMPLES TAKEN
–20
–30
–40
–50
–60
–70
–80
SIGNAL-TO-NOISE RATIO (SNR)
–90
SNR is the measured signal-to-noise ratio at the output of the
ADC. The signal is the rms magnitude of the fundamental.
Noise is the rms sum of all the nonfundamental signals up to
half of the sampling frequency (fS/2) excluding dc. SNR depends
on the number of quantization levels used in the digitization
process; the more levels, the smaller the quantization noise. The
theoretical signal-to-noise ratio for a sine wave input is given by
–100
–110
0
50
100
150
200
250
FREQUENCY (kHz)
Figure 17. FFT Plot
EFFECTIVE NUMBER OF BITS
The formula given in Equation 1 relates the SNR to the number
of bits. Rewriting the formula, as in Equation 2, it is possible to
get a measure of performance expressed in effective number of
bits (N).
SNR = (6.02N + 1.76) dB
(1)
where N is the number of bits.
Thus, for an ideal 12-bit converter, SNR = 74 dB.
SNR −1.76
N =
(2)
Figure 16 shows a histogram plot for 8192 conversions of a dc
input using the AD7864 with a 5 V supply. The analog input was
set at the center of a code. The figure shows that all the codes
appear in the one output bin, indicating very good noise
performance from the ADC.
6.02
The effective number of bits for a device can be calculated
directly from its measured SNR. Figure 18 shows a typical plot
of effective number of bits vs. frequency for an AD7864-2.
12
9000
11
8000
7000
6000
5000
4000
3000
2000
1000
–40°C
10
+25°C
9
8
+105°C
7
6
5
4
0
0
500
1000
1500
2000
2500
3000
1054 1055 1056 1057 1058 1059 1060 1061 1062 1063 1064
FREQUENCY (kHz)
ADC CODE
Figure 18. Effective Numbers of Bits vs. Frequency
Figure 16. Histogram of 8192 Conversions of a DC Input
INTERMODULATION DISTORTION
The output spectrum from the ADC is evaluated by applying a
sine wave signal of very low distortion to the analog input. A
fast fourier transform (FFT) plot is generated from which the
SNR data can be obtained. Figure 17 shows a typical 4096 point
FFT plot of the AD7864 with an input signal of 99.9 kHz and a
sampling frequency of 500 kHz. The SNR obtained from this
With inputs consisting of sine waves at two frequencies, fa and
fb, any active device with nonlinearities creates distortion products
at sum and difference frequencies of mfa nfb where m, n = 0,
1, 2, 3, and so forth. Intermodulation terms are those for which
neither m nor n are equal to zero. For example, the second-order
Rev. D | Page 21 of 28
AD7864
2.5
2.0
terms include (fa + fb) and (fa − fb), whereas the third-order
terms include (2fa + fb), (2fa − fb), (fa + 2fb), and (fa − 2fb).
1.5
Using the CCIF standard where two input frequencies near the
top end of the input bandwidth are used, the second- and third-
order terms are of different significance. The second-order
terms are usually distanced in frequency from the original sine
waves, whereas the third-order terms are usually at a frequency
close to the input frequencies. As a result, the second- and
third-order terms are specified separately. The calculation of the
intermodulation distortion is as per the THD specification
where it is the ratio of the rms sum of the individual distortion
products to the rms amplitude of the fundamental expressed in
decibels. In this case, the input consists of two, equal amplitude,
low distortion sine waves. Figure 19 shows a typical IMD plot
for the AD7864.
1.0
0.5
0
–0.5
–1.0
–1.5
–2.0
–2.5
0
500
1000
1500
2000
2500
3000
3500
4000
ADC CODE
Figure 21. Typical INL Plot
MEASURING APERTURE JITTER
0
AD7864-1 @ 25°C
A convenient way to measure aperture jitter is to use the
relationship it is known to have with SNR (signal-to-noise plus
distortion) given as follows:
5V SUPPLY
SAMPLING AT 131072Hz
–10
INPUT FREQUENCY OF
48,928Hz AND 50,016Hz
–20
4096 SAMPLES TAKEN
–30
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
1
–40
–50
–60
–70
–80
–90
–100
SNRJITTER = 20× log10
(3)
(
2×π × fIN ×σ
)
where:
SNRJITTER is the signal-to-noise due to the rms time jitter.
σ is the rms time jitter.
fIN is the sinusoidal input frequency (1 MHz in this case).
Equation 3 demonstrates that the signal-to-noise ratio due to
jitter degrades significantly with frequency. At low input fre-
quencies, the measured SNR performance of the AD7864 is
indicative of noise performance due to quantization noise and
system noise only (72 dB used as a typical figure in this example).
0
10
20
30
40
50
60
FREQUENCY (kHz)
Figure 19. IMD Plot
AC LINEARITY PLOTS
The plots shown in Figure 20 and Figure 21 show typical DNL
and INL plots for the AD7864.
Therefore, by measuring the overall SNR performance
(including noise due to jitter, system, and quantization) of the
AD7864, a good estimation of the jitter performance of the
AD7864 can be calculated.
3
2
1
12
11
10
9
0
–1
–2
–3
8
7
0
500
1000
1500
2000
2500
3000
3500
4000
6
5
ADC CODE
Figure 20. Typical DNL Plot
900k
950k
1.00M
1.05M
1.10M
FREQUENCY (Hz)
Figure 22. ENOB of the AD7864 at 1 MHz
Rev. D | Page 22 of 28
AD7864
From Figure 22, the ENOB of the AD7864 at 1 MHz is
approximately 11 bits. This is equivalent to 68 dB SNR.
From Equation 3
70.2 dB = 20 × log10[1/(2 × π × 1 MHz × σ)]
SNRTOTAL = SNRJITTER + SNRQUANT = 68 dB
68 dB = SNRJITTER + 72 dB (at 100 kHz)
SNRJITTER = 70.2 dB
σ = 49 ps
where σ is the rms jitter of the AD7864.
Rev. D | Page 23 of 28
AD7864
MICROPROCESSOR INTERFACING
The high speed parallel interface of the AD7864 allows easy
interfacing to most DSPs and microprocessors. This interface
requirements. The following instruction is used to read the
conversion results from the AD7864:
CS RD WR EOC
consists of the data lines (DB0 to DB11),
,
,
,
,
IN D,ADC
and BUSY.
where D is the data memory address and ADC is the AD7864
address.
AD7864 TO ADSP-2100/ADSP-2101/ADSP-2102
INTERFACE
TMS320C5x
Figure 23 shows an interface between the AD7864 and the
A0 TO A13
DS
ADDRESS
DECODE
CONVST
ADSP-210x. The
ADSP-210x or from some other external source. Figure 23
CS DMS
signal can be generated by the
V
CS
RD
IN1
shows the
being generated by a combination of the
V
RD
IN2
V
IN3
WR
WE
signal and the address bus of the ADSP-210x. In this way, the
AD7864 is mapped into the data memory space of the
ADSP-210x.
V
IN4
DB0 TO DB11
D0 TO D15
AD7864
The AD7864 BUSY line provides an interrupt to the ADSP-210x
when the conversion sequence is complete on all the selected
channels. The conversion results can then be read from the
AD7864 using successive read operations. Alternately, one can
BUSY
INTn
PA0
CONVST
Figure 24. AD7864 to TMS320C5x Interface
EOC
use the
pulse to interrupt the ADSP-210x when the
AD7864 TO MC68HC000 INTERFACE
conversion on each channel is complete when reading between
each conversion in the conversion sequence (Figure 9). The
AD7864 is read using the following instruction:
An interface between the AD7864 and the MC68HC000 is
shown in Figure 25. The conversion can be initiated from the
MC68HC000 or from an external source. The AD7864 BUSY
line can be used to interrupt the processor or, alternatively,
software delays can ensure that the conversion has been
completed before a read to the AD7864 is attempted. Because of
the nature of its interrupts, the MC68HC000 requires additional
logic (not shown in Figure 25) to allow it to be interrupted
correctly. For further information on MC68HC000 interrupts,
consult the Addendum to MC68000 Users Manual.
MR0 = DM(ADC)
where MR0 is the ADSP-210x MR0 register and ADC is the
AD7864 address.
ADSP-210x
A0 TO A13
DMS
ADDRESS
DECODE
V
V
V
V
CS
RD
IN1
IN2
IN3
IN4
AS
W
and R/ outputs are used to generate a
The MC68HC000
RD
RD
RD
is used to drive
input to allow the processor to
separate
input signal for the AD7864.
WR
WR
DTACK
the MC68HC000
DB0 TO DB11
D0 TO D24
execute a normal read operation to the AD7864. The conversion
results are read using the following MC68HC000 instruction:
AD7864
BUSY
IRQn
MOVE.W ADC,D0
CONVST
DT1/F0
where D0 is the MC68HC000 D0 register and ADC is the
AD7864 address.
Figure 23. AD7864 to ADSP-210x Interface
AD7864 TO TMS320C5x INTERFACE
Figure 24 shows an interface between the AD7864 and the
TMS320C5x. As with the previous interfaces, conversion can be
initiated from the TMS320C5x or from an external source, and
the processor is interrupted when the conversion sequence is
CS
DS
completed. The
signal and a decode of the address bus. This maps the AD7864
RD
signal to the AD7864 is derived from the
into external data memory. The
signal from the TMS320C5x
is used to enable the ADC data onto the data bus. The AD7864
has a fast parallel bus, consequently there are no wait state
Rev. D | Page 24 of 28
AD7864
Vector control of an ac motor involves controlling phase in
MC68HC000
A0 TO A15
addition to drive and current frequency. Controlling the phase
of the motor requires feedback information on the position of
the rotor relative to the rotating magnetic field in the motor.
Using this information, a vector controller mathematically trans-
forms the three-phase drive currents into separate torque and
flux components. The AD7864, with its 4-channel simultaneous
sampling capability, is ideally suited for use in vector motor
control applications.
ADDRESS
DECODE
V
V
V
V
CS
RD
IN1
IN2
IN3
IN4
DTACK
AS
R/W
AD7864
DB0 TO DB11
D0 TO D15
A block diagram of a vector motor control application using the
AD7864 is shown in Figure 26. The position of the field is derived
by determining the current in each phase of the motor. Only
two phase currents need to be measured because the third can
be calculated if two phases are known. VIN1 and VIN2 of the
AD7864 are used to digitize this information.
CLOCK
CONVST
Figure 25. AD7864 to MC68HC000 Interface
VECTOR MOTOR CONTROL
The current drawn by a motor can be split into two components:
one produces torque and the other produces magnetic flux. For
optimal performance of the motor, control these two compo-
nents independently. In conventional methods of controlling
a three-phase motor, the current (or voltage) supplied to the
motor and the frequency of the drive are the basic control
variables. However, both the torque and flux are functions of
current (or voltage) and frequency. This coupling effect can
reduce the performance of the motor because, for example, if
the torque is increased by increasing the frequency, the flux
tends to decrease.
Simultaneous sampling is critical to maintain the relative phase
information between the two channels. A current sensing isola-
tion amplifier, transformer, or Hall effect sensor is used between
the motor and the AD7864. Rotor information is obtained by
measuring the voltage from two of the inputs to the motor. VIN3
and VIN4 of the AD7864 are used to obtain this information.
Once again, the relative phase of the two channels is important.
A DSP microprocessor is used to perform the mathematical
transformations and control loop calculations on the
information fed back by the AD7864.
DSP MICROPROCESSOR
I
C
DAC
DAC
DAC
TORQUE AND FLUX
CONTROL LOOP
CALCULATIONS AND
TWO TO THREE
DRIVE
CIRCUITRY
V
I
I
B
THREE-
PHASE
MOTOR
B
V
A
A
PHASE INFORMATION
TORQUE
SETPOINT
+
ISOLATION
AMPLIFIERS
+
–
FLUX
SETPOINT
–
V
V
IN1
TRANSFORMATION
TO TORQUE AND
FLUX CURRENT
COMPONENTS
IN2
AD7864*
V
IN3
V
IN4
VOLTAGE
ATTENUATORS
*ADDITIONAL PINS OMITTED FOR CLARITY.
Figure 26. Vector Motor Control Using the AD7864
Rev. D | Page 25 of 28
AD7864
the different channels. The AD7864 has a maximum aperture
delay matching of 4 ns.
MULTIPLE AD7864S IN A SYSTEM
Figure 27 shows a system where a number of AD7864s are
configured to handle multiple input channels. This type of
configuration is common in applications such as sonar and
radar. The AD7864 is specified with maximum limits on
aperture delay match. This means that the user knows the
difference in the sampling instant between all channels. This
allows the user to maintain relative phase information between
All AD7864s use the same external SAR clock (5 MHz).
Therefore, the conversion time for all devices is identical;
consequently, all devices can be read simultaneously. In the
example shown in Figure 27, the data outputs of two AD7864s
EOC
are enabled onto a 32-bit wide data bus when
goes low.
V
V
V
V
EOC
IN1
IN2
IN3
IN4
ADSP-2106x
32
12
AD7864
REF193
V
REF
CS
CLKIN
RD
RD
V
IN1
V
IN2
12
V
IN3
AD7864
V
IN4
V
ADDRESS
DECODE
REF
CS
CLKIN
RD
5MHz
Figure 27. Multiple AD7864s in Multichannel System
Rev. D | Page 26 of 28
AD7864
OUTLINE DIMENSIONS
14.15
13.90 SQ
13.65
1.03
0.88
0.73
2.45
MAX
34
44
1.95 REF
1
33
PIN 1
SEATING
PLANE
10.20
10.00 SQ
9.80
TOP VIEW
(PINS DOWN)
2.20
2.00
1.80
0.23
0.11
23
11
7°
0°
22
12
0.25 MIN
0.10
0.45
0.30
LEAD WIDTH
COPLANARITY
VIEW A
0.80 BSC
LEAD PITCH
VIEW A
ROTATED 90° CCW
COMPLIANT TO JEDEC STANDARDS MO-112-AA-1
Figure 28. 44-Lead Metric Quad Flat Package [MQFP]
(S-44-2)
Dimensions shown in millimeters
ORDERING GUIDE
Relative
Accuracy
Temperature
Range1
Package
Option
Model
AD7864ASZ-12
AD7864ASZ-1REEL2
AD7864BSZ-12
AD7864BSZ-1REEL2
AD7864ASZ-22
AD7864ASZ-2REEL2
AD7864ASZ-32
AD7864ASZ-3REEL2
EVAL-AD7864-2CB3
EVAL-AD7864-3CB3
EVAL-CONTROL BRD24
Input Ranges
5 V, 10 V
5 V, 10 V
5 V, 10 V
5 V, 10 V
0 V to 2.5 V, 0 V to 5 V
0 V to 2.5 V, 0 V to 5 V
2.5 V
Package Description
44-Lead MQFP
44-Lead MQFP
44-Lead MQFP
44-Lead MQFP
44-Lead MQFP
44-Lead MQFP
44-Lead MQFP
44-Lead MQFP
Evaluation Board
Evaluation Board
Controller Board
1 LSB
1 LSB
0.5 LSB
0.5 LSB
1 LSB
1 LSB
1 LSB
1 LSB
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
S-44-2
S-44-2
S-44-2
S-44-2
S-44-2
S-44-2
S-44-2
S-44-2
2.5 V
1 The A version is fully specified up to 105°C with a maximum sample rate of 450 kSPS and IDD maximum (normal mode) of 26 mA.
2 Z = RoHS Compliant Part.
3 This can be used as a stand alone evaluation board or in conjunction with the Evaluation Controller Board for evaluationꢀdemonstration purposes.
4 This board is a complete unit, allowing a PC to control and communicate with all Analog Devices, Inc., evaluation boards ending in the CB designators. To order a
complete evaluation kit, the particular ADC evaluation board needs to be ordered, for example, EVAL-AD7864-1CB, the EVAL-CONTROL BRD2, and a 12 V ac
transformer. See the Evaluation Board application note for more information.
Rev. D | Page 27 of 28
AD7864
NOTES
©1998–2009 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D01341-0-2/09(D)
Rev. D | Page 28 of 28
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