IRU3065CLTRPBF_技术文档

Data Sheet No. PD94703 revA

IRU3065(PbF)

POSITIVE TO NEGATIVE DC TO DC CONTROLLER

PRODUCT DATASHEET

FEATURES

DESCRIPTION

Generate Negative Output from +5V Input

1A Maximum Output Current

The IRU3065 controller is designed to provide solutions

for the applications requiring low power on board switch-

ing regulators. The IRU3065 is specifically designed

for positive to negative conversion and uses few com-

ponents for a simple solution. The IRU3065 operates at

high switching frequency (up to 1.5MHz), resulting in

smaller magnetics. The output voltage can be set by

using an external resistor divider. The stability over all

conditions is inherent with this architecture without any

compensation. The device is available in the standard

6-Pin SOT-23.

1.5MHz maximum Switching Frequency

Few External Components

Available in 6-Pin SOT-23

APPLICATIONS

Hard Disk Drives

Blue Laser for DVD R-W

MR Head Bias

LCD Bias

GaAs FET Bias

Positive-to-Negative Conversion

TYPICAL APPLICATION

5V

C4

D1

10uF

BAT54

VDD

C1 1uF

Vcc

C3

100pF

Q1

V

GATE

IRLML5203

D2

U1

IRU3065

VOUT (-5V)

10BQ015

1.2uH

Gnd

L1

C6

10uF

ISEN

VSEN

R1

0.1

R3

10K

R2

R3

R2

VREF = 5V

VOUT = -VREF ×

10K

Figure 1 - Typical application of IRU3065 for single input voltage.

PACKAGE ORDER INFORMATION

Basic Part (Non Lead-Free)

TA (°C)

DEVICE

PACKAGE

OUTPUT VOLTAGE

0 To 70

IRU3065CLTR

6-Pin SOT-23 (L6)

Adjustable

Lead-Free Part

PACKAGE

TA (°C)

DEVICE

OUTPUT VOLTAGE

0 To 70 IRU3065CLTRPbF 6-Pin SOT-23 (L6)

Adjustable

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1

IRU3065(PbF)

ABSOLUTE MAXIMUM RATINGS

Vcc ......................................................................... 7V

VDD ......................................................................... 12V

Operating Junction Temperature Range ..................... 0°C To 125°C

Operating Ambient Temperature Range ..................... 0°C To 70°C

Storage Temperature Range ...................................... -65°C To +150°C

ESD Capability (Human Body Model) ........................ 2000V

PACKAGE INFORMATION

6-PIN PLASTIC SOT-23 (L6)

TOP VIEW

VGATE 1

6 Vcc

5 VDD

4 I^SEN

Gnd 2

3

VSEN

θJA=230ꢀC/W

ELECTRICAL SPECIFICATIONS

Unless otherwise specified, these specifications apply over Vcc=5V, VDD=7V, CGATE=470pF, RSEN=0.125Ω,

RFDBK1=RFDBK2=10KΩ (to Vcc), fs=1.2MHz, IFL=0.25A and TJ=0°C to 125°C. Typical values refer to TJ=25°C.

PARAMETER

SYM

TEST CONDITION

MIN

4

TYP

MAX

UNITS

Recommended Vcc Supply

Recommended V^DDSupply

Operating Current

Vcc Note.1

V^DD

V

5

4

Icc

mA

3

Initial Output Voltage Accuracy

Measured in application

T^J=25ꢀC, Vout=-5V

Measured in application

over temp. Vout=-5V.

-1%

-2%

1%

Output Accuracy

+2%

Voltage Feedback Sense

Voltage Feedback Input Offset VVoff

Voltage Feedback Bias Current IVBIAS

VVSEN

V

0

mV

µA

mV

µA

-10

10

2

Peak Current Sense Voltage

Min Current Sense Voltage

Current Sense Bias Current

Output Drivers Section

Switching Frequency

Max Output Duty Cycle

Min Output Duty Cycle

Rise Time

VIs

145

50

IIBIAS

2

1.5

0

fs

Dmax

Dmin

Tr

Tf

TD

Note. 1

MHz

%

100

%

10% to 90% Vgate high

ns

40

100

Fall Time

Propagation Delay from

Current Sense to Output

90% to 10% Vgate going low

Vsens=1V. Isens from 0 to

250mV. Delay time between

90% of Isens to 10% of Vgate

ns

Note. 1. guarantted by design

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IRU3065(PbF)

PIN DESCRIPTIONS

PIN# PIN SYMBOL

PIN DESCRIPTION

Output driver for external P Channel MOSFET.

1

2

3

4

5

6

VGATE

Gnd

VSEN

ISEN

This pin serves as ground pin and must be connected to the ground plane.

A resistor divider from this pin to V^OUTand Vcc or an external VREF, sets the output

voltage.

This pin sets the maximum load current by sensing the inductor current.

This pin provides biasing for the output driver.

VDD

Vcc

This pin provides biasing for the internal blocks of the IC.

BLOCK DIAGRAM

Vcc

6

V

DD

5

S

R

Q

1 VGATE

4

3

I

SEN

V

SEN

2

Gnd

Figure 2 - Simplified block diagram of the IRU3065.

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IRU3065(PbF)

APPLICATION INFORMATION

Introduction

The IRU3065 is a controller intended for an inverting When the output current is below a critical current IOCP,

regulator solution. For example, to generate –5V from the output voltage is regulated at the desired value and

a 5V supply. The controller is simple and only has a the switching frequency increases as output current

voltage comparator, current hysteretic comparator, flip- increases. At current IOCP, the switching frequency

flop and MOSFET driver. It controls a typical buck boost reaches its maximum fS(MAX). In this region, the opera-

converter configured by a P-channel MOSFET, an in- tion is in regulation mode. When the current goes above

ductor, a diode and an output capacitor. The sensed IOCP, the operation goes into power limit mode. The out-

inductor current by a sensing resistor compares with put voltage starts to decrease and the output power is

current comparator. The current comparator uses hys- limited. The switching frequency is also reduced.

teresis to control the turn-on and turn-off of the transis-

tor based upon the inductor current and gated by the Analysis shows that the current IOCP is determined by:

output voltage level. When the inductor current rises

VISEN(TH)

VIN

1

2

past the hysteresis set point, the output of the current

comparator goes high. The flip-flop is reset and the P-

channel MOSFET is turned off. In the mean time, the

current sense reference is reduced to near zero, giving

a zero reference threshold voltage level. As the induc-

tor current passes below this threshold, which indicates

that the inductor’s stored energy has been transferred

to the output capacitor, the current comparator output

goes high and turns on the output transistor (if the out-

put voltage is low). By means of hysteresis, the induc-

IOCP =

×

--(1)

Rs

VIN-VOUT(NOM)+VD

Where:

Rs = Current Sensing Resistance

VI^SEN(TH)= Upper Threshold Voltage at the current

comparator (when Vcc=5V, VISEN(TH)=0.145V)

VIN = Input Voltage

VD = Diode Forward Voltage

VOUT(NOM) = Nominal Output Voltage

tor charges and discharges and functions as self oscil- The maximum switching frequency is determined by:

lating. The voltage feedback comparator acts as a de-

VIN×(VD-VOUT(NOM))

mand governor to maintain the output voltage at the de-

sired level.

fS(MAX) =

(VIN+VD-VOUT(NOM)×L×IPEAK

VIN×(VD-VOUT(NOM))×RS

fS(MAX) =

---(2)

By hysteresis control, the maximum switch current (also

equals inductor current) is limited by the internal cur-

rent sensing reference. The power limit is automatically

achieved. The switching frequency is determined by a

combination of factors including the inductance, output

VISEN(TH)×(VIN+VD-VOUT(NOM))×L

Where:

IPEAK = Peak Inductor Current

load current level and peak inductor current. The theo- IPEAK is determined by:

retical output voltage and switching frequency versus

VI^SEN(TH)

IPEAK =

---(3)

output current is shown in Figure 3.

RS

Regulation

Power limit

mode

The detailed operation can be seen in the theoretical

operation section

mode

Output

voltage

V_out

I_out

f_{s max}

Switching

frequency

f_s

I_out

I_ocp

Figure 3 - Theoretical output voltage and switching

frequency vs. output current.

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IRU3065(PbF)

APPLICATION EXAMPLE

The following design example is for the evaluation board

application for IRU3065. The schematic is shown in fig-

ure 1:

Design Example

The modified current IOCP is:

VIN

0.15

1

2

1

2

IOCP =

×

VIN + VD - VOUT(NOM)

RS

5

× 1.5A = 357mA

Where:

5 + 0.5 - (-5)

V^IN= 5V

Output Inductor L

VOUT(NOM) = -5V

IOUT = 200mA

The inductance is chosen by equation (2):

V^IN×(VD - VOUT(NOM))

fS(MAX) = Maximum Frequency

fS(MAX) = 1.2MHz

VD = Diode Forward Voltage

VD = 0.5V

L ꢁ

(VIN+VD-

VOUT(NOM))×fS(MAX)×IPEAK

-(-5 - 0.5)

5

L ꢁ

×

= 1.45µH

(5 - (-5) + 0.5)×1.2MHz

1.5A

Vcc = 5V

Select L = 1.2µH

VISEN(TH)=145mV 150mV

Voltage Sensing Resistor

The maximum inductor current is: IPEAK = 1.5A

The output voltage is determined by the two voltage sens-

ing resistors R2 and R3:

The maximum average inductor current equals

I^AVG=(VISENTH_MAX+VISENTH_MIN)/Rs/2

R3

VOUT(NOM) = -

× VREF

IAVG=(145mV+50mV)/0.1ohm/2=1A

R2

If R3 is chosen as 10K, Then R2 is given by:

MOSFET Selection

A P-channel MOSFET is required. The peak current in

this case is equal to IPEAK=1.5A. The MOSFET

IRLML5203, from international Rectifier with ID=3A and

BVDSS=30V, is a good choice.

V^REF

5V

R2 = -

× R3 = -

× 10K = 10KΩ

VOUT(NOM)

-5V

Current Sensing Resistor R^S

In order to select RS, the desired critical current IOCP

has to be determined. Considering the switching losses, Input Capacitor

for conservative, the critical current should select to be An input capacitor will help to minimize the induced

slightly greater than the nominal output current.

ripple on the +5V supply. A 1µF to 10µF X7R ceramic

capacitor is recommended.

Select:

IOCP = 200mA×1.5 = 300mA

Output Capacitor

Where 1.5 is the coefficient to take the efficiency into An output capacitor is required to store energy from

account.

transfer to the output inductor. Its capacitance and ESR

have a great impact on output voltage ripple. A 10µF to

According to equation (1), the current IOCP is given by: 22µF X7R Tantolum or ceramic capacitor is recom-

mended.

0.15

VIN

1

2

IOCP =

×

= 300mA

RS

VIN - VOUT(NOM) + VD

Output Diode

The current sensing resistance is calculated as:

The average diode current equals output current. In

this case, select the diode average current larger than

300mA. The lowest block voltage is VIN+(-VOUT). In this

case, It is 10V. In order to reduce the switching losses,

the Schottky diode is recommended. The diode 10BQ015

from International Rectifier with ID=1A and VBR=15V is

a good choice.

0.15

VIN

1

2

RS =

×

IOCP

VIN - VOUT(NOM) + VD

0.15

5

1

RS =

×

y0.12Ω

0.3 5 - (-5) + 0.5

2

Select RS = 0.1Ω

Other Components

From equation (3), the modified inductor peak current In order to speed up the turn off of P-channel MOSFET,

is:

a fast diode 1N4148 or a 100ohm resistor and 100pF

capacitor is connected to the pin VDD and VGATE as shown

VISEN(TH)

RS

IPEAK =

= 1.5A

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5

IRU3065(PbF)

in figure 1. The schottky diode can be replaced with a

100Ω resistor (Figure 28.) with a small sacrifice of effi- Demo board Evaluation Results

ciency but lower cost.

Fig.1 shows the evaluation board schematic and the

Thermal Consideration

selected components. The diode D1 can be replaced

with a 100ohm resistor. The measured efficiency ver-

The thermal design is to ensure maximum junction tem- sus load current is shown in Fig. 6. With the boot strap

perature of IRU3065 will not exceed the maximum op- schottky diode, the efficiency is slight higher compar-

eration junction temperature, which is 125ꢀC. The junc- ing with using 100ohm resistor.

tion temperature can be estimated by the following:

If higher efficiency is preferred, lower operation fre-

TJ = PD×ΘJA+TA≤TJ(MAX) = 125ꢀC

quency should be selected. Figure. 5 shows a effi-

ciency curve when 4.7uH inductor is chosen. The maxi-

Where ΘJA is the thermal resistance from junction to mum operation frequency reduces from 800k to 250kHz.

case which is usually provided in the specification. PD As a results, efficiency is more than 10% higher.

is the power dissipation. TA is the ambient temperature.

The package thermal resistance of IRU3065 is estimated

as 230C/W due to compact package.

For the application circuit shown in Fig.1. The mea-

sured output voltage versus output current is shown in

Assuming the maximum allowed ambient temperature Figure 7. When the load current approaches 400mA,

is 70C, the maximum power dissipation of IRU3065 will the output voltage starts to drop and goes into power

be

limit mode. When output is about 1A, the output voltage

will goes almost zero.

P^D<(125-70C)/ΘJA=(125-70)/230=240mW

The measured frequency versus load is listed in Fig-

ure 8. The highest switching frequency occurs at about

For High Power Application

The IR3065 driver is designed to driver PMOS for low 440mA. As load current goes up, the IC goes into

current applications. Figure 4. shows the rise time ver- power limit mode and frequency automatically goes down

sus cap load. For big capacitor load, the rise time is to protect the system.

increasing.

The current sensing comparator threshold voltage ver-

sus VCC is shown in Figure. 9. Since this threshold is

only a divided voltage of VCC, it will changes when

VCC changes. This should be aware in the application.

Rise time versus cap load

90

80

70

60

50

40

30

20

10

0

The output voltage versus Vin=VCC is shown in fig-

ure 11. Since the voltage reference is set by Vin. When

Vin changes, the output voltage will change along Vin.

Sometimes this feature is preferrable since Vout may

want to be tracked with Vin except the polarity. How-

ever, if more accurate output is required, a external

voltage reference should set the output voltage.

For the evaluation board, the measured inductor volt-

age waveforms are listed in Figure 13-17. Figure 15

shows the measured inductor voltage waveform when

output current is 250mA, which the converter is oper-

ated in regulation mode and output voltage is regulated

at desired voltage -5V. Figure 16 shows the measured

inductor voltage waveform when the output current is

equal to the critical current I^OCP. Figure 17 shows the

measured inductor voltage waveform when the output is

in short circuit, which indicates that the converter is in

power limit mode and output voltage is near zero.

0

0.5

1

1.5

2

2.5

Cap(nF)

Fig.4. Rise time versus cap load.

The internal gate driver of IRU3065 is designed for

load current up to 1A. For higher power applications,

external driver is recommended to driver the external

FETs.

.

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6

IRU3065(PbF)

Characteristics of IRU3065

Vout(V) vs Iout(mA)

Efficiency versus load current

6

5

4

3

2

1

0

75

70

65

60

55

50

0

100

200

300

400

Iout(mA)

0

200

400

600

800

1000

Efficiency(%) with diode

Efficiency(%) with 100ohm resistor

Iout(mA)

Figure.5 Efficiency with 4.7uH inductor, 250kHz op-

eration

Fig.7. Output voltage (absolute value) versus load

current. (Vout= -5V, Iocp=400mA)

Efficiency versus load current

Frequency (KHz) versus load current

75

70

65

60

55

50

900

800

700

600

500

400

300

200

100

0

100

200

Iout(mA)

300

400

Efficiency(%) with diode

Efficiency(%) with 100 resistor

0

200

400

600

800

1000

Iout(mA)

Fig.8. Frequency versus load current. (Vout= -5V,

Iocp=400mA).

Figure 6. Efficiency with 1.2uH inductor, 800k

Hz operation

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7

IRU3065(PbF)

Characteristics of IRU3065( Continued)

Output voltage versus

Vin@Iout=200mA, TA=25C

Current comparator threshold versus

A

Vin (T =25C)

-4

-4.2

-4.4

-4.6

-4.8

-5

-5.2

-5.4

-5.6

-5.8

-6

180

4.5

4.7

4.9

5.1

5.3

5.5

170

160

150

140

130

120

4.5

4.7

4.9

5.1

5.3

5.5

Vin

VIn

Figure. 11. Output voltage versus Vin (Vcc=Vin).

Figure 9. Current sensing comparator upper

threshold versus VCC=Vin.

Output voltage versus temperature

(Vin=5V,Iout=200mA)

Current sensing comparator upper

threshold (mV) versus temperature

-5

155

154

153

152

151

150

149

148

147

146

145

144

-25

0

25

50

75

100

125

-5.02

-5.04

-5.06

-5.08

-5.1

-5.12

-5.14

-25

0

25

50

75

100

125

Tempeture(C )

Temperature(C )

Figure. 10. Current sesning comparator upper thresh-

old versus temperature (Vcc=5V)

Figure. 12. Output voltage versus temperature at

Vcc=Vin=5V and Iout=200mA.

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8

IRU3065(PbF)

.

Operation Waveforms of Demo board in Figure. 5

Figure. 13 Start up

Fig. 16. Operation waveform with 450mA, the bound-

ary of continuous mode and discontinuous mode.

The output start out of regulation

Fig. 14. Operation waveform with 20mA load.

Fig. 17. operation waveform with short output.

Fig. 15. Operation waveforms with 250mA load

(normal operation)

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IRU3065(PbF)

THEORETICAL OPERATION

reference of the chip, which is set to be 150mV (for

Vcc=5V), the flip-flop is reset and the PMOS is turned

off. The inductor current is discharged through diode

D2 to the load. The load voltage increases. When the

inductor current decreases to zero, the output current

is supplied by the output capacitor and the output volt-

age decreases until next cycle starts. In this mode, the

voltage at VSEN pin is controlled near zero. Therefore,

the output voltage is regulated at:

Operation-Regulation Mode

Vgate

R3

Vin

-VOUT =

× VREF

R2

Voltage across

the inductor

In the evaluation board, the output voltage is regulated

at -5V, as shown in figure 7. The steady state of the

converter should be operated in this mode. One feature

in this mode is that the shaded inductor current in fig-

ure 18 stays unchanged. The average output diode cur-

rent equals output current. When the switching period

decreases and frequency goes up, the average diode

current increases to support more output current. The

switching frequency increases linearly when the load

current increases as shown in figure 20.

V

−V

D

out

I_peak

Inductor

current

O utputofcurrent

com parator

O utputdiode

current

6

5

I_out

5

4

V

I

3

2

1

0

out out

R 3

R 2

−

V

=

V

ref

out

0.75

0

0.02

0.16

0.32

0.48

0.64

0.8

I

out

t_on

t

1

T_s

Figure 19 - Theoretical output voltage (-V^OUT)

versus output current for IRU3065 controlled buck

boost evaluation board.(assume VIsen=0.2V)

6

5

.

1.5 10

6

.

Figure 18 - Operation waveforms of IRU3065 con-

trolled buck boost converter at regulation mode.

1.083 10

.

1.2 10

.

9 10

In general, IRU3065 controlled buck boost converter

is operated in two modes depending on the load cur-

rent. When the load current is small, the buck boost

operated in first mode (regulation mode). The operation

waveforms are shown in figure 18. In this mode, the

inductor current in the buck converter is discontinuous.

Basic Operation

f

I

s

out

.

6 10

.

3 10

4

.

4.583 10

0

0.2

0.4

0.6

0.8

0.02

I

out

When the voltage at VSEN pin is below zero, the flip-flop

inside the IC is set and the VGATE pin output low, which

trigger the PMOS in the power stage, the output induc-

tor current increases from zero. When sensed inductor

current voltage at ISEN pin reaches the internal current

Figure 20 - Theoretical switching frequency versus

output current for evaluation board.(assume

VIsen=0.2V)

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10

IRU3065(PbF)

Power Limit Mode

Influence of System Parameters

From above section, there is a critical output current

When the output current continuous increases, the

switching period continuous decreases until the induc- IOCP. When the output current is larger than IOCP, the

tor current goes into the boundary of discontinuous and output voltage is out of the regulation and switching fre-

continuous mode as shown in Figure 21. Then the quency starts to decreases. When output current equals

IRU3065 controlled buck boost converter goes into power IOCP, the frequency reaches its maximum fS(MAX). Analy-

limit mode. In this mode, the output power is limited. sis shows that the current IOCP and maximum frequency

The output voltage is no longer regulated. The output fS(MAX) strongly depends on the parameters such as cur-

voltage decreases when the load current increases as rent sensing resistor RS and inductance L as well as the

shown in Figure 19. In this mode, the shaded inductor input and output voltage.

current in Figure 18 keeps same. The turn off time pe-

riod is dependent on the output voltage. When the out-

6

5

put current increases, the output voltage decreases and

5

it takes more time for the inductor current to reset from

.

V

I

,

0.1

Ω

4

3

2

1

0

out out

peak current to zero. Therefore, the turn off period in-

creases. Overall the switching frequency decreases

when load current increases as shown in Figure 20.

.

I

0.11

0.12

Ω

out out

.

I

Ω

out out

0.452

0

0.02

0.1

0.2

0.3

0.4

0.5

0.6

0.7

Vgate

I

out

Figure 22 - Theoretical output voltage versus output

current with different current sensing resistor R^S.

6

.

Vin

1.5 10

6

.

1.3 10

6

5

.

1.2 10

Voltage across

the inductor

.

f

I

,

1

µ

H

s

out

.

9 10

.

1.1

1.2

µH

.

µ

H

6 10

V_out− V_D

.

3 10

I_peak

4

.

4.583 10

0

Inductor

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.02

I

out

current

Figure 23 - Theoretical operation switching frequency

versus output current with different inductance L.

Output of current

comparator

Figure 22 shows the calculated output voltage versus

output current with different current sensing resistor RS.

With different RS, the critical current IOCP varies, and

the power process ability changes. Figure 23 shows

the calculated operation switching frequency versus

output current with different inductance L when RS=0.1Ω.

The inductance L determines the maximum switching

frequency of the buck boost converter.

Output diode

I_out

current

R3

R2

V

ref

− V_out

t₁

t_on

T_s

Figure 21 - Operation waveforms of IRU3065 con-

trolled buck boost converter at power limit mode.

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11

IRU3065(PbF)

Analysis of Operation

The expected switching frequency linearly increases

as output current goes up, as shown in Figure 20.

Regulation Mode

From Figure 18, when the PMOS is on, the inductor

current increases from zero. That is:

Power Limit Mode

When output current continuously increases and

IOUT=IOCP, the converter is in the boundary of regulation

mode and power limit mode with output voltage is regu-

lated to nominal voltage VOUT=VOUT(NOM). As current con-

tinues to increase (IOUT>IOCP), the converter goes into

power limit mode. In this mode, the maximum inductor

current is limited by the internal current reference

VISEN=145mV. Therefore, the turn on time of the PMOS

VIN

IL =

× t

---(4)

L

And the peak current is given by:

VIN

IPEAK =

× tON

---(5)

L

Where tON is the turn on time of the PMOS.

Because the switch is turned off when sensed inductor keeps same as equation (7).

current reaches threshold VISEN, the following equation For turn off time, the inductor current theorectically

holds:

should decrease from IPEAK to zero if the threshold

RS×IPEAK = RS× ^VIN×tON = VISEN=150mV

---(6)

voltage is close to zero , therefore:

L

VI^SEN(TH)

L × IPEAK

VISEN × L

IPEAK =

t1 =

=

---(12)

RS

-(VOUT - VD)

-(VOUT - VD) × RS

The turn on time of the PMOS can be calculated as:

Where V^Dis the forward voltage drop of output di-

ode D2.

L×IPEAK

VISEN×L

RS×VIN

tON =

=

---(7)

VIN

For inductor, by applying voltage and second balance The switching period is given by:

approach, we have:

L × IPEAK

TS = tON + t1 =

+

VIN×tON+(VOUT - VD)×t1 = 0

VIN

-(VOUT - VD)

It can be derived as:

VIN - VOUT + VD

TS = L × IPEAK ×

---(13)

-VIN ×(VOUT - VD)

V^IN×tON

VI^SEN×L

t¹=

=

---(8)

-(VOUT - VD)

-(VOUT - VD)×RS

The combination of equations (12) and (13) result in

Where V^Dis the forward voltage drop of output di- the following:

ode D2.

VIN

t1

=

---(14)

VIN - VOUT + VD

TS

From Figure 18, the average current of output diode

should equals the output current, resulting in:

The output current equals the average diode current,

which is:

1

2

t1

ID(AVG) =

× IPEAK ×

= IOUT

---(9)

1

TS

1

2

t1

×IPEAK×

TS

IOUT =

Where TS is the switching period and fS =

TS

1

2

VISEN

VIN

×

---(15)

Combination of equation (6)(8)(9) results in the rela-

tionship between output current and switching frequency:

RS

VIN - VOUT + VD

Where the peak current is given by equation (6).

2

-RS ×(VOUT - VD)

fS =

×IOUT×2

---(10)

VISEN×VISEN×L

Equation (15) can be rewritten as:

Because at regulation mode, the output voltage is regu-

lated, i.e. VOUT=VOUT(NOM). Then the equation (10) can

be rewritten as:

VI^SEN× VIN

VOUT = VIN + VD -

---(16)

2RS × IOUT

The above equation shows that the output voltage at the

power limit mode is not regulated. It decreases as the

output current increases.

2

-RS ×(VOUT(NOM) - VD)

fS =

×IOUT×2

---(11)

VISEN×VISEN×L

www.irf.com

12

IRU3065(PbF)

When IOUT=IOCP, the output voltage equals nominal volt-

age VOUT=VOUT(NOM). From equation (15),we have

1

VIN

VISEN

I^OCP=

×

---(17)

Vout versus output current

2

VIN - VOUT +VD

RS

6

5

4

3

2

1

0

The above equation is used to select the current sens-

ing resistor RS.

Substitution of equation (16) into equation (13) results

in the relationship between frequency and output cur-

rent, that is

0

0.2

0.4

0.6

0.8

V^IN

2×IOUT

fS =

× 1 -

( )--(18)

Output current (A)

L×IPEAK

IPEAK

Predicted (-Vout)

Measured -Vout

The above equation indicates that the switching fre-

quency decreases when output current increases dur-

ing power limit mode.

Figure 24- The comparison between predicted and

measured output voltage versus output current

When I^OUT=IOCP, the switching frequency reaches its

maximum. Substitution of VOUT=VOUT(NOM) and equation

(6) into equation (13) results in the maximum switching

frequency:

Switching frequency versus output current

V^IN×(VD - VOUT(NOM))

fS(MAX) =

1200

1000

800

600

400

200

0

(VIN + VD -

VOUT(NOM))×L×IPEAK

V^IN×(VD - VOUT(NOM))×RS

fS(MAX) =

---(19)

VISEN×(VIN + VD -

VOUT(NOM))×L

Therefore, the inductance can be selected according

to the maximum desired frequency as shown in the fol-

lowing:

0

0.2

0.4

0.6

0.8

Output current (amp)

Predicted fs(KHz)

Experiment fs (kHz)

V^IN×(VD - VOUT(NOM))

L ꢁ

---(20)

(VIN + VD -

VOUT(NOM))×fS(MAX)×IPEAK

Figure 25 - The comparison between predicted

and measured switching frequency versus

output current

Fig. 24 and Fig.25 shows the theorectical predication

and calculation results for the output voltage and fre-

quency versus output current.

www.irf.com

13

IRU3065(PbF)

Other Applications

5V

100ohm

Vcc

VDD

C1

100pF

Q1

V

GATE

U1

IRU3065

IRLML5203

D2

VOUT (-5V)

10BQ015

Gnd

C2

L1

10uF

1.2uH

ISEN

VSEN

R1

0.1

R3

10K

R2

10K

VREF= 5V

Fig. 26 . IRU3065 application with 100ohm resistor and 100pf cap

www.irf.com

14

IRU3065(PbF)

(L6) SOT-23 Package

B

e

L

E

E1

e1

D

α

C

A2

A

A1

C

L

SYMBOL MIN

MAX

1.45

0.15

1.30

0.50

0.20

3.00

1.75

A

A1

A2

B

C

D

E

E1

e

e1

L

0.90

0.00

0.90

0.35

0.09

2.80

2.60

1.50

0.95 REF

1.90 REF

0.10

0.60

α

0ꢀ

10ꢀ

NOTE: ALL MEASUREMENTS

ARE IN MILLIMETERS.

IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105

TAC Fax: (310) 252-7903

Visit us at www.irf.com for sales contact information

Data and specifications subject to change without notice. 9/6/2005

www.irf.com

15


型号：	IRU3065CLTRPBF
厂家：	Infineon
描述：	POSITIVE TO NEGATIVE DC TO DC CONTROLLER 正到负的直流到直流控制器稳压器开关式稳压器或控制器电源电路开关式控制器光电二极管
文件：	总15页 (文件大小：230K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

IRU3065CLTRPBF [INFINEON]

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