LT5518EUF [Linear]
1.5GHz - 2.4GHz High Linearity Direct Quadrature Modulator; 的1.5GHz - 2.4GHz高线性度直接正交调制器
| 型号： | LT5518EUF |
| 厂家： | 
Linear |
| 描述： | 1.5GHz - 2.4GHz High Linearity Direct Quadrature Modulator |
| 文件： | 总16页 (文件大小：374K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
LT5518
1.5GHz–2.4GHz
High Linearity Direct
Quadrature Modulator
U
DESCRIPTIO
FEATURES
The LT®5518 is a direct I/Q modulator designed for high
performance wireless applications, including wireless
infrastructure. It allows direct modulation of an RF signal
usingdifferentialbasebandIandQsignals.ItsupportsPHS,
GSM, EDGE, TD-SCDMA, CDMA, CDMA2000, W-CDMA
and other systems. It may also be configured as an image
reject up-converting mixer, by applying 90° phase-shifted
signals to the I and Q inputs. The high impedance I/Q
baseband inputs consist of voltage-to-current converters
that in turn drive double-balanced mixers. The outputs of
these mixers are summed and applied to an on-chip RF
transformer, which converts the differential mixer signals
to a 50Ω single-ended output. The balanced I and Q
baseband input ports are intended for DC coupling from a
source with a common mode voltage level of about 2.1V.
The LO path consists of an LO buffer with single-ended
input, and precision quadrature generators that produce
the LO drive for the mixers. The supply voltage range is
4.5V to 5.25V.
■
High Input Impedance Version of the LT5528
■
Direct Conversion to 1.5GHz – 2.4GHz
■
High OIP3: 22.8dBm at 2GHz
■
Low Output Noise Floor at 20MHz Offset:
No RF: –158.2dBm/Hz
P
OUT
= 4dBm: –152.5dBm/Hz
■
■
4-Ch W-CDMA ACPR: –64dBc at 2.14GHz
Integrated LO Buffer and LO Quadrature Phase
Generator
■
■
■
■
50Ω AC-Coupled Single-Ended LO and RF Ports
Low Carrier Leakage: –49dBm at 2GHz
High Image Rejection: 40dB at 2GHz
16-Lead QFN 4mm × 4mm Package
U
APPLICATIO S
■
Infrastructure Tx for DCS, PCS and UMTS Bands
■
Image Reject Up-Converters for DCS, PCS and UMTS
Bands
■
Low Noise Variable Phase-Shifter for 1.5GHz to
, LTC and LT are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
2.4GHz Local Oscillator Signals
U
TYPICAL APPLICATIO
1.5GHz to 2.4GHz Direct Conversion Transmitter Application
with LO Feedthrough and Image Calibration Loop
W-CDMA ACPR, AltCPR and Noise vs RF Output
Power at 2140MHz for 1 and 4 Channels
5V
100nF
×2
V
CC 8, 13
–55
–60
–65
–70
–75
–80
–85
–135
–140
–145
–150
–155
–160
–165
4-CH ACPR
LT5518
14
16
RF = 1.5GHz
TO 2.4GHz
I-DAC
V-I
I-CHANNEL
4-CH ALTCPR
1-CH ACPR
11
PA
0°
1
EN
90°
LO FEEDTHROUGH
CAL OUT
BALUN
Q-CHANNEL
V-I
7
5
Q-DAC
1-CH ALTCPR
IMAGE CAL OUT
1-CH NOISE
CAL
4-CH NOISE
BASEBAND
GENERATOR
3
VCO/SYNTHESIZER
2, 4, 6, 9, 10, 12, 15, 17
DOWNLINK TEST MODEL 64 DPCH
–34 –30 –26 –22 –18 –14 –10
RF OUTPUT POWER PER CARRIER (dBm)
5518 TA01b
ADC
5518 TA01a
5518f
1
LT5518
W W U W
U
W
U
ABSOLUTE AXI U RATI GS
PACKAGE/ORDER I FOR ATIO
(Note 1)
TOP VIEW
ORDER PART
NUMBER
Supply Voltage.........................................................5.5V
Common Mode Level of BBPI, BBMI and
BBPQ, BBMQ .......................................................2.5V
Operating Ambient Temperature
16 15 14 13
LT5518EUF
EN
GND
LO
1
2
3
4
12 GND
11 RF
17
(Note 2) .............................................. –40°C to 85°C
Storage Temperature Range.................. –65°C to 125°C
Voltage on Any Pin
GND
GND
10
9
GND
UF PART
MARKING
5
6
7
8
Not to Exceed...................... –500mV to V + 500mV
CC
5518
T
JMAX
= 125°C, θ = 37°C/W
JA
EXPOSED PAD (PIN 17) IS GND
MUST BE SOLDERED TO THE PCB
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS VCC = 5V, EN = High, TA = 25°C, fLO = 2GHz, fRF = 2.002GHz, PLO = 0dBm.
BBPI, BBMI, BBPQ, BBMQ inputs 2.06VDC, Baseband Input Frequency = 2MHz, I and Q 90° shifted (upper sideband selection).
PRF, OUT = –10dBm, unless otherwise noted. (Note 3)
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
RF Output (RF)
f
RF Frequency Range
RF Frequency Range
–3dB Bandwidth
–1dB Bandwidth
1.5 to 2.4
1.7 to 2.2
GHz
GHz
RF
S
S
RF Output Return Loss
RF Output Return Loss
RF Output Noise Floor
EN = High (Note 6)
EN = Low (Note 6)
–14
–12
dB
dB
22, ON
22, OFF
NFloor
No Input Signal (Note 8)
–158.2
–152.5
–151.1
dBm/Hz
dBm/Hz
dBm/Hz
P
P
= 4dBm (Note 9)
= 4dBm (Note 10)
OUT
OUT
G
G
Conversion Power Gain
Conversion Voltage Gain
Absolute Output Power
3 • LO Conversion Gain Difference
Output 1dB Compression
Output 2nd Order Intercept
Output 3rd Order Intercept
Image Rejection
P
/P , I&Q
10.6
–4
dB
dB
P
OUT IN
20 • log(V
/V
)
V
OUT, 50Ω IN, DIFF, I or Q
P
1V
CW Signal, I and Q
0
dBm
dB
OUT
P-P, DIFF
G
(Note 17)
(Note 7)
–28
8.5
49
3LO vs LO
OP1dB
OIP2
OIP3
IR
dBm
dBm
dBm
dBc
(Notes 13, 14)
(Notes 13, 15)
(Note 16)
22.8
–40
LOFT
Carrier Leakage
(LO Feedthrough)
EN = High, P = 0dBm (Note 16)
–49
–58
dBm
dBm
LO
EN = Low, P = 0dBm (Note 16)
LO
LO Input (LO)
f
LO Frequency Range
LO Input Power
1.5 to 2.4
0
GHz
dBm
dB
LO
P
S
S
–10
5
LO
LO Input Return Loss
LO Input Return Loss
LO Input Referred Noise Figure
LO to RF Small Signal Gain
LO Input Linearity
EN = High (Note 6)
EN = Low (Note 6)
(Note 5) at 2GHz
(Note 5) at 2GHz
(Note 5) at 2GHz
–18
–5
11, ON
11, OFF
dB
NF
14
dB
LO
G
23.8
–9
dB
LO
IIP3
dBm
LO
5518f
2
LT5518
ELECTRICAL CHARACTERISTICS VCC = 5V, EN = High, TA = 25°C, fLO = 2GHz, fRF = 2.002GHz, PLO = 0dBm.
BBPI, BBMI, BBPQ, BBMQ inputs 2.06VDC, Baseband Input Frequency = 2MHz, I and Q 90° shifted (upper sideband selection).
PRF, OUT = –10dBm, unless otherwise noted. (Note 3)
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Baseband Inputs (BBPI, BBMI, BBPQ, BBMQ)
BW
Baseband Bandwidth
–3dB Bandwidth
400
2.06
MHz
V
BB
V
DC Common Mode Voltage
Differential Input Resistance
Common Mode Input Resistance
(Note 4)
CMBB
R
R
Between BBPI and BBMI (or BBPQ and BBMQ)
BBPX and BBMX Shorted Together
2.9
kΩ
Ω
IN, DIFF
IN, CM
105
I
Common Mode Compliance Current Range BBPX and BBMX Shorted Together (Note 18)
–730 to 480
–40
µA
CM, COMP
P
Carrier Feedthrough on BB
Input 1dB Compression Point
I/Q Absolute Gain Imbalance
I/Q Absolute Phase Imbalance
P
= 0 (Note 4)
OUT
dBm
LO2BB
IP1dB
Differential Peak-to-Peak (Note 7)
2.7
V
P-P, DIFF
ΔG
0.06
dB
I/Q
I/Q
Δφ
1
deg
Power Supply (V
)
CC
V
Supply Voltage
4.5
1.0
5
5.25
145
50
V
mA
µA
µs
CC
I
I
t
t
Supply Current
EN = High
128
0.05
0.2
CC, ON
CC, OFF
ON
Supply Current, Sleep Mode
Turn-On Time
EN = 0V
EN = Low to High (Note 11)
EN = High to Low (Note 12)
Turn-Off Time
1.3
µs
OFF
Enable (EN), Low = Off, High = On
Enable
Input High Voltage
Input High Current
EN = High
EN = 5V
V
µA
240
Sleep
Input Low Voltage
EN = Low
0.5
V
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 13: Baseband is driven by 2MHz and 2.1MHz tones. Drive level is set
in such a way that the two resulting RF output tones are –10dBm each.
Note 2: Specifications over the –40°C to 85°C temperature range are
assured by design, characterization and correlation with statistical process
controls.
Note 3: Tests are performed as shown in the configuration of Figure 8.
Note 4: On each of the four baseband inputs BBPI, BBMI, BBPQ and
BBMQ.
Note 14: IM2 measured at LO frequency + 4.1MHz.
Note 15: IM3 measured at LO frequency + 1.9MHz and LO frequency +
2.2MHz.
Note 16: Amplitude average of the characterization data set without image
or LO feedthrough nulling (unadjusted).
Note 17: The difference in conversion gain between the spurious signal at
f = 3 • LO – BB versus the conversion gain at the desired signal at f = LO +
BB for BB = 2MHz and LO = 2GHz.
Note 5: V(BBPI) – V(BBMI) = 1V , V(BBPQ) – V(BBMQ) = 1V
.
DC
DC
Note 6: Maximum value within –1dB bandwidth.
Note 18: Common mode current range where the common mode (CM)
feedback loop biases the part properly. The common mode current is the
sum of the current flowing into the BBPI (or BBPQ) pin and the current
flowing into the BBMI (or BBMQ) pin.
Note 7: An external coupling capacitor is used in the RF output line.
Note 8: At 20MHz offset from the LO signal frequency.
Note 9: At 20MHz offset from the CW signal frequency.
Note 10: At 5MHz offset from the CW signal frequency.
Note 11: RF power is within 10% of final value.
Note 12: RF power is at least 30dB lower than in the ON state.
5518f
3
LT5518
U W
VCC = 5V, EN = High, TA = 25°C, fLO = 2.14GHz,
TYPICAL PERFOR A CE CHARACTERISTICS
P
LO = 0dBm. BBPI, BBMI, BBPQ, BBMQ inputs 2.06VDC, Baseband Input Frequency fBB = 2MHz, I and Q 90° shifted without image or
LO feedthrough nulling. fRF = fBB + fLO (upper sideband selection). PRF, OUT = –10dBm (–10dBm/tone for 2-tone measurements), unless
otherwise noted. (Note 3)
Voltage Gain and Output 1dB
RF Output Power vs LO Frequency
at 1VP-P Differential Baseband Drive
Compression vs LO Frequency
and Temperature
Supply Current vs Supply Voltage
140
130
120
110
100
5
0
15
10
5
4.5V
5.5V
5V
T
= 85°C
A
OP1dB
GAIN
T
= 25°C
A
–5
0
T
= –40°C
A
–5
–10
–15
5V, T = –40°C
A
–10
–15
5V, T = 25°C
A
5V, T = 85°C
A
4.5V, T = 25°C
A
5.5V, T = 25°C
A
5.0
1.3
1.5 1.7 1.9 2.1 2.3 2.5 2.7
1.3
1.5 1.7 1.9 2.1 2.3 2.5 2.7
4.5
5.5
SUPPLY VOLTAGE (V)
LO FREQUENCY (GHz)
LO FREQUENCY (GHz)
5518 G01
5518 G02
5518 G03
Voltage Gain and Output 1dB
Compression vs LO Frequency
and Supply Voltage
Output IP3 and Noise Floor vs LO
Frequency and Temperature
Output IP3 and Noise Floor vs LO
Frequency and Supply Voltage
26
24
22
20
18
16
–146
–148
–150
–152
–154
–156
–158
–160
–162
–164
–166
26
24
22
20
18
16
–146
–148
–150
–152
–154
–156
–158
–160
–162
–164
–166
15
10
OIP3
T
T
T
= –40°C
= 85°C
= 25°C
OIP3
4.5V
5.5V
5V
A
A
A
4.5V
5.5V
5V
OP1dB
f
f
= 2MHz
= 2.1MHz
f
f
= 2MHz
= 2.1MHz
BB, 1
BB, 2
BB, 1
BB, 2
5
0
14 NOISE FLOOR
14 NOISE FLOOR
GAIN
–5
–10
–15
12
10
12
10
NO BASEBAND SIGNAL
= 2.14GHz (FIXED) FOR NOISE
NO BASEBAND SIGNAL
f = 2.14GHz (FIXED) FOR NOISE
LO
8
6
8
6
f
LO
2.1
LO FREQUENCY (GHz)
2.5
2.7
2.1
LO/NOISE FREQUENCY (GHz)
2.5
2.7
2.1
LO/NOISE FREQUENCY (GHz)
2.5
2.7
1.3 1.5
1.7 1.9
2.3
1.3 1.5
1.7 1.9
2.3
1.3 1.5
1.7 1.9
2.3
5518 G04
5518 G05
5518 G06
LO Feedthrough to RF Output vs
LO Frequency
2 • LO Leakage to RF Output vs
2 • LO Frequency
3 • LO Leakage to RF Output vs
3 • LO Frequency
–30
–35
–40
–45
–50
–55
–60
–65
–70
–25
–30
–35
–40
–45
–50
–55
–40
–45
–50
–55
–60
5V, T = –40°C
A
5V, T = 25°C
A
5V, T = 85°C
A
4.5V, T = 25°C
A
5.5V, T = 25°C
A
5V, T = –40°C
5V, T = –40°C
A
A
5V, T = 25°C
5V, T = 25°C
A
A
5V, T = 85°C
5V, T = 85°C
A
A
4.5V, T = 25°C
4.5V, T = 25°C
A
A
5.5V, T = 25°C
5.5V, T = 25°C
A
A
4.5
5.1 5.7 6.3 6.9
8.1
2.1
LO FREQUENCY (GHz)
2.5
2.7
3.9
7.5
1.3 1.5
1.7 1.9
2.3
4.2
2 • LO FREQUENCY (GHz)
5.0
5.4
2.6 3.0
3.4 3.8
4.6
3 • LO FREQUENCY (GHz)
5518 G09
5518 G07
5518 G08
5518f
4
LT5518
U W
TYPICAL PERFOR A CE CHARACTERISTICS
VCC = 5V, EN = High, TA = 25°C, fLO = 2.14GHz,
P
LO = 0dBm. BBPI, BBMI, BBPQ, BBMQ inputs 2.06VDC, Baseband Input Frequency fBB = 2MHz, I and Q 90° shifted without image
or LO feedthrough nulling. fRF = fBB + fLO (upper sideband selection). PRF, OUT = –10dBm (–10dBm/tone for 2-tone measurements),
unless otherwise noted. (Note 3)
Absolute I/Q Gain Imbalance
vs LO Frequency
Absolute I/Q Phase Imbalance
vs LO Frequency
Image Rejection vs LO Frequency
–25
–30
–35
–40
–45
–50
–55
02
0.1
0
5
4
3
2
1
0
5V, T = –40°C
5V, T = –40°C
A
A
5V, T = 25°C
5V, T = 25°C
A
A
5V, T = 85°C
5V, T = 85°C
A
A
4.5V, T = 25°C
4.5V, T = 25°C
A
A
5.5V, T = 25°C
5.5V, T = 25°C
A
A
5V, T = –40°C
A
5V, T = 25°C
A
5V, T = 85°C
A
4.5V, T = 25°C
A
5.5V, T = 25°C
A
1.5 1.7 1.9 2.1
2.7
1.5 1.7 1.9 2.1
2.7
1.5 1.7 1.9 2.1
LO FREQUENCY (GHz)
2.7
1.3
2.3 2.5
1.3
2.3 2.5
1.3
2.3 2.5
LO FREQUENCY (GHz)
LO FREQUENCY (GHz)
5518 G10
5518 G11
5518 G12
RF CW Output Power, HD2 and HD3 vs
Baseband Voltage and Temperature
Voltage Gain vs LO Power
Output IP3 vs LO Power
–2
–4
24
22
20
18
16
14
12
10
8
–10
–20
–30
–40
–50
–60
–70
10
0
RF
HD3
HD2
–6
T
T
T
= –40°C
= 85°C
= 25°C
A
A
A
–10
–8
–10
–12
–14
–16
–18
–20
–30
–40
–50
5V, T = –40°C
5V, T = –40°C
A
A
5V, T = 25°C
5V, T = 25°C
HD2 = MAX POWER AT
+ 2 • f OR f – 2 • f
A
A
A
5V, T = 85°C
5V, T = 85°C
f
A
LO
BB
LO
BB
4.5V, T = 25°C
4.5V, T = 25°C
HD3 = MAX POWER AT
A
A
A
A
6
5.5V, T = 25°C
5.5V, T = 25°C
f
+ 3 • f OR f – 3 • f
LO
BB
LO
BB
5
4
–16 –12 –8 –4
0
8
–16 –12 –8 –4
0
8
–20
4
–20
4
0
2
3
4
1
I AND Q BASEBAND VOLTAGE (V
)
LO INPUT POWER (dBm)
LO INPUT POWER (dBm)
P-P, DIFF
5518 G13
5518 G14
5518 G15
RF CW Output Power, HD2 and
HD3 vs Baseband Voltage and
Supply Voltage
LO Feedthrough to RF Output and
Image Rejection vs Baseband
Voltage and Temperature
LO Feedthrough to RF Output and
Image Rejection vs Baseband
Voltage and Supply Voltage
–10
–20
–30
–40
–50
–60
–70
10
–25
–25
–30
–35
–40
–45
–50
–55
T
T
T
= –40°C
= 85°C
= 25°C
A
A
A
4.5V
LO FT
5.5V
5V
LO FT
0
–30
–35
–40
–45
–50
–55
RF
HD3
4.5V
5.5V
5V
–10
–20
–30
–40
–50
HD2
IR
HD2 = MAX POWER AT
IR
f
+ 2 • f OR f – 2 • f
LO
BB LO
BB
HD3 = MAX POWER AT
f
LO
+ 3 • f OR f – 3 • f
BB
LO
BB
5
0
2
3
4
0
2
3
4
5
0
2
3
4
5
1
1
1
I AND Q BASEBAND VOLTAGE (V
)
I AND Q BASEBAND VOLTAGE (V
)
I AND Q BASEBAND VOLTAGE (V
)
P-P, DIFF
P-P, DIFF
P-P, DIFF
5518 G16
5518 G17
5518 G18
5518f
5
LT5518
U W
VCC = 5V, EN = High, TA = 25°C, fLO = 2.14GHz,
TYPICAL PERFOR A CE CHARACTERISTICS
P
LO = 0dBm. BBPI, BBMI, BBPQ, BBMQ inputs 2.06VDC, Baseband Input Frequency fBB = 2MHz, I and Q 90° shifted without image
or LO feedthrough nulling. fRF = fBB + fLO (upper sideband selection). PRF, OUT = –10dBm (–10dBm/tone for 2-tone measurements),
unless otherwise noted. (Note 3)
LO and RF Port Return Loss
vs RF Frequency
Output IP2 vs LO Frequency
65
60
55
50
45
40
35
0
–10
–20
–30
–40
–50
f
f
f
= 2MHz
BB,1
BB,2
LO PORT, EN = LOW
= 2.1MHz
= f
+ f
+ f
IM2 BB,1 BB,2 LO
LO PORT,
EN = HIGH
RF PORT, EN =
HIGH, NO LO
RF PORT, EN = HIGH,
5V, T = –40°C
A
P
LO
= 0dBm
RF PORT,
EN = LOW
5V, T = 25°C
A
5V, T = 85°C
A
4.5V, T = 25°C
5.5V, T = 25°C
A
A
2.1
2.5
2.7
2.1
RF FREQUENCY (GHz)
2.5
2.7
1.3 1.5
1.7 1.9
2.3
1.3 1.5
1.7 1.9
2.3
LO FREQUENCY (GHz)
5518 G19
5518 G20
U
U
U
PI FU CTIO S
EN (Pin 1): Enable Input. When the enable pin voltage is
higher than 1V, the IC is turned on. When the input voltage
is less than 0.5V, the IC is turned off.
biased at about 2.06V. Applied common mode voltage
must stay below 2.5V.
V
CC
(Pins 8, 13): Power Supply. Pins 8 and 13 are con-
GND (Pins 2, 4, 6, 9, 10, 12, 15): Ground. Pins 6, 9, 15
and17(exposedpad)areconnectedtoeachotherinternally.
Pins 2 and 4 are connected to each other internally and
function as the ground return for the LO signal. Pins 10
and 12 are connected to each other internally and function
as the ground return for the on-chip RF balun. For best RF
performance, pins 2, 4, 6, 9, 10, 12, 15 and the Exposed
Pad (Pin 17) should be connected to the printed circuit
board ground plane.
nected to each other internally. It is recommended to use
0.1µF capacitors for decoupling to ground on each of
these pins.
RF (Pin 11): RF Output. The RF output is an AC-coupled
single-ended output with approximately 50Ω output im-
pedance at RF frequencies. Externally applied DC voltage
should be within the range –0.5V to V + 0.5V in order
CC
to avoid turning on ESD protection diodes.
BBPI,BBMI(Pins14,16):BasebandInputsfortheI-Chan-
nel, with 2.9kΩ Differential Input Impedance. Internally
biased at about 2.06V. Applied common mode voltage
must stay below 2.5V.
LO(Pin3):LOInput.TheLOinputisanAC-coupledsingle-
ended input with approximately 50Ω input impedance at
RF frequencies. Externally applied DC voltage should be
within the range –0.5V to V + 0.5V in order to avoid
CC
Exposed Pad (Pin 17): Ground. This pin must be soldered
to the printed circuit board ground plane.
turning on ESD protection diodes.
BBPQ,BBMQ(Pins7,5):BasebandInputsfortheQ-Chan-
nel, with 2.9kΩ Differential Input Impedance. Internally
5518f
6
LT5518
W
BLOCK DIAGRA
V
CC
8
13
LT5518
BBPI 14
BBMI 16
V-I
V-I
11 RF
0°
90°
BALUN
BBPQ
BBMQ
7
5
1
EN
2
4
6
9
3
10
12
15
17
5518 BD
GND
LO
GND
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APPLICATIO S I FOR ATIO
The LT5518 consists of I and Q input differential voltage-
to-current converters, I and Q up-conversion mixers, an
RF output balun, an LO quadrature phase generator and
LO buffers.
in an RF output balun, which also transforms the output
impedance to 50Ω. The center frequency of the resulting
RF signal is equal to the LO signal frequency. The LO in-
put drives a phase shifter which splits the LO signal into
in-phase and quadrature LO signals. These LO signals
are then applied to on-chip buffers which drive the up-
conversion mixers. Both the LO input and RF output are
single-ended, 50Ω-matched and AC coupled.
External I and Q baseband signals are applied to the dif-
ferential baseband input pins, BBPI, BBMI, and BBPQ,
BBMQ.Thesevoltagesignalsareconvertedtocurrentsand
translated to RF frequency by means of double-balanced
up-converting mixers. The mixer outputs are combined
LT5518
RF
= 5V
C
V
CC
BALUN
FROM
Q
LOMI
LOPI
200Ω
BBPI
V
= 500mV
CM
REF
1.3k
1.3k
1.8pF
1.8pF
200Ω
BBMI
5518 F01
GND
Figure 1. Simplified Circuit Schematic of the LT5518 (Only I-Half is Drawn)
5518f
7
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Baseband Interface
DAC’s differential output current to minimize the LO to RF
feedthrough. Resistors R3A, R3B, R4A and R4B translate
The baseband inputs (BBPI, BBMI), (BBPQ, BBMQ) pres-
ent a differential input impedance of about 2.9kΩ. At each
of the four baseband inputs, a lowpass filter using 200Ω
and 1.8pF to ground is incorporated (see Figure 1), which
limits the baseband bandwidth to approximately 250MHz
the DAC’s output common mode level from about 0.5V
DC
to the LT5518’s input at about 2.06V . For these resis-
DC
tors, 1% accuracy is recommended. For different ambi-
ent temperatures, the LT5518 input common mode level
varies with a temperature coefficient of about –2.7mV/°C.
The internal common mode feedback loop will correct
these level changes in order to bias the LT5518 at the
correct operating point. Resistors R3 and R4 are chosen
high enough that the LT5518 common mode compliance
current value will not be exceeded at the inputs of the
LT5518 as a result of temperature shifts. Capacitors C4A
and C4B minimize the input signal attenuation caused by
the network R3A, R3B, R4A and R4B. This results in a
gaindifferencebetweenlowfrequencyandhighfrequency
baseband signals. The high frequency baseband –3dB
corner point is approximately given by:
(–1dB point). The common mode voltage is about 2.06V
o
and is slightly temperature dependent. At T = –40 C, the
A
o
common mode voltage is about 2.19V and at T = 85 C
A
it is about 1.92V.
If the I/Q signals are DC-coupled to the LT5518, it is
important that the applied common mode voltage level
of the I and Q inputs is about 2.06V in order to properly
bias the LT5518. Some I/Q test generators allow setting
thecommonmodevoltageindependently. Inthiscase, the
common mode voltage of those generators must be set
to 1.03V to match the LT5518 internal bias, because for
DC signals, there is no –6dB source-load voltage division
(see Figure 2).
f
= 1/[2π • C4A • (R3A||R4A||(R
/2)]
–3dB
IN, DIFF
In this example, f
= 58kHz.
–3dB
50Ω
50Ω
1.5k
Thiscornerpointshouldbesetsignificantlylowerthanthe
minimum baseband signal frequency by choosing large
enough capacitors C4A and C4B. For signal frequencies
1.03V
2.06V
DC
DC
+
+
+
2.06V
DC
2.06V
2.06V
DC
DC
–
–
–
50Ω
significantly lower than f
proximately
, the gain is reduced by ap-
GENERATOR
GENERATOR
LT5518
–3dB
5518 F02
Figure 2. DC Voltage Levels for a Generator Programmed at
1.03VDC for a 50Ω Load and for the LT5518 as a Load
G
DC
= 20 • log [R3A||(R
/2)]/[R3A||(R
/2)
IN, DIFF
IN, DIFF
+ R4A]
In this example, G = –11dB.
The LT5518 should be driven differentially; otherwise, the
even-order distortion products will degrade the overall
linearityseverely. Typically, aDACwillbethesignalsource
for the LT5518. A reconstruction filter should be placed
between the DAC output and the LT5518’s baseband
inputs. DC coupling between the DAC outputs and the
LT5518 baseband inputs is recommended. Active level
shifters may be required to adapt the common mode level
of the DAC outputs to the common mode input voltage
of the LT5518. It is also possible to achieve a DC level
shift with passive components, depending on the appli-
cation. For example, if flat frequency response to DC is
not required, then the interface circuit of Figure 3 may be
used. This figure shows a commonly used 0mA – 20mA
DAC output followed by a passive 5th order lowpass
filter. The DC-coupled interface allows adjustment of the
DC
Inserting the network of R3A, R3B, R4A, R4B, C4A and
C4B has the following consequences:
• Reduced LO feedthrough adjustment range. LO to RF
feedthroughcanbereducedbyadjustingthedifferential
DC offset voltage applied to the I and/or Q inputs. Be-
causeoftheDCgainreduction,therangeofadjustment
is reduced. The resolution of the offset adjustment is
improved by the same gain reduction factor.
• DC notch for uneven number of channels. The interface
drawn in Figure 3 might not be practical for an uneven
number of channels, since the gain at DC is lower and
will appear in the center of (one of) the channel(s). In
thatcase,aDC-coupledlevelshiftingcircuitisrequired,
or the LT5528 might be a better solution.
5518f
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• Introduction of a (low frequency) time constant dur-
ing startup. For TDMA-like systems the time constant
introduced by C4A and C4B can cause some delay
during start-up. The associated time constant is ap-
will increase the voltage on the DAC output by dumping
an extra current into resistors R1A, R1B, R2A and R2B.
This current is about (V – V )/(R3A + R4A) = (5
CC
DAC
– 0.5)/(3.01k + 5.63k) = 0.52mA. Maximum impedance
proximately given by T = R
• (C4A + C4B). In
to ground will then be V
/(I
+ I ) = 1.25/0.02052
D
IN, CM
COMPL MAX LS
this example it will result in a delay of about T = 105
= 60.9Ω.
D
• 6.6n = 693ns.
4. Reflection of out-of-band baseband signal power. DAC
outputsignalcomponentshigherthanthecut-offfrequency
of the lowpass filter will not see R2A and R2B as load
resistors and therefore will see only R1A, R1B and the
filter components as a load. Therefore, it is important to
start the lowpass filter with a capacitor (C1), in order to
shunttheDAChigherfrequencycomponentsandthereby,
limit the required extra voltage headroom.
ThemaximumsinusoidalsinglesidebandRFoutputpower
is about 5.5dBm for a full 0mA to 20mA DAC swing.
This maximum RF output level is usually limited by the
compliance voltage range of the DAC (V
) which is
COMPL
assumed here to be 1.25V. When the DAC output voltage
swingislargerthanthiscompliancevoltage,thebaseband
signal will distort and linearity requirements usually will
not be met. The following situations can cause the DAC’s
compliance voltage limit to be exceeded:
The LT5518’s output 1dB compression point is about
8.5dBm, and with the interface network described above,
a common DAC cannot drive the part into compression.
However, it is possible to increase the driving capability
by using a negative supply voltage. For example, if a –1V
supply is available, resistors R1A, R1B, R2A and R2B
can be made 200Ω each and connected with one side to
the –1V supply instead of ground. Typically, the voltage
compliancerangeoftheDACis–1Vto1.25V, sotheDAC’s
outputvoltagewillstaywithinthisrange.Almost6dBextra
voltage swing is available, thus enabling the DAC to drive
the LT5518 beyond its 1dB compression point. Resistors
R3A, R3B, R4A, R4B and the lowpass filter components
must be adjusted for this case.
1. Too high DAC load impedance. If the DC impedance to
ground is higher than V
/I
= 1.25/0.02 = 62.5Ω,
COMPL MAX
thecompliancevoltageisexceededforafullDACswing.In
Figure3, two100Ωresistorsinparallelareused, resulting
in a DC impedance to ground of 50Ω.
2. Too much DC offset. In some DACs, an additional DC
offset current can be set. For example, if the maximum
offset current is set to I
/8 = 2.5mA, then the maxi-
MAX
mum DC DAC load impedance to ground is reduced to
/I • (1 + 1/8) = 1.25/0.0225 = 55Ω.
V
COMPL MAX
3. DC shift caused by R3A, R3B, R4A and R4B if used. The
DC shift network consisting of R3A, R3B, R4A and R4B
LT5518
RF = 5.5dBm, MAX
V
CC
C
5V
BALUN
FROM
Q
LOMI
C4A
3.3nF
LOPI
L1A
L2A
0.53V
200Ω
DC
BBPI
2.1V
V
REF
= 500mV
CM
R4A
3.01k
DC
0mA TO 20mA
0mA TO 20mA
R1A
100Ω
R2A
100Ω
R3A
1.3k
1.3k
1.8pF
5.63k
DAC
C1
C2
C3
R1B
100Ω
R2B
100Ω
R3B
5.63k
1.8pF
R4B
3.01k
L1B
L2B
200Ω
BBMI
2.1V
0.53V
DC
DC
5518 F03
GND
GND
C4B
3.3nF
Figure 3. LT5518 5th Order Filtered Baseband Interface with Common DAC (Only I-Channel is Shown)
5518f
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LT5518
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V
CC
Some DACs use an output common mode voltage of 3.3V.
In that case, the interface circuit drawn in Figure 4 may be
used. The performance is very similar to the performance
of the DAC interface drawn in Figure 3, since the source
and load impedances of the lowpass ladder filter are both
200Ω differential and the current drive is the same. There
are some small differences:
20pF
LO
INPUT
Z
IN
≈ 57Ω
5518 F05
Figure 5. Equivalent Circuit Schematic of the LO Input
• Thebasebanddrivecapabilitycannotbeimprovedusing
an extra supply voltage, since the compliance range of
the DACs in Figure 4 is typically 3.3V – 0.5V to 3.3V +
0.5V, so its range has already been fully used.
significantly below 1.8GHz or above 2.4GHz, the quadra-
ture accuracy will diminish, causing the image rejection
to degrade. The LO pin input impedance is about 50Ω,
and the recommended LO input power is 0dBm. For lower
LO input power, the gain, OIP2, OIP3 and dynamic range
• G and f
are a little different, since R3A (and R3B)
–3dB
DC
is 4.99k instead of 5.6k to accommodate the proper
will degrade, especially below –5dBm and at T = 85°C.
A
DC level shift.
For high LO input power (e.g. 5dBm), the LO feedthrough
will increase, without improvement in linearity or gain.
HarmonicspresentontheLOsignalcandegradetheimage
rejection,becausetheyintroduceasmallexcessphaseshift
in the internal phase splitter. For the second (at 4GHz) and
third harmonics (at 6GHz) at –20dBc level, the introduced
signal at the image frequency is about –55dBc or lower,
corresponding to an excess phase shift much less than 1
degree. For the second and third harmonics at –10dBc,
still the introduced signal at the image frequency is about
–46dBc. Higher harmonics than the third will have less
impact. The LO return loss typically will be better than
14dB over the 1.7GHz to 2.4GHz range. Table 1 shows
the LO port input impedance vs frequency.
LO Section
The internal LO input amplifier performs single-ended to
differential conversion of the LO input signal. Figure 5
shows the equivalent circuit schematic of the LO input.
The internal, differential LO signal is split into in-phase
and quadrature (90° phase shifted) signals that drive LO
buffer sections. These buffers drive the double balanced I
andQmixers.ThephaserelationshipbetweentheLOinput
and the internal in-phase LO and quadrature LO signals
is fixed, and is independent of start-up conditions. The
phase shifters are designed to deliver accurate quadrature
signals for an LO frequency near 2GHz. For frequencies
LT5518
RF = 5.5dBm, MAX
V
CC
C
5V
BALUN
FROM
Q
LOMI
C4A
3.3nF
LOPI
3.3V
0mA TO
20mA
L1A
L2A
3.3V
DC
200Ω
BBPI
2.1V
V
REF
= 500mV
CM
R4A
3.01k
DC
R3A
1.3k
1.3k
1.8pF
4.99k
DAC
C1
C2
C3
GND
R3B
4.99k
1.8pF
R4B
3.01k
L1B
L2B
200Ω
BBMI
0mA TO
20mA
2.1V
3.3V
DC
DC
5518 F04
GND
C4B
3.3nF
Figure 4. LT5518 5th Order Filtered Baseband Interface with 3.3VCM DAC (Only I-Channel is Shown).
5518f
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Table 1. LO Port Input Impedance vs Frequency for EN = High
The RF output S with no LO power applied is given in
22
Frequency
Input Impedance
S
11
Table 4.
MHz
Ω
Mag
Angle
95
Table 4. RF Port Output Impedance vs Frequency for EN = High
and No LO Power Applied
1000
1400
1600
1800
2000
2200
2400
2600
44.5 + j18.2
60.3 + j6.8
62.8 – j0.6
62.4 – j9.0
56.7 – j15.6
50.9 – j16.5
46.6 – j15.2
42.9 – j13.9
0.197
0.112
0.113
0.136
0.157
0.161
0.159
0.165
30
Frequency
Input Impedance
S
22
–2.4
–32
–58
–77
–94
–109
MHz
Ω
Mag
Angle
153
119
102
103
154
160
152
144
1000
1400
1600
1800
2000
2200
2400
2600
21.7 + j9.9
32.3 + j19.5
42.2 + j18.5
46.8 + j9.6
41.8 + j3.7
36.1 + j4.3
32.8 + j7.4
31.2 + j10.5
0.416
0.312
0.214
0.104
0.098
0.170
0.226
0.264
The input impedance of the LO port is different if the part
is in shut-down mode. The LO input impedance for EN =
Low is given in Table 2.
Table 2. LO Port Input Impedance vs Frequency for EN = Low
For EN = Low the S is given in Table 5.
22
Table 5. RF Port Output Impedance vs Frequency for EN = Low
Frequency
Input Impedance
S
11
MHz
Ω
Mag
Angle
75
15
–11
–33
–53
–70
–87
–104
Frequency
Input Impedance
S
22
1000
1400
1600
1800
2000
2200
2400
2600
42.1 + j43.7
121 + j34.9
134 – j31.6
91.3 – j68.5
56.4 – j66.3
37.7 – j54.9
27.9 – j43.6
22.1 – j33.9
0.439
0.454
0.483
0.510
0.532
0.544
0.550
0.553
MHz
Ω
Mag
Angle
154
1000
1400
1600
1800
2000
2200
2400
2600
20.9+j9.6
28.5 + j20.2
36.7 + j24.5
48.7 + j23.1
55.7 + j11.0
48.9 + j0.6
39.8 – j0.02
34.2 + j3.2
0.428
0.365
0.311
0.229
0.116
0.013
0.115
0.193
123
103
80.2
56.7
158.9
–179
167
RF Section
Afterup-conversion,theRFoutputsoftheIandQmixersare
combined. An on-chip balun performs internal differential
tosingle-endedoutputconversion,whiletransformingthe
output signal impedance to 50Ω. Table 3 shows the RF
port output impedance vs frequency.
To improve S for lower frequencies, a shunt capacitor
22
can be added to the RF output. At higher frequencies, a
shunt inductor can improve the S . Figure 6 shows the
22
equivalent circuit schematic of the RF output.
V
CC
Table 3. RF Port Output Impedance vs Frequency for EN = High
and PLO = 0dBm
20pF
RF
OUTPUT
Frequency
Input Impedance
S
22
MHz
Ω
Mag
Angle
153
21pF 3nH
52.5Ω
1000
1400
1600
1800
2000
2200
2400
2600
21.3 + j9.7
29.8 + j20.3
39.1 + j23.5
50.8 + j18.4
52.1 + j5.4
43.2 – j0.1
36.0 + j2.0
32.1 + j5.6
0.421
0.348
0.280
0.180
0.057
0.073
0.164
0.228
5518 F06
121
100
Figure 6. Equivalent Circuit Schematic of the RF Output
77.1
65.5
–179
171
Note that an ESD diode is connected internally from
the RF output to ground. For strong output RF signal
levels (higher than 3dBm) this ESD diode can degrade
the linearity performance if the 50Ω termination imped-
159
ance is connected directly to ground. To prevent this, a
5518f
11
LT5518
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J2
APPLICATIO S I FOR ATIO
coupling capacitor can be inserted in the RF output line.
This is strongly recommended during a 1dB compression
measurement.
BBIM
BBIP
V
CC
C2
16
BBMI GND BBPI
EN
15
14
13
R1
100nF
V
CC
GND
100
1
2
3
4
12
11
10
9
V
EN
J3
CC
RF
OUT
GND
LO
RF
GND
GND
GND
J4
LO
IN
LT5518
Enable Interface
GND
17
Figure 7 shows a simplified schematic of the EN pin inter-
face. The voltage necessary to turn on the LT5518 is 1.0V.
To disable (shutdown) the chip, the Enable voltage must
be below 0.5V. If the EN pin is not connected, the chip is
disabled. This EN = Low condition is guaranteed by the
75kΩ on-chip pull-down resistor. It is important that the
BBMQ GND BBPQ
V
CC
5
6
7
8
C1
100nF
J5
BBQM
J6
GND
BBQP
BOARD NUMBER: DC729A
5518 F08
voltage at the EN pin does not exceed V by more than
Figure 8. Evaluation Board Circuit Schematic
CC
0.5V.Ifthisshouldoccur,thefullchipsupplycurrentcould
be sourced through the EN pin ESD protection diodes.
Damage to the chip may result.
R3
3.01k
R4
3.01k
J1
J2
BBIM
BBIP
R5
52.3Ω
R6
52.3Ω
C1
3.3nF
C2
R2
5.62k
V
CC
E2
V
CC
3.3nF
R1
5.62k
BOARD NUMBER: DC831A
EN
C3
16
BBMI GND BBPI
EN
15
14
13
R7
100nF
V
CC
GND
75k
25k
100
1
2
3
4
12
11
10
9
V
EN
J3
CC
E1
RF
OUT
GND
LO
RF
GND
GND
GND
J4
LO
IN
LT5518
GND
5518 F07
17
BBMQ GND BBPQ
V
CC
5
6
7
8
Figure 7. EN Pin Interface
R10
3.01k
C4
100nF
J5
R9
5.62k
BBQM
R8
5.62k
R12
52.3Ω
Evaluation and Demo Boards
C5
3.3nF
R11
3.01k
J6
Figure 8 shows the schematic of the evaluation board that
was used for the measurements summarized in the Elec-
trical Characteristics tables and the Typical Performance
Characteristic plots.
BBQP
GND
E4
GND
E3
R13
52.3Ω
C6
3.3nF
5518 F09
Figure 9. Demo Board Circuit Schematic
Figure 9 shows the demo board schematic. Resistors R3,
R4, R10 and R11 may be replaced by shorting wires if a
flat frequency response to DC is required. A good ground
connection is required for the exposed pad of the LT5518
package. If this is not done properly, the RF performance
will degrade. The exposed pad also provides heat sink-
ing for the part and minimizes the possibility of the chip
overheating. R7 (optional) limits the Enable pin current in
the event that the Enable pin is pulled high while the V
CC
inputs are low. In Figures 10, 11 and 12 the silk screen
and the demo board PCB layouts are shown. If improved
LOandImagesuppressionisrequired, anLOfeedthrough
calibration and an Image suppression calibration can be
performed.
Figure 10. Component Side Silk Screen of Demo Board
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Figure 11. Component Side Layout of Demo Board
Figure 12. Bottom Side Layout of Demo Board
5V
100nF
×2
V
CC 8, 13
LT5518
14
RF = 1.5GHz
TO 2.4GHz
I-DAC
V-I
I-CHANNEL
16
11
PA
0°
1
EN
90°
BALUN
LO FEEDTHROUGH CAL OUT
IMAGE CAL OUT
Q-CHANNEL
V-I
7
5
Q-DAC
CAL
BASEBAND
GENERATOR
3
VCO/SYNTHESIZER
2, 4, 6, 9, 10, 12, 15, 17
ADC
5518 F13
Figure 13. 1.5GHz to 2.4GHz Direct Conversion Transmitter
Application with LO Feedthrough and Image Calibration Loop
Application Measurements
low, the ACPR will be limited by the noise performance of
the part. In the middle, an optimum ACPR is obtained.
TheLT5518isrecommendedforbase-stationapplications
using various modulation formats. Figure 13 shows a
typical application. The CAL box in Figure 13 allows for
LO feedthrough and Image suppression calibration. Fig-
ure 14 shows the ACPR performance for W-CDMA using
one or four channel modulation. Figures 15, 16 and 17
illustrate the 1, 2 and 4-channel W-CDMA measurement.
To calculate ACPR, a correction is made for the spectrum
analyzer noise floor.
Because of the LT5518’s very high dynamic range, the test
equipment can limit the accuracy of the ACPR measure-
ment. Consult the factory for advice on ACPR measure-
ment, if needed.
TheACPRperformanceissensitivetotheamplitudematch
of the BBIP and BBIM (or BBQP and BBQM) input voltage.
This is because a difference in AC voltage amplitude will
giverisetoadifferenceinamplitudebetweentheeven-order
harmonic products generated in the internal V-I converter.
If the output power is high, the ACPR will be limited by the
linearity performance of the part. If the output power is
As a result, they will not cancel out entirely. Therefore, it
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APPLICATIO S I FOR ATIO
is important to keep the amplitudes at the BBIP and BBIM
inputs (or BBQP and BBQM) as equal as possible.
and Image Rejection can also change as function of the
baseband drive level, as is depicted in Figure 19. The RF
output power, IM2 and IM3 vs two-tone baseband drive
level are given in Figure 20.
When the temperature is changed after calibration, the LO
feedthrough and the Image Rejection performance will
change.ThisisillustratedinFigure18.TheLOfeedthrough
–30
–55
–60
–65
–70
–75
–80
–85
–135
–140
–145
–150
–155
–160
–165
DOWNLINK TEST
4-CH ACPR
MODEL 64 DPCH
–40
–50
4-CH ALTCPR
1-CH ACPR
–60
–70
–80
UNCORRECTED
SPECTRUM
–90
1-CH ALTCPR
CORRECTED
SPECTRUM
1-CH NOISE
–100
–110
–120
4-CH NOISE
SYSTEM NOISE FLOOR
DOWNLINK TEST MODEL 64 DPCH
–34 –30 –26 –22 –18 –14 –10
RF OUTPUT POWER PER CARRIER (dBm)
2127.5 2132.5 2137.5
2152.5
2142.5 2147.5
RF FREQUENCY (MHz)
5518 F15
5518 F14
Figure 14. W-CDMA ACPR, ALTCPR and Noise vs
RF Output Power at 2140MHz for 1 and 4 Channels
Figure 15. 1-Channel W-CDMA Spectrum
–30
–40
–40
–40
DOWNLINK TEST
CALIBRATED WITH P = –30dBm
RF
UNCORRECTED
SPECTRUM
DOWNLINK
TEST
MODEL 64
DPCH
MODEL 64 DPCH
–45
–50
–55
–60
–65
–70
–75
–80
–85
–90
–50
–60
IMAGE REJECTION
–50
–60
–70
–70
–80
CORRECTED
SPECTRUM
–80
LO FEEDTHROUGH
–90
UNCORRECTED
SPECTRUM
–90
–100
–110
–120
–130
CORRECTED
SPECTRUM
–100
–110
–120
SYSTEM NOISE FLOOR
SYSTEM NOISE FLOOR
2125 2130 2135 2140 2145 2150 2155
2115
2125
2135
2145
2155
2165
–40
–20
0
20
40
60
80
RF FREQUENCY (MHz)
RF FREQUENCY (MHz)
TEMPERATURE (°C)
5518 F16
5518 F17
5518 F18
Figure 16. 2-Channel W-CDMA
Spectrum
Figure 17. 4-Channel W-CDMA
Spectrum
Figure 18. LO Feedthrough and
Image Rejection vs Temperature
after Calibration at 25°C
5518f
14
LT5518
U
W U U
APPLICATIO S I FOR ATIO
20
10
0
P
RF
RF
10
0
IM3
–10
–20
–30
–40
–50
–60
–70
–80
–90
–10
–20
–30
–40
–50
–60
–70
–80
–90
LO FT
IM2
f
f
P
f
= 2MHz, 2.1MHz, 0°
BBI
BBQ
LO
= 2MHz, 2.1MHz, 90°
= 0dBm
= f + f
f
f
P
f
= 2MHz, 0°
= 2MHz, 90°
= 0dBm
RF BB LO
BBI
BBQ
f
LO
= 2.14GHz
IR
IM2 = POWER AT f + 4.1MHz
LO
LO
= f + f
IM3 = MAX POWER AT
RF BB LO
T
A
T
A
T
A
= –40°C
= 85°C
= 25°C
T
T
T
= –40°C
= 85°C
= 25°C
A
A
A
f
V
+ 1.9MHz or f + 2.2MHz
f
= 2.14GHz
LO
LO
CC
LO
= 5V
V
= 5V
CC
EN = HIGH
EN = HIGH
0
1
2
3
4
5
0.1
1
10
5518 F20
I AND Q BASEBAND VOLTAGE (V
)
I AND Q BASEBAND VOLTAGE (V
)
P-P, DIFF, EACH TONE
P-P, DIFF
5518 F19
Figure 19. Image Rejection and LO Feedthrough
vs Baseband Drive Voltage After Calibration at 25°C
and VBBI = 0.2VP-P, DIFF
Figure 20. RF Two-Tone Power, IM2 and
IM3 at 2140MHz vs Baseband Voltage
U
PACKAGE DESCRIPTIO
UF Package
16-Lead Plastic QFN (4mm × 4mm)
(Reference LTC DWG # 05-08-1692)
NOTE:
0.72 0.05
1. DRAWING CONFORMS TO JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGC)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
4.35 0.05
2.90 0.05
2.15 0.05
(4 SIDES)
PACKAGE OUTLINE
0.30 0.05
0.65 BSC
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
BOTTOM VIEW—EXPOSED PAD
PIN 1 NOTCH R = 0.20 TYP
R = 0.115
OR 0.35 × 45° CHAMFER
0.75 0.05
4.00 0.10
(4 SIDES)
TYP
15
16
0.55 0.20
PIN 1
TOP MARK
(NOTE 6)
1
2
2.15 0.10
(4-SIDES)
(UF16) QFN 10-04
0.200 REF
0.30 0.05
0.65 BSC
0.00 – 0.05
5518f
InformationfurnishedbyLinearTechnologyCorporationisbelievedtobeaccurateandreliable.However,
no responsibility is assumed for its use. Linear Technology Corporation makes no representation that
the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT5518
RELATED PARTS
PART NUMBER
Infrastructure
LT5511
DESCRIPTION
COMMENTS
High Linearity Up-Converting Mixer
RF Output to 3GHz, 17dBm IIP3, Integrated LO Buffer
DC to 3GHz, 17dBm IIP3, Integrated LO Buffer
LT5512
DC-3GHz High Signal Level Down-Converting Mixer
LT5514
Ultralow Distortion, IF Amplifier/ADC Driver with
Digitally Controlled Gain
850MHz Bandwidth, 47dBm OIP3 at 100MHz,
10.5dB to 33dB Gain Control Range
LT5515
LT5516
LT5517
LT5519
1.5GHz to 2.5GHz Direct Conversion Quadrature Demodulator
0.8GHz to 1.5GHz Direct Conversion Quadrature Demodulator
40MHz to 900MHz Quadrature Demodulator
20dBm IIP3, Integrated LO Quadrature Generator
21.5dBm IIP3, Integrated LO Quadrature Generator
21dBm IIP3, Integrated LO Quadrature Generator
0.7GHz to 1.4GHz High Linearity Up-Converting Mixer
17.1dBm IIP3 at 1GHz, Integrated RF Output Transformer with 50Ω
Matching, Single-Ended LO and RF Ports Operation
LT5520
LT5521
LT5522
LT5524
LT5525
LT5526
LT5528
1.3GHz to 2.3GHz High Linearity Up-Converting Mixer
10MHz to 3700MHz High Linearity Up-Converting Mixer
600MHz to 2.7GHz High Signal Level Down-Converting Mixer
15.9dBm IIP3 at 1.9GHz, Integrated RF Output Transformer with
50Ω Matching, Single-Ended LO and RF Ports Operation
24.2dBm IIP3 at 1.95GHz, NF = 12.5dB, 3.15V to 5.25V Supply,
Single-Ended LO Port Operation
4.5V to 5.25V Supply, 25dBm IIP3 at 900MHz, NF = 12.5dB, 50Ω
Single-Ended RF and LO Ports
Low Power, Low Distortion ADC Driver with
Digitally Programmable Gain
450MHz Bandwidth, 40dBm OIP3, 4.5dB to 27dB Gain Control
High Linearity, Low Power Downconverting Mixer
High Linearity, Low Power Downconverting Mixer
1.5GHz – 2.4GHz High Linearity Direct Quadrature Modulator
Single-Ended 50Ω RF and LO Ports, 17.6dBm IIP3 at 1900MHz,
I
= 28mA
CC
3V to 5.3V Supply, 16.5dBm IIP3, 100kHz to 2GHz RF, NF = 11dB,
= 28mA
I
CC
4.5V to 5.25V Supply, 22dBm OIP3 at 2GHz, NFloor = 159dBm/Hz,
50Ω Single-Ended BB, RF and LO Ports
RF Power Detectors
LT5504
800MHz to 2.7GHz RF Measuring Receiver
80dB Dynamic Range, Temperature Compensated,
2.7V to 5.25V Supply
LTC®5505
LTC5507
LTC5508
LTC5509
LTC5530
LTC5531
LTC5532
LT5534
RF Power Detectors with >40dB Dynamic Range
100kHz to 1000MHz RF Power Detector
300MHz to 3GHz, Temperature Compensated, 2.7V to 6V Supply
100kHz to 1GHz, Temperature Compensated, 2.7V to 6V Supply
44dB Dynamic Range, Temperature Compensated, SC70 Package
36dB Dynamic Range, Low Power Consumption, SC70 Package
300MHz to 7GHz RF Power Detector
300MHz to 3GHz RF Power Detector
300MHz to 7GHz Precision RF Power Detector
300MHz to 7GHz Precision RF Power Detector
300MHz to 7GHz Precision RF Power Detector
50MHz to 3GHz RF Power Detector with 60dB Dynamic Range
Precision V
Precision V
Precision V
Offset Control, Shutdown, Adjustable Gain
Offset Control, Shutdown, Adjustable Offset
Offset Control, Adjustable Gain and Offset
OUT
OUT
OUT
1dB Output Variation Overtemperature, 38ns Response Time
Low Voltage RF Building Block
LT5546 500MHz Quadrature Demodulator with VGA
and 17MHz Baseband Bandwidth
Wide Bandwidth ADCs
17MHz Baseband Bandwidth, 40MHz to 500MHz IF,
1.8V to 5.25V Supply, –7dB to 56dB Linear Power Gain
LT1749
LT1750
12-Bit, 80Msps
500MHz BW S/H, 71.8dB SNR
500MHz BW S/H, 75.5dB SNR
14-Bit, 80Msps
5518f
LT/TP 0205 1K • PRINTED IN USA
LinearTechnology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
16
●
●
(408) 432-1900 FAX: (408) 434-0507 www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2005
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