LM363H-10 [NSC]
LM363 Precision Instrumentation Amplifier; LM363精密仪表放大器
| 型号： | LM363H-10 |
| 厂家： | 
National Semiconductor |
| 描述： | LM363 Precision Instrumentation Amplifier |
| 文件： | 总22页 (文件大小：452K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
April 1991
LM363 Precision Instrumentation Amplifier
General Description
The LM363 is a monolithic true instrumentation amplifier. It
requires no external parts for fixed gains of 10, 100 and
1000. High precision is attained by on-chip trimming of off-
set voltage and gain. A super-beta bipolar input stage gives
very low input bias current and voltage noise, extremely low
offset voltage drift, and high common-mode rejection ratio.
A two-stage amplifier design yields an open loop gain of
10,000,000 and a gain bandwidth product of 30 MHz, yet
remains stable for all closed loop gains. The LM363 oper-
eliminate bandwidth loss due to cable capacitance. Com-
pensation pins allow overcompensation to reduce band-
width and output noise, or to provide greater stability with
capacitive loads. Separate output force, sense and refer-
ence pins permit gains between 10 and 10,000 to be pro-
grammed using external resistors.
On the 8-pin metal can package, gain is internally set at 10,
100 or 500 but may be increased with external resistors.
The shield driver and offset adjust pins are omitted on the
8-pin versions.
g
g
18V with only
ates with supply voltages from
1.5 mA current drain.
5V to
The LM363 is rated for 0 C to 70 C.
§
§
The LM363’s low voltage noise, low offset voltage and off-
set voltage drift make it ideal for amplifying low-level, low-
impedance transducers. At the same time, its low bias cur-
rent and high input impedance (both common-mode and
differential) provide excellent performance at high imped-
ance levels. These features, along with its ultra-high com-
mon-mode rejection, allow the LM363 to be used in the
most demanding instrumentation amplifier applications, re-
placing expensive hybrid, module or multi-chip designs. Be-
cause the LM363 is internally trimmed, precision external
resistors and their associated errors are eliminated.
Features
Y
Offset and gain pretrimmed
e
Y
Y
Y
Y
Y
Y
Y
12 nV/ Hz input noise (G 500/1000)
0
2 nA bias current typical
e
130 dB CMRR typical (G 500/1000)
No external parts required
Dual shield drivers
Can be used as a high performance op amp
Low supply current (1.5 mA typ)
The 16-pin dual-in-line package provides pin-strappable
gains of 10, 100 or 1000. Its twin differential shield drivers
Typical Connections
16-Pin Package
8-Pin Package
e
G
10 2, 3, 4, open
100 3–4 shorted
000 2–4 shorted
TL/H/5609–33
TL/H/5609–1
Connection Diagrams
Metal Can Package
16-Pin Dual-In-Line Package
Order Number LM363H-10,
LM363H-100 or LM363H-500
See NS Package Number H08C
TL/H/5609–2
Order Number 363D
See NS Package Number D16C
C
1995 National Semiconductor Corporation
TL/H/5609
RRD-B30M115/Printed in U. S. A.
Absolute Maximum Ratings (Note 5)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales
Office/Distributors for availability and specifications.
Input Voltage
Equal to Supply Voltage
g
Reference and Sense Voltage
Lead Temp. (Soldering, 10 sec.)
ESD rating to be determined.
25V
300 C
§
g
g
Supply Voltage
18V
10V
Differential Input Voltage
Input Current
g
20 mA
LM363 Electrical Characteristics (Notes 1 and 2)
LM363
Tested
Design
Limit
Parameter
Conditions
Units
Typ
Limit
(Note 3)
(Note 4)
FIXED GAIN (8-PIN)
e
e
e
Input Offset Voltage
Input Offset Voltage Drift
Gain Error
G
G
G
500
100
10
30
50
150
250
2.5
400
700
6
mV
mV
mV
0.5
e
e
e
G
G
G
500
100
10
1
2
4
8
mV/ C
§
mV/ C
§
20
75
mV/ C
§
e
e
e
G
G
G
500
100
10
0.1
0.8
0.7
0.6
0.9
0.8
0.7
%
%
%
g
(
10V Swing, 2 kX Load)
0.07
0.05
PROGRAMMABLE GAIN (16-PIN)
e
e
e
Input Offset Voltage
G
G
G
1000
100
10
50
100
1
250
450
3.5
500
900
8
mV
mV
mV
e
e
e
Input Offset Voltage Drift
Gain Error
G
G
G
1000
100
10
1
2
5
mV/ C
§
10
mV/ C
§
10
100
mV/ C
§
e
e
e
G
G
G
1000
100
10
2.0
0.1
0.6
3.0
0.7
2.0
3.5
0.8
2.3
%
%
%
g
(
10V Swing, 2 kX Load)
FIXED GAIN AND PROGRAMMABLE
e
e
e
Gain Temperature Coefficient
G
G
G
1000
500
40
20
10
ppm/ C
§
ppm/ C
§
100, 10
ppm/ C
§
e
e
Gain Non-Linearity
G
G
10, 100
0.01
0.01
0.03
0.05
0.04
0.06
%
%
g
(
10V Swing, 2 kX Load)
500, 1000
2
LM363 Electrical Characteristics (Continued) (Notes 1 and 2)
LM363
Tested
Design
Limit
Parameter
Conditions
Units
Typ
Limit
(Note 3)
(Note 4)
e
e
e
Common-Mode Rejection
s
10V)
G
G
G
1000, 500
100
130
120
105
114
94
104
84
dB
dB
dB
s
b
Ratio ( 10V
V
CM
10
90
80
e
e
e
Positive Supply Rejection
Ratio (5V to 15V)
G
G
G
1000, 500
100
130
120
100
110
100
85
100
95
dB
dB
dB
10
78
e
e
e
Negative Supply Rejection
G
G
G
1000, 500
100
120
106
86
100
85
90
75
60
dB
dB
dB
b
b
Ratio ( 5V to 15V)
10
70
Input Bias Current
2
1
10
3
20
5
nA
nA
Input Offset Current
Common-Mode Input
Resistance
100
8
GX
e
e
e
Differential Mode Input
Resistance
G
G
G
1000, 500
100
0.2
2
GX
GX
GX
10
20
s
s
13V
b
Input Offset Current Change
11V
V
CM
20
50
100
300
pa/V
Reference and Sense
Resistance
kX
kX
kX
Min
30
80
27
83
Max
e
Open Loop Gain
Supply Current
G
1000, 500
10
1
V/mV
CL
Positive
1.2
1.6
2.4
2.8
3.0
3.4
mA
mA
Negative
b
Note 1: These conditions apply unless otherwise noted; Va 15V, V
15V, V
CM
e
eb
e
e
0V, R 2 kX, reference pin grounded, sense pin connected to output and
L
e
T
j
25 C.
§
Note 2: Boldface limits are guaranteed over full temperature range. Operating ambient temperature range is 0 C to 70 C for the LM363.
§
§
Note 3: Guaranteed and 100% production tested.
Note 4: Guaranteed but not 100% tested. These limits are not used in determining outgoing quality levels.
Note 5: Maximum rated junction temperature is 100 C for the LM363. Thermal resistance, junction to ambient, is 150 C/W for the TO-99(H) package and 100 C/W
§
§
§
for the ceramic DIP (D).
3
e
Typical Performance Characteristics T 25 C
§
A
Fixed Gain and Programmable
Parameter
Units
1000/500
100
18
10
90
Input Voltage Noise, rms, 1 kHz
Input Voltage Noise (Note 6)
Input Current Noise, rms, 1 kHz
Input Current Noise (Note 6)
Bandwidth
12
0.4
0.2
40
30
1
nV/
S
Hz
1.5
0.2
40
10
mVp-p
0.2
40
pA/
S
pAp-p
kHz
V/ms
ms
Hz
100
0.36
25
200
0.24
20
Slew Rate
Settling Time, 0.1% of 10V
Offset Voltage Warm-Up Drift (Note 7)
Offset Voltage Stability (Note 8)
Gain Stability (Note 8)
70
5
15
50
mV
5
10
100
0.05
mV
0.01
0.005
%
Note 6: Measured for 100 seconds in a 0.01 Hz to 10 Hz bandwidth.
b
Note 7: Measured for 5 minutes in still air, Va 15V, V
e
eb
15V. Warm-up drift is proportionally reduced at lower supply voltages.
Common-Mode Input
Voltage Limit
Supply Current vs Supply
Voltage
Input Bias Current vs
Temperature
TL/H/5609–3
4
Typical Performance Characteristics (Continued)
Output Current Limit
Input Noise Voltage
Input Current Noise
TL/H/5609–4
5
Typical Performance Characteristics (Continued)
CMRR with Balanced
Source Resistance
CMRR with Balanced
Source Resistance
CMRR with Balanced
Source Resistance
TL/H/5609–5
6
Typical Performance Characteristics (Continued)
Shield Driver Bias Voltage
Shield Driver Loading Error
Shield Driver Loading Error
TL/H/5609–6
7
Simplified Schematic (pin numbers in parentheses are for 8-pin package)
TL/H/5609–7
Theory of Operation
Referring to the Simplified Schematic, it can be seen that
the input voltage is applied across the bases of Q1 and Q2
This voltage divided by the attenuation factor
R4 R2
e
and appears between their emitters. If R
is the resist-
a a
R3 R4 R1 R2
E1-2
ance across these emitters, a differential current equal to
/R flows from Q1’s emitter to Q2’s. The second
is equal to the output-to-reference voltage. Hence, the over-
all gain is given by
V
IN E1-2
stage amplifier shown maintains Q1 and Q2 at equal collec-
tor currents by negative feedback to Q4. The emitter cur-
rents of Q3 and Q4 must therefore be unbalanced by an
a
R3 R4
V
R
OUT
E3-4
E1-2
e
e
c
G
.
V
R4
R
IN
amount equal to the current flow across R
. Defining
E1-2
R5 R6, the differential voltage across the emitters
e
a
R
E3-4
of Q4 to Q3 is equal to
V
IN
c
R
E3-4
.
R
E 1-2
8
Application Hints
The LM363 was designed to be as simple to use as possi-
ble, but several general precautions must be taken. The dif-
ferential inputs are directly coupled and need a return path
to power supply common. Worst-case bias currents are only
10 nA for the LM363, so the return impedance can be as
high as 100 MX. Ground drops between signal return and IC
supply common should not be ignored. While the LM363
has excellent common-mode rejection, signals must remain
within the proper common-mode range for this specification
to apply. Operating common-mode range is guaranteed
overdrives these diodes conduct, greatly increasing input
currents. This behavior is illustrated in the I vs V plot in
the Typical Performance Characteristics. (The graph is not
symmetrical because at large input currents a portion of the
current into the device flows out the Vb terminal.)
IN IN
The input protection resistors allow a full 10V differential
e
input voltage without degradation even at G 1000. At input
voltages more than one diode drop below Vb or two diode
drops above Va input, current increases rapidly. Diode
clamps to the supplies, or external resistors to limit current
to 20 mA, will prevent damage to the device.
b
a
from 10V to 10V with 15V supplies.
g
The high-gain (500 or 1000) versions have large gain-band-
width products (15 MHz or 30 MHz) so board layout is fairly
critical. The differential input leads should be kept away
from output force and sense leads, especially at high imped-
ances. Only 1 pF from output to positive input at 100 kX
source impedance can cause oscillations. The gain adjust
leads on the 16-pin package should be treated as inputs
and kept away from the output wiring.
REFERENCE AND SENSE INPUTS
The equivalent circuit is shown in the schematic diagram.
Limitations for correct operation are as follows. Maximum
differential swing between reference and sense pins is typi-
g
g
cally 15V ( 10V guaranteed). If this limit is exceeded, the
sense pin no longer controls the output, which then pegs
high or low. The negative common-mode limit is 1.5V below
Vb. (This is permissible because R2 and R4 are returned to
a node biased higher than Vb.) If largepositive voltages are
applied to the reference and sense pins, the common-mode
range of the signal inputs begins to suffer as the drop
POWER SUPPLY
g
The LM363 may be powered from split supplies from 5V
to 18V (or single-ended supplies from 10V to 36V). Posi-
g
g
across R13 and R16 increases. For example, at 15V sup-
tive supply current is typically 1.2 mA independent of supply
voltage. The negative supply current is higher than the posi-
tive by the current drawn through the voltage dividers for the
reference and sense inputs (typ 600 mA total). The LM363’s
excellent PSRR often makes regulated supplies unneces-
sary. Actually, supply voltage can be as low as 7V total but
PSRR is severely degraded, so that well-regulated supplies
are recommended below 10V total. Split supplies need not
be balanced; output swing and input common-mode range
will simply not be symmetrical with unbalanced supplies. For
e
a
e
plies, V
b
V
0V, signal input range is typically
REF
12V to
SENSE
13.5V. At V
e
e
V 15V, signal input
SENSE
REF
b
a
range drops to 11V to 13.5V. The reference and sense
pins can be as much as 10V above Va as long as a restrict-
b
ed signal common-mode range ( 10V min) can be tolerat-
ed.
g
For maximum bipolar output swing at 15V supplies, the
reference pin should be returned to a voltage close to
ground. At lower supply voltages, the reference pin need
not be halfway between the supplies for maximum output
a
b
example, at 12V and 5V supplies, input common-mode
range is typically 10.5V to 2V and output swing is 11V
a
b
a
swing. For example, at Va
12V and Vb
5V,
e
a
grounding the reference pin still allows a
e b
b
to 4V.
a
b
11V to 4V
When using ultra-low offset versions, best results are ob-
g
swing. For single-supply systems, the reference pin can be
tied to either supply if a single output polarity is all that is
required. For a bipolar input and output, create a low imped-
ance reference with an op amp and voltage divider or a
regulator (e.g., LM336, LM385, LM317L). This forms the ref-
erence for all succeeding signal-processing stages. (Don’t
connect the reference terminal directly to a voltage divider;
this degrades gain error.) See Figure 1.
tained at 15V supplies. For example, the LM363-500’s off-
g
set voltage is guaranteed within 150 mV at 15V at 25 C.
§
Running at 5V results in a worst-case negative PSRR er-
g
b
6
b
b
ror of 10V ( 15V to 5V) multiplied by 3.2X10
(110 dB)
or 32 mV, increasing the worst-case offset. Positive PSRR
results in another 10 mV worst-case change.
INPUTS
The LM363 input circuitry is depicted in the Simplified Sche-
matic. The input stage is run relatively rich (50 mA) for low
voltage noise and wide bandwidth; super-beta transistors
and bias-current cancellation (not shown) keep bias cur-
rents low. Due to the bias-current cancellation circuitry, bias
current may be either polarity at either input. While input
current noise is high relative to bias current, it is not signifi-
cant until source resistance approaches 100 kX.
Input common-mode range is typically from 3V above Vb to
1.5V below Va, so that a large potential drop between the
input signal and output reference can be accommodated.
However, a return path for the input bias current must be
provided; the differential input stage is not isolated from the
supplies. Differential input swing in the linear region is equal
to output swing divided by gain, and typically ranges from
a. Usual configuration swing.
e
e
1.3V at G 10 to 13 mV at G 1000.
TL/H/5609–8
b. Unequal supplies, output ground referred. Full output swing pre-
Clamp diodes are provided to prevent zener breakdown and
resulting degradation of the input transistors. At large input
served referred to supplies.
FIGURE 1. Reference Connections
9
Application Hints (Continued)
TL/H/5609–9
c. Single Supply, Unipolar Output
d. Single Supply, Bipolar Output
FIGURE 1. Reference Connections (Continued)
OUTPUTS
duce an offset shift. A simple low-pass RC filter will usually
cure this problem (Figure 2). Use film type resistors for their
low thermal EMF. In highly noisy environments, LC filters
can be substituted for increased RF attenuation.
The LM363’s output can typically swing within 1V of the
supplies at light loads. While specified to drive a 2 kX load
g
to 10V, current limit is typically 15 mA at room tempera-
ture. The output can stably drive capacitive loads up to
400 pF. For higher load capacitance, the amplifier may be
overcompensated (see COMPENSATION section, follow-
ing). The output may be continuously shorted to ground
without damaging the device.
OFFSET VOLTAGE
The LM363’s offset voltage is internally trimmed to a very
low value. Note that data sheet values are given at
a
b
e
e
e
e
T
25 C, V
§
0V and V
V
15V. For other condi-
j
CM
TL/H/5609–10
tions, warm-up drift, temperature drift, common-mode rejec-
tion and power supply rejection must be taken into account.
Warm-up drift, due to chip and package thermal gradients, is
an effect separate from temperature drift. Typical warm-up
drift is tabulated in the Electrical Characteristics; settling
time is approximately 5 minutes in still air. At load currents
FIGURE 2. Low Pass Filter Prevents RF Rectification
Instrumentation amplifiers have both an input offset voltage
(V ) and an output offset voltage (V
IOS
). The total input-
OOS
) is related to the instrumen-
referred offset voltage (V
OSRTI
e
a
V
OOS
tation amplifier gain (G) as follows: V
V
IOS
/
OSRTI
G. The offset voltage given in the LM363 specifications is
the total input-referred offset. As long as only one gain is
used, offset voltage can be nulled at either input or output
as shown in Figures 3a and 3b. When the 16-pin device is
up to
(DV
5 mA, thermal feedback effects are negligible
e
2mV at G 1000).
s
OS
Care must be taken in measuring the extremely low offset
voltages of the high gain amplifiers. Input leads must be
held isothermal to eliminate thermocouple effects. Oscilla-
tions, due to either heavy capacitive loading or stray capaci-
tance from input to output, can cause erroneous readings.
In either case, overcompensation will help. High frequency
noise fed into the inputs may be rectified internally, and pro-
used at multiple gain settings, both V
IOS
and V should
OOS
be nulled to get minimum offset at all gains, as shown in
Figure 3c. The correct procedure is to trim V for zero
OOS
e
output at G 10, then trim V
e
at G 1000.
IOS
TL/H/5609–11
FIGURE 3. Offset Voltage Trimming
10
Application Hints (Continued)
Because the LM363’s offset voltage is so low to begin with,
offset nulling has a negligible effect on offset temperature
drift. For example, zeroing a 100 mV offset, assuming external
worst-case output offset of 50 mV, creating an input-re-
e
e
ferred error of 5 mV at G 10 or 50 mV at G 1000.
Increasing gain this way increases output offset error. An
LM363H-100 may have an output offset of 5 mV, resulting in
input referred offset component of 50 mV. Raising the gain
to 200 yields a 10 mV error at the output and changes input
referred error by an additional 50 mV.
resistor TC of 200 ppm/ C and worst-case internal resistor
§
TC, results in an additional drift component of 0.08 mV/ C.
§
For this reason, drift specifications are guaranteed, with or
without external offset nulling.
External resistors connected to the reference and sense
pins can only increase the gain. If ultra-low output imped-
ance is not critical, the technique in Figure 5 can be used to
GAIN ADJUSTMENT
Gain may be increased by adding an external voltage divid-
er between output force and sense and reference; the pre-
ferred connection is shown in Figure 4. Since both the
sense and reference pins look like 50 kX ( 20 kX) to V ,
impedances presented to both pins must be equal to avoid
trim the gain to nominal value. Alternatively, the V
OS
adjust-
ment terminals on the 16-pin package may be used to trim
the gain (Figure 10b).
b
g
offset error. For example, a 100X imbalance can create a
R1 and R2 should be as low as possible to avoid errors due to 50 kX
input impedance of reference and sense pins. Total resistance
a
(R2 2R1) should be above 4 kX, however, to prevent excessive load
on the LM363 output. The exact formula for calculating gain (G) is:
2R1 R1
e
a
1
O
a
G
G
G
R2 50k
#
J
e
preset gain
O
The last term may be ignored in applications where gain accuracy is not
critical. The table below gives suggested values for R1 and R2 along
with the calculated error due to ‘‘closest value’’ standard 1% resistors.
Total gain error tolerance includes contributions from LM363 G error
O
g
and resistor tolerance ( 1%) and works out to approximately 2.5% in
every case.
Pinout shown is for 16-pin package. This same technique can also be
used with 8-pin versions.
TL/H/5609–12
Gain Increase
1.5
1.21k
5k
2
2.5
2k
3
4
5
6
7
8
9
10
4.42k
1k
R1
R2
1.21k
2k
1.78k
1.21k
2k
2.49k
1k
2.94k
1k
3.48k
1k
3.92k
1k
2.49k
2.74k
0
2.05k
1k
a
b
b
b
a
a
b
a
b
b
0.7%
Error (typ)
0.6%
0.2%
0.3%
0.6%
0.8%
0.5%
0.9%
0.4%
0.9%
FIGURE 4. Increasing Gain
Pinout shown is for 8-pin versions.
This same technique can also be used
with 16-pin version.
TL/H/5609–13
FIGURE 5. Adjusting Gain, Alternate Technique
11
Application Hints (Continued)
COMPENSATION AND OUTPUT CLAMPING
Heavy Miller overcompensation on the 16-pin package can
degrade AC PSRR. A large capacitor between pins 15 and
16 couples transients on the positive supply to the output
buffer. Since the amplifier bandwidth is severely rolled off it
cannot keep the output at the correct state at moderate
frequencies. Hence, for good PSRR, either keep the Miller
capacitance under 1000 pF or use the pin 15-to-ground
compensation shown in Table I.
The LM363 is internally compensated for unity feedback
from output to sense. Increasing gain with external dividers
will decrease the bandwidth and increase stability margin.
Without external compensation, the amplifier can stably
drive capacitive loads up to 400 pF. When used as an op
amp (sense and reference pins grounded, feedback to in-
verting input), the LM363 is stable for gains of 100 or more.
For greater stability, the device may be over-compensated
as in Figure 6. Tables I and II depict suggested compensa-
tion components along with the resulting changes in large
and small signal bandwidth for the 8-pin and 16-pin pack-
ages, respectively.
Note that the RC network from pin 8 of the 8-pin device to
ground has a large effect on power bandwidth, especially at
low gains. The Miller capacitance utilized for overcompen-
sating the 16-pin device permits higher slew rate and larger
load capacitance for the same bandwidth, and is preferred
when bandwidth must be greatly reduced (e.g., to reduce
output noise).
TL/H/5609–14
FIGURE 6. Overcompensation
TABLE I. Overcompensation on 8-Pin Package
Small Signal
3 dB
Bandwidth
(kHz)
Power
Bandwidth
Maximum
Capacitive
Load
Compensation Network
²
g
(Pin 8 to Ground)
(
10V Swing)
(Hz)
Gain
(pF)
Ð
125
95
45
10
1
100k
15k
1.8k
200
20
400
600
800
1000*
1000*
100 pF, 15k
1000 pF, 5k
0.01 mF,500X
0.1 mF
500
Ð
240
170
80
20
2
100k
15k
1.8k
200
20
400
900
1200
1600*
2000*
100 pF, 15k
1000 pF, 5k
0.01 mF, 500X
0.1 mF
100
10
Ð
240
170
90
20
2
100k
15k
1.8k
200
20
400
900
1200
1600*
2000*
100 pF, 15k
1000 pF, 5k
0.01 mF, 500X
0.1 mF
t
*Also stable for C
0.05 mF
Pin 15 to ground on 16-pin package
L
²
TABLE II. Overcompensation on 16-Pin Package
Small Signal
3 dB
Bandwidth
(Hz)
Power
Bandwidth
Maximum
Compensation
Capacitor
(Pin 15 to 16)
Capacitive
Load
(pF)
Gain
g
(
10V Swing)
(Hz)
Ð
10 pF
100 pF
1000 pF
0.01 mF
45k
16k
2.5k
250
25
45k
16k
2.5k
250
25
1000*
2000*
2500*
3000*
3000*
1000
Ð
10 pF
100 pF
1000 pF
0.01 mF
140k
50k
7.5k
750
75
100k
50k
7.5k
750
75
900
1600
2000*
2000*
2000*
100
Ð
10 pF
100 pF
1000 pF
0.01 mF
180k
60k
9k
900
90
90k
50k
9k
900
90
600
1100
1600
2000*
2000*
10
t
*Also stable for C
0.05 mF
L
12
Application Hints (Continued)
Because the LM363’s output voltage is approximately one
diode drop below the voltage at pin 15 (pin 8 for the 8-pin
device), this point may be used to limit output swing as seen
inFigure 7a. Current available from this pin is only 50 mA, so
that zeners must have a sharp breakdown to clamp accu-
rately. Alternatively, a diode tied to a voltage source could
be used as in Figure 7b.
50 pF to ground at both shield driver outputs. Do not use
only one shield driver for a single-ended signal as oscilla-
tions can result; shield driver to input capacitance must be
g
roughly balanced ( 30%). To further reduce noise pickup,
the shielded signal lines may be enclosed together in a
grounded shield. If a large amount of RF noise is the prob-
lem, the only sure cure is a filter capacitor at both inputs;
otherwise the RFI may be internally rectified, producing an
offset.
DC loading on the shield drivers should be minimized. The
drivers can only source approximately 40 mA; above this
value the input stage bias voltages change, degrading V
OS
and CMRR. While the shield drivers can sink several mA,
may degrade severely at loads above 100 mA (see
V
OS
Shield Driver Loading Error curve in Typical Performance
Characteristics). Because the shield drivers are one diode
drop above the input levels, unbalanced leakage paths from
shield to input can produce an input offset at high source
impedances. Buffering with emitter-followers (Figure 8b) re-
duces this leakage current by reducing the voltage differen-
tial and eliminates any loading on the amplifier.
TL/H/5609–15
FIGURE 7. Output Clamp
SHIELD DRIVERS
When differential signals are sent through long cables, three
problems occur. First, noise, both common-mode and differ-
ential, is picked up. Second, signal bandwidth is reduced by
the RC low-pass filter formed by the source impedance and
the cable capacitance. Finally, when these RC time con-
stants are not identical (unbalanced source impedance
and/or unbalanced capacitance), AC common-mode rejec-
tion is degraded, amplifying both induced noise and
‘‘ground’’ noise. Either filtering at the amplifier inputs or
slowing down the amplifier by overcompensating will indeed
reduce the noise, but the price is slower response. The
LM363D’s dual shield drivers can actually increase band-
width while reducing noise.
TL/H/5609–16
The way this is done is by bootstrapping out shield capaci-
tance. The shield drivers follow the input signal. Since both
sides of the shield capacitance swing the same amount, it is
effectively out of the circuit at frequencies of interest.
Hence, the input signal is not rolled off and AC CMRR is not
degraded (Figure 8). The LM363D’s shield drivers can han-
dle capacitances (shield to center conductor) as high as
1000 pF with source resistances up to 100 kX.
FIGURE 8. Driving Shielded Cables
MISCELLANEOUS TRIMMING
The V
adjust and shield driver pins available on the 16-
OS
pin package may be used to trim the other parameters be-
sides offset voltage, as illustrated inFigure 10. The bias-cur-
rent trim relies on the fact that the voltage on the shield
driver and gain setting pins is one diode drop respectively
above and below the input voltage. Input bias current can
be held to within 100 pA over the entire common-mode
range, and input offset current always stays under 30 pA.
For best results, identical shielded cables should be used
for both signal inputs, although small mismatches in shield
s
driver to ground capacitance ( 500 pF) do not cause prob-
lems. At certain low values of cable capacitance (50 pF–
200 pF), high frequency oscillations can occur at high
The CMRR trims use the shield driver pins to drive the V
OS
adjust pins, thus maintaining the LM363’s ultra-high input
impedance.
t
source resistance ( 10 kX). This is alleviated by adding
13
Application Hints (Continued)
If power supply rejection is critical, frequently only the nega-
tive PSRR need be adjusted, since the positive PSRR is
more tightly specified. Any or all of the trim schemes of
Figure 10 can be combined as desired. As long as the cen-
ter tap of the 100k trimpot is returned to a voltage 200 mV
below Va, the trim schemes shown will not greatly affect
V
. Both the gain and DC CMRR trims can degrade posi-
OS
tive PSRR; the positive PSRR can then be nulled out if de-
sired. The correct order of trimming from first to last is bias
current, gain, CMRR, negative PSRR, positive PSRR and
V
OS
.
Top Trace: Cable Shield Grounded
TL/H/5609–17
TL/H/5609–18
FIGURE 9. Improved Response using Shield Drivers
TL/H/5609–19
FIGURE 10. Other Trims for 16-Pin Package
14
Typical Applications
4 mA-20 mA Two Wire Current Transmitter
TL/H/5609–20
The LM329 reference provides excellent line regulation and gain stability. When bridge is balanced
e
(I
4 mA), there’s no drop across R3 and R4, so that gain and offset adjustments are non-in-
OUT
teractive. The LM334 configured as a zero-TC current source supplies quiescent current to circuit.
R11 provides current limiting.
Design Equations
R2
e
a
I
R7
a
e
4 mA
I
(I
)
1
OS
R6
DI
R1
#
J
a a
R2 R3 R4 10 mA
A
OUT
V
j
j
e
Gain
X
a
R3 R4
DV
R1
mV
IN
e
when A
LM363 voltage gain
0.68V 68 mV
V
j
e
a
Pick I
3.8 mA
334
R9
R10
b
V
2.4V
Z
e
a
e
26 mA
I
I
I
MAX
334
R11
j
j
I
-I -I
334 363
1.5mA
BRIDGE(MAX)
Z
Precision Current Source (Low Output Current)
e
R1
I
R2
V
IN
s
10V
e
,
V
IN
OUT
l
l
GR1
TL/H/5609–21
Precision Voltage to Current Converter (Low Input Voltage)
e
R1
R2
e
Req
R1 50 kX
ll
G V
G V
IN
IN
e
e
I
OUT
Req
1 kX
TL/H/5609–22
15
Typical Applications (Continued)
Curvature Corrected Platinum RTD Thermometer
*70k and 2k should track to 5 ppm/ C
§
**Less than 5 ppm/ C drift
§
Less than 100 ppm/ C drift
²
§
These resistors should track to 20 ppm/ C
²²
§
Equivalent circuit, showing lead resistance
³
b
150 C. A unique trim arrangement eliminates cumbersome trim inter-
This thermometer is capable of 0.01 C accuracy over
a
50 C to
§
§
§
actions so that zero, gain, and nonlinearity correction can be trimmed in
one oven trip. Extra op amps provide full Kelvin sensing on the sensor
without adding drift and offset terms found in other designs. A2 is con-
figured as a Howland current pump, biasing the sensor with a fixed
current.
Resistors R2, R3, R4 and R5 from a bridge driven into balance by A1. In
e
TL/H/5609–23
balance, both inputs of A1 are at the same voltage. Since R6 R7, A1
draws equal currents from both legs of the bridge. Any loading of the
R4/R5 leg by the sensor would unbalance the bridge; therefore, both
bridge taps are given to the sensor open circuit voltage and no current
is drawn.
Precision Temperature Controller
TL/H/5609–24
*Ultronix 105A wirewound
e
Thermistor Yellow Springs 44032
Setpoint stability 2.5X10
Ý
b
4
e
C/Hr
§
16
Typical Applications (Continued)
Low Frequency Rolloff (AC Coupling)
1
e
e
1 Hz
f1
2qC1(50 kX)
e
e
100Hz
f2 100 f1
Reduced DC voltage gain
attenuates offset error and
1/f noise by a factor of 100.
TL/H/5609–25
Precision Comparator with Balanced Inputs and Variable Offset
Boosted Current Source with Limiting
e
R1 R2
G V
IN
e
I
O
R2
V
BE
e
I
MAX
R2
j
j
60 mA
t
15 mS at 1 mV overdrive
pd
e
a
DV
V2 0.6V
DV
OUT
OUT
e
e
2 mV
Hysteresis
Offset
a
G(R1 R2)
/G
e
V
SENSE
g
1.3V range
TL/H/5609–26
Thermocouple Amplifier with Cold Junction Compensation
Input protection circuitry allows
thermocouple to short to 120 V without
AC
damaging amplifier.
Calibration:
1) Apply 50 mV signal in place of thermocouple.
e
Trim R3 for V
OUT
12.25V.
2) Reconnect thermocouple. Trim R9 for correct
output.
TL/H/5609–27
17
Typical Applications (Continued)
Synchronous Demodulator
TL/H/5609–28
*Use square wave drive produced by optical chopper to run LF13333 switch inputs.
Pulsed Bridge Driver/Amplifier
TL/H/5609–29
18
Typical Applications (Continued)
Precision Barometer
e
**Parallel trim for 28.00 Hg 0V
×
e
Parallel trim for 32.00 Hg 4V out
²
×
Ý
*B.L.H. Electronics DHF-444114
Pressure Transducer,
350X input impedance.
e
Output 1 mV/volt excitation/psi
TL/H/5609–30
Removing Large DC Offsets
*Optional bandlimiting to reduce noise.
e
e
Pick R1C1 R2C2 R3C3/10
1
TL/H/5609–31
e
2qf
l
e
LM363 bias currents flowing into R1 and R2.
f
0.1 Hz for values shown. Integrator nulls out offset error to
l
Removing Small DC Offsets
*Optional bandlimiting to reduce noise.
Low frequency break
1
e
e
0.01 Hz
frequency f
l
2qR1C1
Accommodates out referred offset of several volts. Limit is set by max
differential between reference and sense terminals.
TL/H/5609–32
19
20
Physical Dimensions inches (millimeters)
Metal Can Package (H)
Order Number LM363H-10, LM363H-100 or LM363H-500
NS Package Number H08C
21
Physical Dimensions inches (millimeters) (Continued)
Hermetic Dual-In-Line Package (D)
Order Number LM363D
NS Package Number D16C
LIFE SUPPORT POLICY
NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT
DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF NATIONAL
SEMICONDUCTOR CORPORATION. As used herein:
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with instructions for use provided in the labeling, can
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to the user.
2. A critical component is any component of a life
support device or system whose failure to perform can
be reasonably expected to cause the failure of the life
support device or system, or to affect its safety or
effectiveness.
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