RT2858B_技术文档

®

RT2858B

3A, 18V, 650kHz ACOT^TMSynchronous Step-Down Converter

General Description

Features

^zACOT^TMControl for Fast Transient, f_SWStability, and

Robust Loop Stability with all-MLCC C_OUT

^z4.5V to 18V Input Voltage Range

The RT2858B is a synchronous DC/DC step-down

converter with Advanced Constant On-Time (ACOT^TM

)

mode control. It achieves high power density to deliver up

to 3Aoutput current from a 4.5V to 18V input supply. The

proprietary ACOT^TMmode offers an optimal transient

response over a wide range of loads and all kinds of ceramic

capacitors, which allows the device to adopt very low ESR

output capacitors for ensuring performance stabilization.

In addition, the RT2858B keeps an excellent constant

switching frequency under line and load variation and the

integrated synchronous power switches with theACOT^TM

mode operation provides high efficiency in whole output

current load range. Cycle-by-cycle current limit provides

an accurate protection by a valley detection of low-side

MOSFET and external soft-start setting eliminates input

current surge during startup. Protection functions also

include output under voltage protection and thermal

shutdown.

^z3A Output Current

^zR_DSON120mΩ/50mΩ for High Efficiency Across I_OUT

Range and Competitive Advantage for I_OUT> 1.5A

^zAdvanced Constant On-Time Control

^zSupport All Ceramic Capacitors

^zUp to 95% Efficiency

^z650kHz f_SW; Start-Up into Pre-Biased Load;

Adjustable Soft-Start; Internal Bootstrap

^zAdjustable Output Voltage from 0.765V to 8V

z Enable; UVLO; OCP (Cycle-by-Cycle); and OTP

(150°C)

z RoHS Compliant and Halogen Free

Applications

z Industrial and Commercial Low Power Systems

z Computer Peripherals

z LCDMonitors and TVs

z Green Electronics/Appliances

z Point of Load Regulation for High-Performance DSPs,

FPGAs, and ASICs

Simplified Application Circuit

Load Transient Response

V_OUT

(50mV/Div)

RT2858B

L1

VIN

SW

V

OUT

V

IN

C7

C1

C2

C5

C6

C3 R1

R2

BOOT

Enable

EN

SS

FB

PVCC

V

PVCC

C4

GND

I_OUT

(1A/Div)

V_IN= 12V, V_OUT= 1.05V, I_OUT= 0A to 3A

Time (100μs/Div)

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RT2858B

Ordering Information

Marking Information

RT2858B

RT2858BHGSP

Package Type

RT2858BHGSP : Product Number

SP : SOP-8 (Exposed Pad-Option 2)

RT2858BH

GSPYMDNN

YMDNN : Date Code

Lead Plating System

G : Green (Halogen Free and Pb Free)

H : Hiccup Mode OVP and UVP

N : OVP and UVP disable

RT2858BNGSP

RT2858BNGSP : Product Number

YMDNN : Date Code

Note :

RT2858BN

GSPYMDNN

Richtek products are :

` RoHS compliant and compatible with the current require-

ments of IPC/JEDEC J-STD-020.

` Suitable for use in SnPb or Pb-free soldering processes.

Pin Configurations

(TOP VIEW)

8

7

6

5

EN

FB

VIN

2

3

4

BOOT

SW

GND

PVCC

SS

9

GND

SOP-8 (Exposed Pad)

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RT2858B

Absolute Maximum Ratings (Note 1)

z Supply Voltage, VIN ----------------------------------------------------------------------------------------------- −0.3V to 21V

z Switch Voltage, SW ----------------------------------------------------------------------------------------------- −0.8V to (V_IN+ 0.3V)

<10ns ----------------------------------------------------------------------------------------------------------------- −5V to 25V

z BOOT to SW -------------------------------------------------------------------------------------------------------- −0.3V to 6V

z PVCC to VIN--------------------------------------------------------------------------------------------------------- −18V to 0.3V

z Other Pins------------------------------------------------------------------------------------------------------------ −0.3V to 21V

z Power Dissipation, P_D@ T_A= 25°C

SOP-8 (Exposed Pad) -------------------------------------------------------------------------------------------- 2.041W

z Package Thermal Resistance (Note 2)

SOP-8 (Exposed Pad), θ_JA--------------------------------------------------------------------------------------- 49°C/W

SOP-8 (Exposed Pad), θ_JC-------------------------------------------------------------------------------------- 8°C/W

z Junction Temperature Range------------------------------------------------------------------------------------- 150°C

z Lead Temperature (Soldering, 10 sec.)------------------------------------------------------------------------ 260°C

z Storage Temperature Range ------------------------------------------------------------------------------------- −65°C to 150°C

z ESD Susceptibility (Note 3)

HBM (Human Body Model)--------------------------------------------------------------------------------------- 2kV

Recommended Operating Conditions (Note 4)

z Supply Voltage, VIN ----------------------------------------------------------------------------------------------- 4.5V to 18V

z Junction Temperature Range------------------------------------------------------------------------------------- −40°C to 125°C

z Ambient Temperature Range------------------------------------------------------------------------------------- −40°C to 85°C

Electrical Characteristics

(V_IN= 12V, T_A= −40°C to 85°C, unless otherwise specified)

Parameter

Supply Current

Symbol

Test Conditions

Min

Typ

Max

Unit

Shutdown Current

Quiescent Current

Logic Threshold

I_SHDN

I_Q

EN = 0V, T_A= 25°C

--

1

10

μA

EN = 5V, V_FB= 0.8V, T_A= 25°C

1.3

mA

Logic-High

Logic-Low

2

--

18

EN Input Voltage

V

--

0.4

V_FBVoltage and Discharge Resistance

Feedback Threshold Voltage V_FB

T_A= 25°C

0.757 0.765 0.773

V

Feedback Input Current

I_FB

V_FB= 0.8V, T_A= 25°C

−0.1

0

0.1

μA

V_PVCCOutput

6V ≤ V_IN≤ 18V, 0 < I_PVCC< 5mA, T_A

= 25°C

V_PVCCOutput Voltage

V_PVCC

4.8

5.1

5.4

V

Line Regulation

Load Regulation

Output Current

6V ≤ V_IN≤ 18V, I_PVCC= 5mA

0 < I_PVCC< 5mA

--

20

100

--

mV

mA

I_PVCC

V_IN= 6V, V_PVCC= 4V, T_A= 25°C

70

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RT2858B

Parameter

R_DS(ON)

Symbol

Test Conditions

Min

Typ

Max

Unit

mΩ

A

R_{DS(ON)_H}

R_{DS(ON)_L}

High-Side

Low-Side

V_BOOT– SW = 5V, T_A= 25°C

T_A= 25°C

--

120

50

--

Switch-On

Resistance

Current Limit

I_LIM

4

5

6

Thermal Shutdown

Thermal Shutdown Threshold T_SD

Thermal Shutdown Hysteresis ΔT_SD

On-Time Timer Control

--

150

20

--

°C

On-Time

t_ON

V_IN= 12V, V_OUT= 1.05V

--

135

260

--

ns

Minimum Off-Time

Soft-Start

t_OFF(MIN)

V_FB= 0.7V, T_A= 25°C

310

SS Charge Current

SS Discharge Current

UVLO

V_SS= 0V

1.4

0.1

2

2.6

--

μA

V_SS= 0.5V

0.2

mA

UVLO Threshold

Hysteresis

VIN Rising to Wake up V_PVCC

3.6

3.85

350

4.1

V

130

400

mV

Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are

stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in

the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may

affect device reliability.

Note 2. θ_JAis measured at T_A= 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. θ_JCis

measured at the exposed pad of the package. The PCB copper area with exposed pad is 70mm²(please see PCB

Layout section for recommended shape & board physical design guidance).

Note 3. Devices are ESD sensitive. Handling precaution is recommended.

Note 4. The device is not guaranteed to function outside its operating conditions.

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RT2858B

Typical Operating Characteristics

Efficiency vs. Load Current

Output Voltage vs. Input Voltage

100

1.10

1.09

1.08

1.07

1.06

1.05

1.04

1.03

1.02

1.01

1.00

V_IN= 4.5V

90

80

70

60

V_IN= 12V

50

40

V_IN= 18V

30

20

10

V_IN= 4.5V to 18V, V_OUT= 1.05V, I_OUT= 0A

V_OUT= 1.05V

0

0.001

0.01

0.1

1

10

4

6

8

10

12

14

16

18

Load Current (A)

Input Voltage (V)

Output Voltage vs. Temperature

Output Voltage vs. Output Current

5.20

1.065

1.060

1.055

1.050

1.045

1.040

1.035

1.030

1.025

5.15

5.10

5.05

5.00

4.95

4.90

4.85

4.80

V_IN= 18V

V_IN= 12V

V_OUT= 1.05V

V_IN= 12V, V_OUT= 5V, I_OUT= 0A

-50

-25

0

25

50

75

100

125

0

0.5

1

1.5

2

2.5

3

Temperature (°C)

Output Current (A)

Frequency vs. Input Voltage

Reference Voltage vs. Temperature

700

680

660

640

620

600

0.80

0.78

0.76

0.74

0.72

0.70

V_OUT= 1.05V, I_OUT= 0.3A

12 14 16 18

V_IN= 12V, V_OUT= 0.765V

4

6

8

10

-50

-25

0

25

50

75

100

125

Input Voltage (V)

Temperature (°C)

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RT2858B

Current Limit vs. Temperature

Current Limit vs. Input Voltage

6.0

5.5

5.0

4.5

4.0

3.5

7.0

6.5

6.0

5.5

5.0

4.5

4.0

3.5

3.0

Peak Current

Valley Current

V_OUT= 1.05V

14 16 18

V_IN= 12V, V_OUT= 1.05V

50 75 100 125

-50

-25

0

25

4

6

8

10

12

Temperature (°C)

Input Voltage (V)

Load Transient Response

V_OUTRipple

V_OUT

(10mV/Div)

V_OUT

(50mV/Div)

V_SW

(5V/Div)

I_OUT

(1A/Div)

V_IN= 12V, V_OUT= 1.05V, I_OUT= 3A

Time (500ns/Div)

V_IN= 12V, V_OUT= 1.05V, I_OUT= 0A to 3A

Time (100μs/Div)

Power On from VIN

Power Off from VIN

V_IN

(5V/Div)

V_OUT

(1V/Div)

I_OUT

(2A/Div)

I_SW

(2A/Div)

V_IN= 12V, V_OUT= 1.05V, I_OUT= 3A

Time (1ms/Div)

V_IN= 12V, V_OUT= 1.05V, I_OUT= 3A

Time (5ms/Div)

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RT2858B

Power On from EN

Power Off from EN

EN

(2V/Div)

V_OUT

(1V/Div)

I_OUT

(2A/Div)

I_SW

(2A/Div)

V_IN= 12V, V_OUT= 1.05V, I_OUT= 3A

Time (5ms/Div)

V_IN= 12V, V_OUT= 1.05V, I_OUT= 3A

Time (1ms/Div)

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RT2858B

Functional Pin Description

Pin No.

Pin Name

Pin Function

Enable Control Input. A logic-high enables the converter; a logic-low forces

the IC into shutdown mode reducing the supply current to less than 10μA.

1

EN

Feedback Voltage Input. It is used to regulate the output of the converter to a

set value via an external resistive voltage divider. The feedback threshold

voltage is 0.765V typically.

2

3

4

FB

Regulator Output for Internal Circuit. Connect a 1μF capacitor to GND to

stabilize output voltage.

PVCC

SS

Soft-Start Time Setting. SS controls the soft-start period. Connect a capacitor

from SS to GND to set the soft-start period. A 3.9nF capacitor sets the

soft-start period of V_OUTto 2.6ms.

5, 9

(Exposed Pad)

Ground. The Exposed pad should be soldered to a large PCB and connected

to GND for maximum thermal dissipation.

GND

SW

6

7

Switch Node. Connect this pin to an external L-C filter.

Bootstrap Supply for High-Side Gate Driver. Connect a 0.1μF or greater

ceramic capacitor between the BOOT to SW pins.

BOOT

Power Input. The input voltage range is from 4.5V to 18V. Must bypass with a

suitably large ( ≥10μF x 2) ceramic capacitor.

8

VIN

Function Block Diagram

BOOT

PVCC

Internal

Regulator

VIN

PVCC

Over Current

Protection

PVCC

VIBIAS

V

REF

UGATE

LGATE

Switch

Controller

SW

Driver

SW

PVCC

2µA

Ripple

Gen.

GND

+

SS

FB

Comparator

On-Time

EN

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RT2858B

Operation

In normal operation, the high-side N-MOSFET is turned

on when the FB Comparator sets the Switch Controller,

and it is turned off when On-Time Controller resets the

Switch Controller. While the high-sideN-MOSFET is turned

off, the low-sideN-MOSFET is turned on and waits for the

FB Comparator to set the beginning of next cycle.

Internal Regulator

Provide internal power for logic control and switch gate

drivers.

On-Time Controller

Control on-time according to VIN and SW to obtain

constant switching frequency.

The FB Comparator sets the Switch Controller by

comparing the feedback signal (FB) from output voltage

with the internal 0.765V reference. When load transient

induces VOUT drop, the FB voltage will be less than its

threshold voltage. This means that the high-side N-

MOSFET will turn on again immediately after minimum

off-time expired. The switching frequency will vary during

the transient period thus can provide a very fast transient

response. After the load transient finished, the RT2858B

will be back to steady state with a constant switching

frequency.

OVP/UVP Protection

The RT2858B detects over and under voltage conditions

by monitoring the feedback voltage on FB pin. The two

functions are enabled after approximately 1.7 times the

soft-start time. When the feedback voltage becomes

higher than 120% of the target voltage, the OVP

comparator will go high to turn off both internal high side

and low side MOSFETs. When the feedback voltage is

lower than 70% of the target voltage for 250μs, the UVP

comparator will go high to turn off both internal high side

and low side MOSFETs.

Enable

Activate internal regulator once EN input level is higher

than the target level. Force IC to enter shutdown mode

when the EN input level is lower than 0.4V

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RT2858B

Typical Application Circuit

L1

1.4µH

RT2858B

VIN

V

6

OUT

8

1

SW

V

IN

1.05V/3A

C1

C2

C7

22µF x 2

C6

0.1µF

10µF x 2

0.1µF

C3

R1

7

2

8.25k

BOOT

FB

Enable

EN

5, 9 (Exposed Pad)

4

GND

SS

R2

22.1k

3

PVCC

V

PVCC

C5

3.9nF

C4

1µF

Table 1. Suggested Component Values (V_IN= 12V)

V

(V)

R1 (kΩ)

6.81

R2 (kΩ)

22.1

C3 (pF)

L1 (μH)

C7 (μF)

OUT

1

1.05

1.2

1.8

2.5

3.3

5

--

1.4

2

22 to 68

8.25

22.1

12.7

22.1

30.1

49.9

73.2

124

180

22.1

5 to 22

2

3.3

7

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RT2858B

Design Procedure

Inductor Selection

For best efficiency, choose an inductor with a low DC

resistance that meets the cost and size requirements.

For low inductor core losses some type of ferrite core is

usually best and a shielded core type, although possibly

larger or more expensive, will probably give fewer EMI

and other noise problems.

Selecting an inductor involves specifying its inductance

and also its required peak current. The exact inductor value

is generally flexible and is ultimately chosen to obtain the

best mix of cost, physical size, and circuit efficiency.

Lower inductor values benefit from reduced size and cost

and they can improve the circuit's transient response, but

they increase the inductor ripple current and output voltage

ripple and reduce the efficiency due to the resulting higher

peak currents. Conversely, higher inductor values increase

efficiency, but the inductor will either be physically larger

or have higher resistance since more turns of wire are

required and transient response will be slower since more

time is required to change current (up or down) in the

inductor. A good compromise between size, efficiency,

and transient response is to use a ripple current (ΔI_L) about

20-50% of the desired full output load current. Calculate

the approximate inductor value by selecting the input and

output voltages, the switching frequency (f_SW), the

maximum output current (I_OUT(MAX)) and estimating a ΔI_L

as some percentage of that current.

Considering the Typical Operating Circuit for 1.05V output

at 3Aand an input voltage of 12V, using an inductor ripple

of 1A (33%), the calculated inductance value is :

1.05V× 12V −1.05V

12V×650kHz×1A

(

)

= 1.47μH

L =

The ripple current was selected at 1A and, as long as we

use the calculated 1.47μH inductance, that should be the

actual ripple current amount. Typically the exact calculated

inductance is not readily available and a nearby value is

chosen. In this case 1.4μH was available and actually used

in the typical circuit. To illustrate the next calculation,

assume that for some reason is was necessary to select

a 1.8μH inductor (for example). We would then calculate

the ripple current and required peak current as below :

1.05V× 12V −1.05V

(

)

ΔI_L=

= 0.82A

V

× V − V

IN OUT

(

)

OUT

12V×650kHz×1.8μH

L =

V ×f

×ΔI

L

IN SW

0.82

2

and I_L(PEAK)= 3A +

= 3.41A

Once an inductor value is chosen, the ripple current (ΔI_L)

is calculated to determine the required peak inductor

current.

For the 1.8μH value, the inductor's saturation and thermal

rating should exceed 3.41A. Since the actual value used

was 1.4μH and the ripple current exactly 1A, the required

peak current is 3.53A.

V

OUT

× V − V

IN OUT

(

)

and I

ΔI

2

L

ΔI =

L

= I

+

L(PEAK)

OUT(MAX)

V ×f

×L

IN SW

To guarantee the required output current, the inductor

needs a saturation current rating and a thermal rating that

exceeds I_L(PEAK). These are minimum requirements. To

maintain control of inductor current in overload and short-

circuit conditions, some applications may desire current

ratings up to the current limit value. However, the IC's

output under-voltage shutdown feature make this

unnecessary for most applications.

Input Capacitor Selection

The input filter capacitors are needed to smooth out the

switched current drawn from the input power source and

to reduce voltage ripple on the input. The actual

capacitance value is less important than the RMS current

rating (and voltage rating, of course). The RMS input ripple

current (I_RMS) is a function of the input voltage, output

voltage, and load current :

I_L(PEAK)should not exceed the minimum value of IC's upper

current limit level or the IC may not be able to meet the

desired output current. If needed, reduce the inductor ripple

current (ΔI_L) to increase the average inductor current (and

the output current) while ensuring that I_L(PEAK)does not

exceed the upper current limit level.

V

OUT

× V

− V

(

)

VIN OUT

I

= I

×

RMS

OUT

V

VIN

Ceramic capacitors are most often used because of their

low cost, small size, high RMS current ratings, and robust

surge current capabilities. However, take care when these

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RT2858B

capacitors are used at the input of circuits supplied by a

wall adapter or other supply connected through long, thin

wires. Current surges through the inductive wires can

induce ringing at the RT2858B's input which could

potentially cause large, damaging voltage spikes VIN. If

this phenomenon is observed, some bulk input capacitance

may be required. Ceramic capacitors (to meet the RMS

current requirement) can be placed in parallel with other

types such as tantalum, electrolytic, or polymer (to reduce

ringing and overshoot).

1A

V

=

= 4.4mV

RIPPLE(C)

8×44μF×0.65MHz

V_RIPPLE= 5mV + 4.4mV = 9.4mV

Output Transient Undershoot and Overshoot

In addition to voltage ripple at the switching frequency,

the output capacitor and its ESR also affect the voltage

sag (undershoot) and soar (overshoot) when the load steps

up and down abruptly. The ACOT transient response is

very quick and output transients are usually small.

However, the combination of small ceramic output

capacitors (with little capacitance), low output voltages

(with little stored charge in the output capacitors), and

low duty cycle applications (which require high inductance

to get reasonable ripple currents with high input voltages)

increases the size of voltage variations in response to

very quick load changes. Typically, load changes occur

slowly with respect to the IC's 650kHz switching frequency.

But some modern digital loads can exhibit nearly

instantaneous load changes and the following section

shows how to calculate the worst-case voltage swings in

response to very fast load steps.

Choose capacitors rated at higher temperatures than

required. Several ceramic capacitors may be paralleled to

meet the RMS current, size, and height requirements of

the application. The typical operating circuit uses two 10μF

and one 0.1μF low ESR ceramic capacitors on the input.

Output Capacitor Selection

The RT2858B are optimized for ceramic output capacitors

and best performance will be obtained using them. The

total output capacitance value is usually determined by

the desired output voltage ripple level and transient response

requirements for sag (undershoot on positive load steps)

and soar (overshoot on negative load steps).

The output voltage transient undershoot and overshoot each

have two components : the voltage steps caused by the

output capacitor's ESR, and the voltage sag and soar due

to the finite output capacitance and the inductor current

slew rate. Use the following formulas to check if the ESR

is low enough (typically not a problem with ceramic

capacitors) and the output capacitance is large enough to

prevent excessive sag and soar on very fast load step

edges, with the chosen inductor value.

Output Ripple

Output ripple at the switching frequency is caused by the

inductor current ripple and its effect on the output

capacitor's ESR and stored charge. These two ripple

components are called ESR ripple and capacitive ripple.

Since ceramic capacitors have extremely low ESR and

relatively little capacitance, both components are similar

in amplitude and both should be considered if ripple is

critical.

The amplitude of the ESR step up or down is a function of

the load step and the ESR of the output capacitor:

V_{ESR_STEP}= ΔI_OUT×R_ESR

V_RIPPLE= V_RIPPLE(ESR)+ V_RIPPLE(C)

V_RIPPLE(ESR)= ΔI_L×R_ESR

The amplitude of the capacitive sag is a function of the

load step, the output capacitor value, the inductor value,

the input-to-output voltage differential, and the maximum

duty cycle. The maximum duty cycle during a fast transient

is a function of the on-time and the minimum off-time since

the ACOT^TMcontrol scheme will ramp the current using

on-times spaced apart with minimum off-times, which is

ΔI

OUT

L

V

=

RIPPLE(C)

8×C

× f

SW

For the Typical Operating Circuit for 1.05V output and an

inductor ripple of 1A, with 2 x 22μF output capacitance

each with about 10mΩ ESR including PCB trace

resistance, the output voltage ripple components are :

V_RIPPLE(ESR)= 1A×5mΩ = 5mV

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RT2858B

as fast as allowed. Calculate the approximate on-time

(neglecting parasitics) and maximum duty cycle for a given

input and output voltage as :

and an even greater percentage of the output voltage. This

is illustrated by comparing the previous to the next

example.

V_OUT

V ×f_SW

IN

t_ON

The Typical Operating Circuit for 12V to 3.3V with a 2μH

inductor and 2 x 22μF output capacitance can be used to

illustrate the effect of a higher output voltage. The output

voltage sag and soar in response to full 0A-3A-0A

instantaneous transients are calculated as follows :

3.3V

t_ON

=

and D_MAX=

t_ON+ t_OFF(MIN)

The actual on-time will be slightly longer as the IC

compensates for voltage drops in the circuit, but we can

neglect both of these since the on-time increase

compensates for the voltage losses. Calculate the output

t_ON

=

= 423ns

voltage sag as :

12V×650kHz

2

L×(ΔI

)

OUT

423ns

V

SAG

=

and D_MAX

=

= 0.62

2×C

× V

×D

− V

MAX OUT

(

)

OUT

IN(MIN)

423ns+ 260ns

2μH×(3A)²

The amplitude of the capacitive soar is a function of the

V

SAG

=

= 49.5mV

2×44μF× 12V×0.62−3.3V

(

)

load step, the output capacitor value, the inductor value

2μH×(3A)²

and the output voltage :

V

SOAR

=

= 62mV

2×44μF×3.3V

2

L×(ΔI

)

OUT

V

SOAR

=

In this case the sag is about 1.5% of the output voltage

and the soar is only 2% of the output voltage.

2×C

× V

OUT

For the Typical Operating Circuit for 1.05V output, the

circuit has an inductor 1.4μH and 2 x 22μF output

capacitance with 5mΩ ESR each. The ESR step is 3A x

2.5mΩ = 7.5mV which is small, as expected. The output

voltage sag and soar in response to full 0A-3A-0A

instantaneous transients are :

Any sag is always short-lived, since the circuit quickly

sources current to regain regulation in only a few switching

cycles. With the RT2858B, any overshoot transient is

typically also short-lived since the converter will sink

current, reversing the inductor current sharply until the

output reaches regulation again.

1.05V

12V×650kHz

t_ON

=

= 135ns

Most applications never experience instantaneous full load

steps and the RT2858B's high switching frequency and

fast transient response can easily control voltage regulation

at all times. Also, since the sag and soar both are

proportional to the square of the load change, if load steps

were reduced to 1A(from the 3Aexamples preceding) the

voltage changes would be reduced by a factor of almost

ten. For these reasons sag and soar are seldom an issue

except in very low-voltage CPU core or DDR memory

supply applications, particularly for devices with high clock

frequencies and quick changes into and out of sleep

modes. In such applications, simply increasing the amount

of ceramic output capacitor (sag and soar are directly

proportional to capacitance) or adding extra bulk

capacitance can easily eliminate any excessive voltage

transients.

135ns

and D_MAX

=

= 0.34

135ns+ 260ns

1.4μH×(3A)²

V

SAG

=

= 47mV

2×44μF× 12V×0.34−1.05V

(

)

1.4μH×(3A)²

V

SOAR

=

= 136mV

2×44μF×1.05V

The sag is about 4% of the output voltage and the soar is

a full 13% of the output voltage. The ESR step is negligible

here but it does partially add to the soar, so keep that in

mind whenever using higher-ESR output capacitors.

The soar is typically much worse than the sag in high-

input, low-output step-down converters because the high

input voltage demands a large inductor value which stores

lots of energy that is all transferred into the output if the

load stops drawing current. Also, for a given inductor, the

soar for a low output voltage is a greater voltage change

©

is a registered trademark of Richtek Technology Corporation.

DS2858B-00 September 2013

www.richtek.com

13

RT2858B

Output Capacitors Stability Criteria

Any ESR in the output capacitor lowers the required

minimum output capacitance, sometimes considerably.

For the rare application where that is needed and useful,

the equation including ESR is given here :

The RT2858B's ACOT^TMcontrol architecture uses an

internal virtual inductor current ramp and other

compensation that ensures stability with any reasonable

output capacitor. The internal ramp allows the IC to operate

with very low ESR capacitors and the IC is stable with

very small capacitances. Therefore, output capacitor

selection is nearly always a matter of meeting output

voltage ripple and transient response requirements, as

discussed in the previous sections. For the sake of the

unusual application where ripple voltage is unimportant

and there are few transients (perhaps battery charging or

LED lighting) the stability criteria are discussed below.

V

ESR

OUT

+13647×L× V

C

OUT

≥

2×f

× V ×(R

)

SW

IN

OUT

As can be seen, setting R_ESRto zero and simplifying the

equation yields the previous simpler equation. To allow

for the capacitor's temperature and bias voltage coefficients,

use at least double the calculated capacitance and use a

good quality dielectric such as X5R or X7R with an

adequate voltage rating since ceramic capacitors exhibit

considerable capacitance reduction as their bias voltage

increases.

The equations giving the minimum required capacitance

for stable operation include a term that depends on the

output capacitor's ESR. The higher the ESR, the lower

the capacitance can be and still ensure stability. The

equations can be greatly simplified if the ESR term is

removed by setting ESR to zero. The resulting equation

gives the worst-case minimum required capacitance and

it is usually sufficiently small that there is usually no need

for the more exact equation.

The required output capacitance (C_OUT) is a function of

the inductor value (L) and the input voltage (V_IN) :

5.64×10⁻¹¹

C

OUT

≥

V ×L

IN

The worst-case high capacitance requirement is for low

VIN and small inductance, so a 5V to 3.3V converter is

used for an example. Using the inductance equation in a

previous section to determine the required inductance :

3.3V× 5V −3.3V

(

)

= 1.73μH

L =

5V×650kHz×1A

Therefore, the required minimum capacitance for the 5V

to 3.3V converter is :

5.64×10⁻¹¹

5V×1.73μH

C_OUT

≥

= 6.5μF

Using the 12V to 1.05V typical application as another

example :

5.64×10⁻¹¹

12V×1.4μH

C_OUT

≥

= 3.4μF

©

is a registered trademark of Richtek Technology Corporation.

www.richtek.com

14

DS2858B-00 September 2013

RT2858B

Applications Information

Soft-Start (SS)

Current-Sinking Applications

The RT2858B soft-start uses an external capacitor at SS

to adjust the soft-start timing according to the following

equation :

The RT2858B's is not recommended for current sinking

applications even though its continuous switching

operation allows the IC to sink some current. Sinking

enables a fast recovery from output voltage overshoot

caused by load transients and is normally useful for

applications requiring negative currents, such asDDR V_TT

bus termination applications and changing-output voltage

applications where the output voltage needs to slew

quickly from one voltage to another. However, the IC's

negative current limit is set low (about 1.6A) and the current

limit behavior latches the synchronous rectifier off until

the high-side switch's next pulse, to prevent the possibility

of IC damage from large negative currents. Therefore,

sinking current is not necessarily available at all times.

C_SS(nF)×1.065V

t_SS(ms) =

I_SS(μA)

The available capacitance range is from 2.7nF to 220nF. If

a 3.9nF capacitor is used, the typical soft-start will be

2ms. Do not leave SS unconnected.

Enable Operation (EN)

For automatic start-up the high-voltage EN pin can be

connected to V_IN, either directly or through a 100kΩ

resistor. Its large hysteresis band makes EN useful for

simple delay and timing circuits. EN can be externally

pulled to V_INby adding a resistor-capacitor delay (R_EN

and C_ENin Figure 1). Calculate the delay time using EN's

internal threshold where switching operation begins (1.2V,

typical).

If implementing applications where current-sinking may

occur, take care to allow for the current that is delivered

to the input supply. A step-down converter in sinking

operation functions like a backwards step-up converter.

The current that is sunk at its output terminals is delivered

up to its input terminals. If this current has no outlet, the

input voltage will rise.

An external MOSFET can be added to implement digital

control of EN when no system voltage above 2V is available

(Figure 2). In this case, a 100kΩ pull-up resistor, REN, is

connected between VINand the ENpin. MOSFET Q1 will

be under logic control to pull down the ENpin. To prevent

enabling circuit when VINis smaller than the VOUT target

value or some other desired voltage level, a resistive voltage

divider can be placed between the input voltage and ground

and connected to EN to create an additional input under-

voltage lockout threshold (Figure 3).

Agood arrangement for long-term sinking applications is

for a sinking supply (supply A) that is sinking current

sourced from supply B, to both be powered by the same

input supply. That way, any current delivered back to the

input by supplyAis current that just left the input through

supply B. In this way, the current simply makes a round

trip and the input supply will not rise.

EN

R

EN

In cases where this is not possible, make sure that there

are sufficient other loads on the input supply to prevent

that supply's voltage from rising high enough to cause

damage to itself or any of its loads. In cases where the

sinking is not long-term, such as output-voltage slewing

applications, make sure there is sufficient input capacitance

to control any input voltage rise. The worst-case voltage

rise is :

V

EN

RT2858B

IN

C

EN

GND

Figure 1. External Timing Control

R

100k

EN

V

EN

RT2858B

GND

IN

C

OUT

× ΔV

OUT

ΔV

=

IN

Q1

Enable

C

IN

Figure 2. Digital Enable Control Circuit

©

is a registered trademark of Richtek Technology Corporation.

DS2858B-00 September 2013

www.richtek.com

15

RT2858B

External BOOT Bootstrap Diode

R

EN1

V

IN

EN

When the input voltage is lower than 5.5V it is

recommended to add an external bootstrap diode between

VIN(or VCC) and the BOOT pin to improve enhancement

of the internal MOSFET switch and improve efficiency.

The bootstrap diode can be a low cost one such as 1N4148

or BAT54.

R

EN2

RT2858B

GND

Figure 3. ResistorDivider for Lockout Threshold Setting

Output Voltage Setting

5V

Set the desired output voltage using a resistive divider

from the output to ground with the midpoint connected to

FB. The output voltage is set according to the following

BOOT

equation :

R1

RT2858B

SW

0.1µF

V_OUT= 0.765×(1+

)

R2

V

OUT

Figure 5. External Bootstrap Diode

R1

External BOOT Capacitor Series Resistance

FB

RT2858B

GND

R2

The internal power MOSFET switch gate driver is

optimized to turn the switch on fast enough for low power

loss and good efficiency, but also slow enough to reduce

EMI. Switch turn-on is when most EMI occurs since V_SW

rises rapidly. During switch turn-of, SW is discharged

relatively slowly by the inductor current during the dead-

time between high-side and low-switch on-times.

Figure 4. Output Voltage Setting

Place the FB resistors within 5mm of the FB pin. Choose

R2 between 10kΩ and 100kΩ to minimize power

consumption without excessive noise pick-up and

calculate R1 as follows :

In some cases it is desirable to reduce EMI further, at the

expense of some additional power dissipation. The switch

turn-on can be slowed by placing a small (<10Ω)

resistance between BOOT and the external bootstrap

capacitor. This will slow the high-side switch turn-on and

V_SW's rise. To remove the resistor from the capacitor

charging path (avoiding poor enhancement due to under-

charging the BOOT capacitor), use the external diode

shown in Figure 5 to charge the BOOT capacitor and place

the resistance between BOOT and the capacitor/diode

connection.

R2×(V

− 0.765V)

0.765V

OUT

R1 =

For output voltage accuracy, use divider resistors with 1%

or better tolerance.

Under-Voltage Lockout Protection

The RT2858B feature an Under-Voltage Lockout (UVLO)

function that monitors the internal linear regulator output

(V_PVDD) and prevents operation if V_PVDDis too low. In some

multiple input voltage applications, it may be desirable to

use a power input that is too low to allow V_PVDDto exceed

the UVLO threshold. In this case, if there is another low-

power supply available that is high enough to operate the

V_PVDDregulator, connecting that supply to VCC will allow

the IC to operate, using the lower-voltage high-power supply

for the DC/DC power path. Because of the internal linear

regulator, any supply regulated or unregulated) between

4.5V and 18V will operate the IC.

V_PVDDCapacitor Selection

Decouple V_PVDDto PGND with a 1μF ceramic capacitor.

High grade dielectric (X7R, or X5R) ceramic capacitors

are recommended for their stable temperature and bias

voltage characteristics.

©

is a registered trademark of Richtek Technology Corporation.

www.richtek.com

16

DS2858B-00 September 2013

RT2858B

Thermal Considerations

Recommendations for PCB Layout

For continuous operation, do not exceed absolute

maximum junction temperature. The maximum power

dissipation depends on the thermal resistance of the IC

package, PCB layout, rate of surrounding airflow, and

difference between junction and ambient temperature. The

maximum power dissipation can be calculated by the

following formula :

` 1 Ounce Copper on Top Layer, plated-up through SMT

PCB Mfg Process

` 1 Ounce Copper on Top Layer will improve Thermal

performance Minimum 4 Layer PCB Stack up.

` Place the shape with 70mm²as Figure 7 around the

PSOP-8 Footprint to achieve best thermal performance.

P_D(MAX)= (T_J(MAX)− T_A) / θ_JA

where T_J(MAX)is the maximum junction temperature, T_Ais

the ambient temperature, and θ_JAis the junction to ambient

thermal resistance.

For recommended operating condition specifications, the

maximum junction temperature is 125°C. The junction to

ambient thermal resistance, θ_JA, is layout dependent. For

SOP-8 (Exposed Pad) package, the thermal resistance,

θ_JA, is 49°C/W on a standard JEDEC 51-7 four-layer

thermal test board. The maximum power dissipation at

T_A= 25°C can be calculated by the following formula :

Copper Area = 70mm², θ_JA= 49°C/W

Figure 7. PCB CopperArea

` Utilize Standard PTH (Plated Through Hole, 25mil

diameter, as Figure 8) to Via down from Exposed Pad

on Top Layer, to GND Plane on PCB.

P_D(MAX)= (125°C − 25°C) / (49°C/W) = 2.041W for

SOP-8 (Exposed Pad) package

The maximum power dissipation depends on the operating

ambient temperature for fixed T_J(MAX)and thermal

resistance, θ_JA. The derating curve in Figure 6 allows the

designer to see the effect of rising ambient temperature

on the maximum power dissipation.

Figure 8. Standard PTH toGNDPlane

3.0

Four-Layer PCB

2.5

2.0

1.5

1.0

0.5

0.0

0

25

50

75

100

125

Ambient Temperature (°C)

Figure 6. Derating Curve of Maximum PowerDissipation

©

is a registered trademark of Richtek Technology Corporation.

DS2858B-00 September 2013

www.richtek.com

17

RT2858B

Layout Consideration

` SW node is with high frequency voltage swing and

should be kept at small area. Keep sensitive

components away from the SW node to prevent stray

capacitive noise pickup.

Follow the PCB layout guidelines for optimal performance

of the RT2858B

` Keep the traces of the main current paths as short and

wide as possible.

` Connect feedback network behind the output capacitors.

Keep the loop area small. Place the feedback

components near the RT2858B feedback pin.

` Put the input capacitor as close as possible to the device

pins (VINandGND).

` The GND and Exposed Pad should be connected to a

strong ground plane for heat sinking and noise protection.

The resistor divider must be connected

as close to the device as possible.

Input capacitor must be placed

as close to the IC as possible.

C1

C2

V

OUT

SW should be connected to inductor by

wide and short trace. Keep sensitive

components away from this trace.

R1

8

7

6

5

EN

FB

VIN

R2

C4

C5

2

3

4

BOOT

GND

C6

GND

PVCC

SS

SW

9

L1

GND

C7

Figure 9. PCB Layout Guide

©

is a registered trademark of Richtek Technology Corporation.

www.richtek.com

18

DS2858B-00 September 2013

RT2858B

Outline Dimension

H

A

Y

M

EXPOSED THERMAL PAD

(Bottom of Package)

J

B

X

F

C

I

D

Dimensions In Millimeters Dimensions In Inches

Symbol

Min

Max

5.004

4.000

1.753

0.510

1.346

0.254

0.152

6.200

1.270

2.300

2.500

3.500

Min

Max

A

B

C

D

F

H

I

4.801

3.810

1.346

0.330

1.194

0.170

0.000

5.791

0.406

2.000

2.100

3.000

0.189

0.150

0.053

0.013

0.047

0.007

0.000

0.228

0.016

0.079

0.083

0.118

0.197

0.157

0.069

0.020

0.053

0.010

0.006

0.244

0.050

0.091

0.098

0.138

J

M

X

Y

X

Y

Option 1

Option 2

8-Lead SOP (Exposed Pad) Plastic Package

Richtek Technology Corporation

14F, No. 8, Tai Yuen 1^stStreet, Chupei City

Hsinchu, Taiwan, R.O.C.

Tel: (8863)5526789

Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should

obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot

assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be

accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third

parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.

DS2858B-00 September 2013

www.richtek.com

19


型号：	RT2858B
厂家：	RICHTEK TECHNOLOGY CORPORATION
描述：	暂无描述
文件：	总19页 (文件大小：279K)
中文：	中文翻译
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