MIC26400YJL [MICREL]
5A Hyper Speed Control Synchronous DC/DC Buck Regulator; 5A超调速同步DC / DC降压稳压器
| 型号： | MIC26400YJL |
| 厂家： | 
MICREL SEMICONDUCTOR |
| 描述： | 5A Hyper Speed Control Synchronous DC/DC Buck Regulator |
| 文件： | 总27页 (文件大小：1044K) |
| 中文： | 中文翻译 | 下载： | 下载PDF数据表文档文件 |
MIC26400
5A Hyper Speed ControlTM
Synchronous DC/DC Buck Regulator
SuperSwitcher IITM
General Description
Features
•
Hyper Speed ControlTM architecture enables
- High delta V operation (VIN = 26V and VOUT = 0.8V)
- Small output capacitance
The Micrel MIC26400 is a constant-frequency, synchronous
buck regulator featuring a unique digitally modified adaptive
ON-time control architecture. The MIC26400 operates over
an input supply range of 4.5V to 26V and provides a
regulated output at up to 5A of output current. The output
voltage is adjustable down to 0.8V with a typical accuracy of
±1%, and the device operates at a switching frequency of
300kHz.
•
•
•
4.5V to 26V input voltage
Output down to 0.8V with ±1% accuracy
Any CapacitorTM Stable
- Zero ESR to high-ESR output capacitance
Micrel’s Hyper Speed ControlTM architecture allows for ultra-
fast transient response while reducing the output capacitance
and also makes (High VIN)/(Low VOUT) operation possible.
This digitally modified adaptive tON ripple control architecture
combines the advantages of fixed frequency operation and
fast transient response in a single device.
•
•
•
•
•
•
5A output current capability
300kHz switching frequency
Internal compensation
Up to 95% efficiency
6ms Internal soft-start
Foldback current limit and “hiccup” mode short-circuit
protection
The MIC26400 offers a full suite of protection features to
ensure protection of the IC during fault conditions. These
include undervoltage lockout to ensure proper operation
under power-sag conditions, internal soft-start to reduce
inrush current, foldback current limit, “hiccup” mode short-
circuit protection and thermal shutdown.
•
•
•
•
Thermal shutdown
Supports safe start-up into a pre-biased load
–40°C to +125°C junction temperature range
28-pin 5mm X 6mm MLF® package
All support documentation can be found on Micrel’s web
site at: www.micrel.com.
Applications
•
•
•
•
Distributed power systems
Communications/networking infrastructure
Set-top box, gateways and routers
Printers, scanners, graphic cards and video cards
____________________________________________________________________________________________________________
Typical Application
Hyper Speed Control, SuperSwitcher II and Any Capacitor are trademarks of Micrel, Inc.
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc.
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
M9999-070110-A
July 2010
Micrel, Inc.
MIC26400
Ordering Information
Junction Temperature
Range
Lead
Finish
Part Number
Voltage
Switching Frequency
Package
MIC26400YJL
Adjustable
300kHz
–40°C to +125°C
28-Pin 5mm X 6mm MLF®
Pb-Free
Pin Configuration
28-Pin 5mm X 6mm MLF® (YJL)
Pin Description
Pin Number
Pin Name Pin Function
High-Side N-internal MOSFET Drain Connection (Input): The PVIN operating voltage range is from
4.5V to 26V. Input capacitors between the PVIN pins and the power ground (PGND) are required.
Note that the connection must be kept short.
13, 14, 15,
PVIN
EN
16, 17, 18, 19
Enable (Input): A logic level control of the output. The EN pin is CMOS-compatible. Logic high or
floating = enable, logic low = shutdown. In the off state, the VDD supply current of the device is
reduced (typically 0.7mA).
24
Feedback (Input): Input to the transconductance amplifier of the control loop. The FB pin is regulated
to 0.8V. A resistor divider connecting the feedback to the output is used to adjust the desired output
voltage.
25
26
27
FB
Signal ground. SGND must be connected directly to the ground planes. Do not route the SGND pin
to the PGND Pad on the top layer. (see PCB Layout Guidelines for details.)
SGND
VDD
VDD Bias (Input): Power to the internal reference and control sections of the MIC26400. The VDD
operating voltage range is from 4.5V to 5.5V. A 2.2µF ceramic capacitor from the VDD pin-to-PGND
must be placed next to the IC.
Power Ground. PGND is the ground path for the MIC26400 buck converter power stage. The PGND
pin connects to the sources of low-side N-Channel internal MOSFETs, the negative terminals of
input capacitors, and the negative terminals of output capacitors. The loop for the power ground
should be as small as possible and separate from the Signal ground (SGND) loop.
2, 5, 6, 7, 8, 21
PGND
CS
Current Sense (Input): High current output driver return. The CS pin connects directly to the switch
node. Due to the high speed switching on this pin, the CS pin should be routed away from sensitive
nodes. CS pin also senses the current by monitoring the voltage across the low-side internal
MOSFET during OFF-time.
22
Boost (Output): Bootstrapped voltage to the high-side N-channel internal MOSFET driver. A Schottky
diode is connected between the VDD pin and the BST pin. A boost capacitor of 0.1μF is connected
between the BST pin and the SW pin.
20
BST
SW
Switch Node (Output): Internal connection for the high-side MOSFET source and low-side MOSFET
drain.
4, 9, 10, 11, 12
23
VIN
NC
Power Supply Voltage (Input): Requires bypass capacitor to SGND.
No Connect.
1, 3, 28
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M9999-070110-A
July 2010
Micrel, Inc.
MIC26400
Absolute Maximum Ratings(1, 2)
Operating Ratings(3)
Supply Voltage (PVIN, VIN)................................. 4.5V to 26V
Bias Voltage (VDD)............................................ 4.5V to 5.5V
Enable Input (VEN)................................................. 0V to VDD
Junction Temperature (TJ) ........................−40°C to +125°C
Maximum Power Dissipation......................................Note 4
Package Thermal Resistance(4)
PVIN to PGND................................................ −0.3V to +28V
VIN to PGND ....................................................−0.3V to PVIN
VDD to PGND ................................................... −0.3V to +6V
VSW, VCS to PGND..............................−0.3V to (PVIN +0.3V)
VBST to VSW ........................................................ −0.3V to 6V
VBST to PGND.................................................. −0.3V to 34V
5mm x 6mm MLF® (θJA) ....................................36°C/W
VEN to PGND ......................................−0.3V to (VDD + 0.3V)
VFB to PGND.......................................−0.3V to (VDD + 0.3V)
PGND to SGND ........................................... −0.3V to +0.3V
Junction Temperature ..............................................+150°C
Storage Temperature (TS).........................−65°C to +150°C
Lead Temperature (soldering, 10sec)........................ 260°C
Electrical Characteristics(5)
PVIN = VIN =12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Power Supply Input
Input Voltage Range (VIN, PVIN)
VDD Bias Voltage
4.5
26
V
Operating Bias Voltage (VDD
)
4.5
2.4
5
5.5
3.2
V
V
Under-Voltage Lockout Trip Level
UVLO Hysteresis
VDD Rising
VFB = 1.5V
2.7
50
1.4
mV
mA
Quiescent Supply Current
3
2
VDD = VBST = 5.5V, VIN = 26V
Shutdown Supply Current
Reference
0.7
mA
SW = unconnected, VEN = 0V
0°C ≤ TJ ≤ 85°C (±1.0%)
−40°C ≤ TJ ≤ 125°C (±1.5%)
IOUT = 0A to 5A
0.792
0.788
0.8
0.8
0.2
0.1
5
0.808
0.812
Feedback Reference Voltage
V
Load Regulation
Line Regulation
FB Bias Current
Enable Control
EN Logic Level High
EN Logic Level Low
EN Bias Current
Oscillator
%
%
VIN = (VOUT + 3.0V) to 26V
VFB = 0.8V
nA
4.5V < VDD < 5.5V
4.5V < VDD < 5.5V
VEN = 0V
1.2
0.85
0.78
50
V
V
0.4
µA
Switching Frequency (6)
Maximum duty cycle (7)
Minimum duty cycle
Minimum Off-time
Soft-Start
225
300
87
375
kHz
%
VFB = 0V
VFB > 0.8V
0
%
360
ns
Soft-Start time
6
ms
3
M9999-070110-A
July 2010
Micrel, Inc.
MIC26400
Electrical Characteristics(5) (Continued)
PVIN = VIN =12V, VDD = 5V; VBST – VSW = 5V; TA = 25°C, unless noted. Bold values indicate −40°C ≤ TJ ≤ +125°C.
Parameter
Condition
Min.
Typ.
Max.
Units
Short Circuit Protection
Current-Limit Threshold
Short Circuit Current
Internal FETs
VFB = 0.8V
VFB = 0V
6
13
6
A
A
Top-MOSFET RDS (ON)
Bottom-MOSFET RDS (ON)
SW Leakage Current
VIN Leakage Current
Thermal Protection
Over-temperature Shutdown
ISW = 1A
43
mΩ
mΩ
µA
ISW = 1A
12.5
VIN = 26V, VSW = 26V, VEN = 0V, VBST = 31.5 V
VIN = 26V, VSW = 0V, VEN = 0V, VBST = 31.5V
60
25
µA
TJ Rising
155
10
°C
°C
Over-temperature Shutdown
Hysteresis
Notes:
1. Exceeding the absolute maximum rating may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5kΩ in series with 100pF.
3. The device is not guaranteed to function outside operating range.
4. PD(MAX) = (TJ(MAX) – TA)/ θJA, where θJA depends upon the printed circuit layout. See “Applications Information.”
5. Specification for packaged product only.
6. Measured in test mode.
7. The maximum duty-cycle is limited by the fixed mandatory off-time tOFF of typically 360ns.
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July 2010
Micrel, Inc.
MIC26400
Typical Characteristics
VIN Shutdown Current
vs. Input Voltage
VDD Operating Supply Current
vs. Input Voltage
VIN Operating Supply Current
vs. Input Voltage
10.0
20
16
12
8
10
8
8.0
6.0
6
4.0
4
VOUT = 1.2V
VOUT = 1.2V
IOUT = 0A
VDD = 5V
VEN = 0V
VDD= 5V
2.0
0.0
4
2
VDD = 5V
SWITCHING
SWITCHING
0
0
4
10
16
22
28
28
28
4
10
16
22
28
4
10
16
22
28
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Feedback Voltage
vs. Input Voltage
Current Limit
vs. Input Voltage
Total Regulation
vs. Input Voltage
0.808
0.804
0.800
0.796
0.792
1.0%
0.8%
0.6%
0.4%
0.2%
0.0%
20
15
10
5
VOUT = 1.2V
VDD = 5V
IOUT = 0A to 5A
VOUT = 1.2V
VOUT = 1.2V
VDD = 5V
IOUT = 0A
VDD = 5V
0
4
10
16
22
4
10
16
22
28
4
10
16
22
28
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Switching Frequency
vs. Input Voltage
390
345
300
255
210
VOUT = 1.2V
VDD = 5V
IOUT = 0A
4
10
16
22
INPUT VOLTAGE (V)
5
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July 2010
Micrel, Inc.
MIC26400
Typical Characteristics (Continued)
VDD Operating Supply Current
vs. Temperature
VDD Shutdown Current
vs. Temperature
VDD UVLO Threshold
vs. Temperature
10.0
8.0
6.0
4.0
2.0
0.0
1
0.8
0.6
0.4
0.2
0
2.8
2.7
2.6
2.5
2.4
2.3
VIN = 12V
VOUT = 1.2V
VDD = 5V
Rising
IOUT = 0A
SWITCHING
Falling
VIN = 12V
IOUT = 0A
VDD = 5V
VEN = 0V
VIN = 12V
-50
-20
10
40
70
100
130
-50
-20
10
40
70
100
130
130
130
130
-50
-20
10
40
70
100
130
130
130
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
VIN Shutdown Current
vs. Temperature
VIN Operating Supply Current
vs. Temperature
Current Limit
vs. Temperature
10.0
8.0
6.0
4.0
2.0
0.0
10.0
8.0
6.0
4.0
2.0
0.0
25
20
15
10
5
VIN = 12V
VOUT = 1.2V
VDD = 5V
IOUT = 0A
SWITCHING
VIN = 12V
VDD = 5V
VIN = 12V
VOUT = 1.2V
VDD = 5V
IOUT = 0A
0
-50
-20
10
40
70
100
130
-50
-20
-20
-20
10
40
70
100
-50
-20
10
40
70
100
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
Feedback Voltage
vs. Temperature
Line Regulation
vs. Temperature
Load Regulation
vs. Temperature
0.808
0.804
0.800
0.796
0.792
1.0%
0.8%
0.6%
0.4%
0.2%
0.0%
0.5%
VIN = 12V
VIN = 12V
VOUT = 1.2V
VDD = 5V
IOUT = 0A
VIN = 6V to 26V
VOUT = 1.2V
VDD = 5V
0.4%
0.3%
0.2%
0.1%
0.0%
VOUT = 1.2V
VDD = 5V
IOUT = 0A to 5A
-50
-20
10
40
70
100 130
-50
10
40
70
100
-50
-20
10
40
70
100
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
EN Bias Current
vs. Temperature
Switching Frequency
vs. Temperature
345
330
315
300
285
270
255
100
80
60
40
20
0
VIN = 12V
VOUT = 1.2V
VDD = 5V
IOUT = 0A
VIN = 12V
VOUT = 1.2V
VDD = 5V
-50
-20
10
40
70
100
130
-50
10
40
70
100
TEMPERATURE (°C)
TEMPERATURE (°C)
6
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July 2010
Micrel, Inc.
MIC26400
Typical Characteristics (Continued)
Output Voltage
vs. Output Current
Efficiency
vs. Output Current
Feedback Voltage
vs. Output Current
1.212
1.208
1.204
1.2
100
95
90
85
80
75
70
65
60
55
50
0.808
0.804
0.800
0.796
0.792
12VIN
24VIN
1.196
1.192
1.188
VIN = 12V
OUT = 1.2V
VDD = 5V
VIN = 12V
VOUT = 1.2V
VDD = 5V
VOUT = 1.2V
VDD = 5V
V
0
1
2
3
4
5
0
1
2
3
4
5
0
1
2
3
4
5
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Line Regulation
vs. Output Current
Switching Frequency
vs. Output Current
Output Voltage (VIN = 5V)
vs. Output Current
0.5%
0.4%
0.3%
0.2%
0.1%
0.0%
390
360
330
300
270
240
210
5
4.8
4.6
4.4
4.2
4
VIN = 6V to 26V
VOUT = 1.2V
VDD = 5V
VIN = 5V
VFB < 0.8V
VDD = 5V
TA
25ºC
85ºC
125ºC
VIN = 12V
VOUT = 1.2V
VDD = 5V
3.8
3.6
3.4
0
0
1
2
3
4
5
0
1
2
3
4
5
1
2
3
4
5
6
7
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Die Temperature* (VIN = 5V)
vs. Output Current
Die Temperature* (VIN = 12V)
vs. Output Current
Die Temperature* (VIN = 24V)
vs. Output Current
60
40
20
60
40
20
0
60
40
20
VIN = 12V
VOUT = 1.2V
VDD= 5V
VIN = 24V
VOUT = 1.2V
VDD = 5V
VIN = 5V
VOUT = 1.2V
VDD= 5V
0
0
0
0
1
2
3
4
5
0
1
2
3
4
5
1
2
3
4
5
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
Die Temperature* : The temperature measurement was taken at the hottest point on the MIC26400 case mounted on a 5 square inch PCB, see
Thermal Measurement section. Actual results will depend upon the size of the PCB, ambient temperature and proximity to other heat emitting
components.
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July 2010
Micrel, Inc.
MIC26400
Typical Characteristics (Continued)
Efficiency (VIN = 5V)
vs. Output Current
Efficiency (VIN = 24V)
vs. Output Current
Efficiency (VIN = 12V)
vs. Output Current
100
95
100
95
90
85
80
75
70
95
90
85
80
75
70
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
5.0V
3.3V
2.5V
1.8V
1.5V
1.2V
1.0V
0.9V
0.8V
3.3V
2.5V
1.8V
1.5V
90
1.0V
0.9V
0.8V
1.2V
1.0V
0.9V
0.8V
85
80
0
1
2
3
4
5
6
7
0
1
2
3
4
5
6
7
0
1
2
3
4
5
6
7
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
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Micrel, Inc.
MIC26400
Functional Characteristics
9
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Micrel, Inc.
MIC26400
Functional Characteristics (Continued)
10
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Micrel, Inc.
MIC26400
Functional Characteristics (Continued)
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Micrel, Inc.
MIC26400
Functional Diagram
Figure 1. MIC26400 Block Diagram
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Micrel, Inc.
MIC26400
The maximum duty cycle is obtained from the 360ns
Functional Description
tOFF(min)
:
The MIC26400 is an adaptive ON-time synchronous
step-down DC/DC regulator. It is designed to operate
over a wide input voltage range from 4.5V to 26V and
provides a regulated output voltage at up to 5A of output
current. A digitally modified adaptive ON-time control
scheme is employed in to obtain a constant switching
frequency and to simplify the control compensation.
Over-current protection is implemented without the use
of an external sense resistor. The device includes an
internal soft-start function which reduces the power
supply input surge current at start-up by controlling the
output voltage rise time.
tS − tOFF(min)
360ns
tS
Dmax
=
= 1−
(2)
tS
where tS = 1/300kHz = 3.33μs. It is not recommended to
use MIC26400 with a OFF-time close to tOFF(min) during
steady-state operation.
The actual ON-time and resulting switching frequency
will vary with the part-to-part variation in the rise and fall
times of the internal MOSFETs, the output load current,
and variations in the VDD voltage. Also, the minimum tON
results in a lower switching frequency in high VIN to VOUT
applications, such as 26V to 1.0V. The minimum tON
measured on the MIC26400 evaluation board is about
184ns. During load transients, the switching frequency is
changed due to the varying OFF-time.
Theory of Operation
Figure 1 illustrates the block diagram for the control loop
of the MIC26400. The output voltage is sensed by the
MIC26400 feedback pin FB via the voltage divider R1
and R2, and compared to a 0.8V reference voltage VREF
at the error comparator through
a
low gain
To illustrate the control loop operation, we will analyze
both the steady-state and load transient scenarios. For
easy analysis, the gain of the gm amplifier is assumed to
be 1. With this assumption, the inverting input of the
error comparator is the same as the feedback voltage.
transconductance (gm) amplifier. If the feedback voltage
decreases and the output of the gm amplifier is below
0.8V, then the error comparator will trigger the control
logic and generate an ON-time period. The ON-time
period length is predetermined by the “FIXED tON
ESTIMATION” circuitry:
Figure 2 shows the MIC26400 control loop timing during
steady-state operation. During steady-state, the gm
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ON-
time is predetermined by the tON estimator. The
termination of the OFF-time is controlled by the feedback
voltage. At the valley of the feedback voltage ripple,
which occurs when VFB falls below VREF, the OFF period
ends and the next ON-time period is triggered through
the control logic circuitry.
VOUT
tON(estimated)
=
(1)
VIN × 300kHz
Where VOUT is the output voltage and VIN is the power
stage input voltage.
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases and
the output of the gm amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
360ns, the MIC26400 control logic will apply the tOFF(min)
instead. tOFF(min) is required to maintain enough energy in
the boost capacitor (CBST) to drive the high-side
MOSFET.
Figure 2. MIC26400 Control Loop Timing
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MIC26400
Figure 3 shows the operation of the MIC26400 during a
load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF. This will cause the error comparator to trigger
an ON-time period. At the end of the ON-time period, a
minimum OFF-time tOFF(min) is generated to charge CBST
since the feedback voltage is still below VREF. Then, the
next ON-time period is triggered due to the low feedback
voltage. Therefore, the switching frequency changes
during the load transient, but returns to the nominal fixed
frequency once the output has stabilized at the new load
current level. With the varying duty cycle and switching
frequency, the output recovery time is fast and the
output voltage deviation is small in MIC26400 converter.
Soft-Start
Soft-start reduces the power supply input surge current
at startup by controlling the output voltage rise time. The
input surge appears while the output capacitor is
charged up. A slower output rise time will draw a lower
input surge current.
The MIC26400 implements an internal digital soft-start
by making the 0.8V reference voltage VREF ramp from 0
to 100% in about 6ms with 9.7mV steps. Therefore, the
output voltage is controlled to increase slowly by a stair-
case VFB ramp. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption. VDD must be powered up at the same time
or after VIN to make the soft-start function correctly.
Current Limit
The MIC26400 uses the RDS(ON) of the internal low-side
power MOSFET to sense over-current conditions. This
method will avoid adding cost, board space and power
losses taken by a discrete current sense resistor. The
low-side MOSFET is used because it displays much
lower parasitic oscillations during switching than the
high-side MOSFET.
In each switching cycle of the MIC26400 converter, the
inductor current is sensed by monitoring the low-side
MOSFET in the OFF period. If the inductor current is
greater than 13A, then the MIC26400 turns off the high-
side MOSFET and a soft-start sequence is triggered.
This mode of operation is called “hiccup mode” and its
purpose is to protect the downstream load in case of a
hard short. The current-limit threshold has a foldback
characteristic related to the feedback voltage, as shown
in Figure 4.
Figure 3. MIC26400 Load Transient Response
Unlike true current-mode control, the MIC26400 uses the
output voltage ripple to trigger an ON-time period. The
output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC26400 control loop has the advantage
of eliminating the need for slope compensation.
In order to meet the stability requirements, the
MIC26400 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gm amplifier and the error comparator.
The recommended feedback voltage ripple is
20mV~100mV. If a low-ESR output capacitor is selected,
the feedback voltage ripple may be too small to be
sensed by the gm amplifier and the error comparator.
Also, the output voltage ripple and the feedback voltage
ripple are not necessarily in phase with the inductor
current ripple if the ESR of the output capacitor is very
low. In these cases, ripple injection is required to ensure
proper operation. Please refer to “Ripple Injection”
subsection in “Application Information” for more details
about the ripple injection technique.
Current-Limit Thresold
vs. Feedback Voltage
20.0
16.0
12.0
8.0
4.0
0.0
0.0
0.2
0.4
0.6
0.8
1.0
FEEDBACK VOLTAGE (V)
Figure 4. MIC26400 Current Limit Foldback Characteristic
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MIC26400
the power stroke (high-side switching) cycle, i.e. ΔBST =
10mA x 3.33μs/0.1μF = 333mV. When the low-side
MOSFET is turned back on, CBST is recharged through
D1. A small resistor RG, which is in series with CBST, can
be used to slow down the turn-on time of the high-side
N-channel MOSFET.
MOSFET Gate Drive
The Block Diagram of Figure 1 shows a bootstrap circuit,
consisting of D1 (a Schottky diode is recommended) and
CBST. This circuit supplies energy to the high-side drive
circuit. Capacitor CBST is charged, while the low-side
MOSFET is on, and the voltage on the SW pin is
approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the high-side MOSFET turns on, the voltage on
the SW pin increases to approximately VIN. Diode D1 is
reversed biased and CBST floats high while continuing to
keep the high-side MOSFET on. The bias current of the
high-side driver is less than 10mA so a 0.1μF to 1μF is
sufficient to hold the gate voltage with minimal droop for
The drive voltage is derived from the VDD supply voltage.
The nominal low-side gate drive voltage is VDD and the
nominal high-side gate drive voltage is approximately
VDD – VDIODE, where VDIODE is the voltage drop across
D1. An approximate 30ns delay between the high-side
and low-side driver transitions is used to prevent current
from simultaneously flowing unimpeded through both
MOSFETs.
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MIC26400
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels. The
winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by the equation below:
Application Information
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by the equation below:
PINDUCTOR(Cu) = IL(RMS)2 × RWINDING
(8)
The resistance of the copper wire, RWINDING, increases
with the temperature. The value of the winding
resistance used should be at the operating temperature.
VOUT × (V
− VOUT )
IN(max)
L =
(4)
V
IN(max) × fsw × 20%×IOUT(max)
PWINDING(Ht)
=
RWINDING(20°C) × (1 + 0.0042 × (TH – T20°C))
(9)
Where:
SW = switching frequency, 300kHz
20% = ratio of AC ripple current to DC output current
IN(max) = maximum power stage input voltage
The peak-to-peak inductor current ripple is:
Where:
TH = temperature of wire under full load
20°C = ambient temperature
f
T
V
RWINDING(20°C) = room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
VOUT ×(V
− VOUT )
IN(max)
ΔIL(pp)
=
(5)
The type of the output capacitor is usually determined by
its ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors for
selecting the output capacitor. Recommended capacitor
types are tantalum, low-ESR aluminum electrolytic, OS-
CON and POSCAP. The output capacitor’s ESR is
usually the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. The maximum value of ESR is
calculated:
VIN(max) × fsw ×L
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor current
ripple.
IL(pk) =IOUT(max) + 0.5 × ΔIL(pp)
(6)
The RMS inductor current is used to calculate the I2R
losses in the inductor.
ΔVOUT(pp)
ESRC
≤
(10)
OUT
ΔIL(PP)
2
ΔIL(PP)
2
IL(RMS) = IOUT(max)
+
(7)
12
Where:
ΔVOUT(pp) = peak-to-peak output voltage ripple
ΔIL(PP) = peak-to-peak inductor current ripple
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC26400 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used
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MIC26400
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated below:
The peak input current is equal to the peak inductor
current, so:
ΔVIN = IL(pk) × ESRCIN
(14)
2
ΔIL(PP)
⎛
⎜
⎜
⎝
⎞
⎟
⎟
⎠
2
ΔVOUT(pp)
=
+
(
ΔIL(PP) × ESRC
)
OUT
COUT × fSW × 8
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
(11)
Where:
D = duty cycle
OUT = output capacitance value
SW = switching frequency
ICIN(RMS) ≈ IOUT(max) × D×(1−D)
(15)
C
f
The power dissipated in the input capacitor is:
As described in the “Theory of Operation” subsection in
“Functional Description”, the MIC26400 requires at least
20mV peak-to-peak ripple at the FB pin to make the gm
amplifier and the error comparator behave properly. Also,
the output voltage ripple should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” subsection for more details.
P
DISS(CIN) = ICIN(RMS)2 × ESRCIN
(16)
Voltage Setting Components
The MIC26400 requires two resistors to set the output
voltage as shown in Figure 5.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated below:
Figure 5. Voltage-Divider Configuration
ΔIL(PP)
IC
=
(12)
The output voltage is determined by the equation:
OUT (RMS)
12
R1
VOUT = VFB ×(1+
)
(17)
The power dissipated in the output capacitor is:
R2
2
Where VFB = 0.8V. A typical value of R1 can be between
3kꢀ and 10kꢀ. If R1 is too large, it may allow noise to be
introduced into the voltage feedback loop. If R1 is too
small in value, it will decrease the efficiency of the power
supply, especially at light loads. Once R1 is selected, R2
can be calculated using:
PDISS(C
= IC
× ESRC
(13)
)
OUT (RMS)
OUT
OUT
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR.
VFB ×R1
R2 =
(18)
VOUT − VFB
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Micrel, Inc.
MIC26400
Ripple Injection
The VFB ripple required for proper operation of the
MIC26400 gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. For a low
output voltage, such as a 1V, the output voltage ripple is
only 10mV to 20mV, and the feedback voltage ripple is
less than 20mV. If the feedback voltage ripple is so small
that the gm amplifier and error comparator can’t sense it,
the MIC26400 will lose control and the output voltage is
not regulated. In order to have some amount of VFB
ripple, a ripple injection method is applied for low output
voltage ripple applications.
Figure 6b. Inadequate Ripple at FB
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
1) Enough ripple at the feedback voltage due to the large
ESR of the output capacitors.
As shown in Figure 6a, the converter is stable without
any ripple injection. The feedback voltage ripple is:
R2
ΔVFB(pp)
=
× ESRC
× ΔIL
(19)
(pp)
OUT
Figure 6c. Invisible Ripple at FB
R1+ R2
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 6c. The injected ripple
is:
where ΔIL(pp) is the peak-to-peak value of the inductor
current ripple.
2) Inadequate ripple at the feedback voltage due to the
small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin through a
feedforward capacitor Cff in this situation, as shown in
Figure 6b. The typical Cff value is between 1nF and
100nF. With the feedforward capacitor, the feedback
voltage ripple is very close to the output voltage ripple:
1
ΔVFB(pp) = VIN ×Kdiv ×D×(1-D)×
(21)
(22)
fSW ×τ
R1//R2
Kdiv
=
Rinj + R1//R2
ΔVFB(pp) ≈ ESR × ΔIL
(20)
(pp)
Where:
3) Virtually no ripple at the FB pin voltage due to the very
low ESR of the output capacitors.
VIN = Power stage input voltage
D = duty cycle
fSW = switching frequency
τ = (R1//R2//Rinj) × Cff
In equations (21) and (22), it is assumed that the time
constant associated with Cff must be much greater than
the switching period:
1
T
=
<< 1
(23)
fSW ×τ
τ
Figure 6a. Enough Ripple at FB
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Micrel, Inc.
MIC26400
If the voltage divider resistors R1 and R2 are in the kꢀ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
Thermal Measurements
Measuring the IC’s case temperature is recommended to
insure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kꢀ range.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer.
If a thermal couple wire is used, it must be constructed
of 36 gauge wire or higher (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36)
is adequate for most applications.
Step 2. Select Rinj according to the expected feedback
voltage ripple using equation (24),
ΔVFB(pp)
fSW ×τ
D× (1− D)
Kdiv
=
×
(24)
V
IN
Then the value of Rinj is obtained as:
1
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
Rinj = (R1//R2) × (
− 1)
(25)
Kdiv
Step 3. Select Cinj as 100nF, which could be considered
as short for a wide range of the frequencies.
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Micrel, Inc.
MIC26400
Inductor
PCB Layout Guidelines
•
•
•
•
Keep the inductor connection to the switch node
(SW) short.
Warning!!! To minimize EMI and output noise, follow
these layout recommendations.
Do not route any digital lines underneath or close to
the inductor.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
Keep the switch node (SW) away from the feedback
(FB) pin.
The following guidelines should be followed to insure
proper operation of the MIC26400 converter.
The CS pin should be connected directly to the SW
pin to accurate sense the voltage across the low-
side MOSFET.
IC
•
To minimize noise, place a ground plane underneath
the inductor.
•
The 2.2µF ceramic capacitor, which is connected to
the VDD pin, must be located right at the IC. The
VDD pin is very noise sensitive and placement of the
capacitor is very critical. Use wide traces to connect
to the VDD and PGND pins.
Output Capacitor
•
Use a wide trace to connect the output capacitor
ground terminal to the input capacitor ground
terminal.
•
The signal ground pin (SGND) must be connected
directly to the ground planes. Do not route the
SGND pin to the PGND Pad on the top layer.
•
Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
•
•
Place the IC close to the point of load (POL).
Use fat traces to route the input and output power
lines.
•
The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high current
load trace can degrade the DC load regulation.
•
Signal and power grounds should be kept separate
and connected at only one location.
RC Snubber
Place the RC snubber on the same side of the board
and as close to the SW pin as possible.
Input Capacitor
•
•
•
Place the input capacitor next.
Place the input capacitors on the same side of the
board and as close to the IC as possible.
•
•
Keep both the PVIN and PGND connections short.
Place several vias to the ground plane close to the
input capacitor ground terminal.
•
•
Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.
•
•
If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.
In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the over-
voltage spike seen on the input supply with power is
suddenly applied.
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Micrel, Inc.
MIC26400
Evaluation Board Schematic
Figure 7. Schematic of MIC26400 Evaluation Board
(J13, R13, R15 are for testing purposes)
21
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Micrel, Inc.
MIC26400
Bill of Materials
Item
Part Number
Manufacturer
EPCOS(1)
AVX(2)
Murata(3)
AVX
Description
Qty
C1
B41125A7227M
220µF Aluminum Capacitor, SMD, 35V
1
12105C475KAZ2A
C2, C3
4.7µF Ceramic Capacitor, X7R, Size 1210, 50V
2
GRM32ER71H475KA88L
12106D107MAT2A
C4, C5, C13
100µF Ceramic Capacitor, X5R, Size 1210, 6.3V
0.1µF Ceramic Capacitor, X7R, Size 0603, 50V
1
4
GRM32ER60J107ME20L
06035C104KAT2A
Murata
AVX
C6, C7, C10
C8, C9
C11
GRM188R71H104KA93D
C1608X7R1H104K
Murata
TDK(4)
AVX
0805ZC225MAT2A
GRM21BR71A225KA01L
C2012X7R1A225K
2.2µF Ceramic Capacitor, X7R, Size 0805, 10V
1nF Ceramic Capacitor, X7R, Size 0603, 50V
22nF Ceramic Capacitor, X7R, Size 0603, 50V
2
1
1
Murata
TDK
06035C102KAT2A
AVX
GRM188R71H102KA01D
C1608X7R1H102K
Murata
TDK
AVX
06035C223KAZ2A
GRM188R71H223K
C12
Murata
TDK
C1608X7R1H223K
Open
C14
C15
Open
SD103AWS-7
SD103AWS
CMDZ5L6
Diodes Inc(6)
Vishay(7)
Central Semi(8)
D1
Small Signal Schottky Diode
5.6V Zener Diode
1
D2
1
1
1
1
1
2
1
1
1
1
Cooper Bussmann(9) 4.0µH Inductor, 12A Saturation Current
L1
HCF1305-4R0-R
FCX619
Q1
ZETEX
50V NPN Transistor
R1
CRCW06034R75FKEA
CRCW08051R21FKEA
CRCW060310K0FKEA
CRCW060380K6FKEA
CRCW060340K2FKEA
CRCW060320K0FKEA
CRCW060311K5FKEA
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
4.75ꢀ Resistor, Size 0603, 1%
1.21ꢀ Resistor, Size 0805, 1%
10kꢀ Resistor, Size 0603, 1%
80.6kꢀ Resistor, Size 0603, 1%
40.2kꢀ Resistor, Size 0603, 1%
20kꢀ Resistor, Size 0603, 1%
11.5kꢀ Resistor, Size 0603, 1%
R2, R16
R3, R4
R5
R6
R7
R8
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July 2010
Micrel, Inc.
MIC26400
Bill of Materials (Continued)
Item
R9
Part Number
Manufacturer
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Vishay Dale
Micrel. Inc.(10)
Description
Qty
1
CRCW06038K06FKEA
CRCW06034K75FKEA
CRCW06033K24FKEA
CRCW06031K91FKEA
CRCW06030000FKEA
CRCW06035K23FKEA
CRCW060349R9FKEA
MIC26400YJL
8.06kꢀ Resistor, Size 0603, 1%
4.75kꢀ Resistor, Size 0603, 1%
3.24kꢀ Resistor, Size 0603, 1%
1.91kꢀ Resistor, Size 0603, 1%
0ꢀ Resistor, Size 0603, 5%
R10
R11
R12
R13
R14
R15
U1
1
1
1
1
5.23kꢀ Resistor, Size 0603, 1%
49.9ꢀ Resistor, Size 0603, 1%
26V/5A Synchronous Buck DC/DC Regulator
1
1
1
Notes:
1. EPCOS: www.epcos.com.
2. AVX: www.avx.com.
3. Murata: www.murata.com.
4. TDK: www.tdk.com.
5. SANYO: www.sanyo.com.
6. Diode Inc.: www.diodes.com.
7. Vishay: www.vishay.com.
8. Central Semi: www.centralsemi.com.
9. Cooper Bussmann: www.cooperbussmann.com.
10. Micrel, Inc.: www.micrel.com.
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Micrel, Inc.
MIC26400
PCB Layout
Figure 8. MIC26400 Evaluation Board Top Layer
Figure 9. MIC26400 Evaluation Board Mid-Layer 1 (Ground Plane)
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Micrel, Inc.
MIC26400
PCB Layout (Continued)
Figure 10. MIC26400 Evaluation Board Mid-Layer 2
Figure 11. MIC26400 Evaluation Board Bottom Layer
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Micrel, Inc.
MIC26400
Recommended Land Pattern
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M9999-070110-A
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Micrel, Inc.
MIC26400
Package Information
28-Lead 5mm x 6mm MLF® (YJL)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2010 Micrel, Incorporated.
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M9999-070110-A
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