RT5779A_技术文档

®

RT5779A/B

6V, 5A, 1.5MHz ACOT^TMSynchronous Step-Down Converter

in TSOT-23-8 package

General Description

Features

^5A Converter With Built-In 20mΩ/18mΩ Low R_DS(ON)

Power FETs

The RT5779A/B is a simple, easy-to-use, 5Asynchronous

step-downDC-DC converter with an input supply voltage

range of 2.5V to 6V. The device build-in an accurate 0.6V

( 1.5%) reference voltage and can operate in 100% duty

cycle as a very low dropout voltage regulator. the

RT5779A/B integrates low R_DS(ON)power MOSFETs to

achieve high efficiency and is available in TSOT-23-8 (FC)

package.

^Input Supply Voltage Range : 2.5V to 6V

^Output Voltage Range : 0.6V to 6V

^Operates Up to 100% Duty Cycle

^Advanced Constant On-Time (ACOT^TM) Control

^Ultrafast Transient Response

^No Needs For External Compensations

^Optimized for Low-ESR Ceramic Output Capacitors

^0.6V 1.5% High-Accuracy Feedback Reference

Voltage

The RT5779A/B adopts Advanced Constant On-Time

(ACOT^TM) control architecture to provide an ultrafast

transient response with few external components and to

operate in nearly constant switching frequency over the

line, load, and output voltage range. The RT5779A/B is

designed to operate at 1.5MHz fixed frequency. While

the RT5779A automatically enters power saving mode

(PSM) operation at light load to maintain high efficiency.

RT5779B operates in Forced PWM over the loading range,

that helps meet tight voltage regulation accuracy

requirements.

 35μA Operation Quiescent Current (RT5779A)

 Robust Over-Current Protection for Both FETs

 Optional for Operation Modes :

 Power Saving Mode (PSM) (RT5779A)

 Forced PWM Mode (RT5779B)

 Fixed Switching Frequency : 1.5MHz

 Monotonic Start-Up for Pre-Biased Output

 Individual Enable Control Input

 Power Good Indicator

The RT5779A/B senses both FETs current for a robust

over-current protection. It prevents the device from the

catastrophic damage in output short circuit, over current

or inductor saturation. The individual enable control input

and power good indicator provides flexible system power

sequence control a built-in soft-start function prevents

inrush current during start-up. The device also includes

input under-voltage lockout, output under-voltage

protection, and over-temperature protection (thermal

shutdown) to provide safe and smooth operation in all

operating conditions.

 Built-In Internally Fixed Soft-Start (Typ. 1.5ms)

 100% Duty Cycle Mode

 Internal Output Discharge

 Input Under-Voltage Lockout (UVLO)

 Output Under-Voltage Protection (UVP) with Hiccup

Mode

 Over-Temperature Protection

 Available in TSOT-23-8 (FC) Package

Simplified Application Circuit

L

V

IN

VIN

RT5779A/B

LX

FB

V

OUT

C

IN

R

PGOOD

R1

R2

C

OUT

PGOOD

Enable

PGOOD

EN

VOUT

AGND

PGND

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RT5779A/B

Ordering Information

RT5779A/B

Pin Configuration

(TOP VIEW)

Package Type

J8F : TSOT-23-8 (FC)

Lead Plating System

G : Green (Halogen Free and Pb Free)

8

7

2

6

3

5

4

PWM Operation Mode

A : Automatic PSM

B : Forced PWM

Note :

Richtek products are :

TSOT-23-8 (FC)

 RoHS compliant and compatible with the current require-

ments of IPC/JEDEC J-STD-020.

 Suitable for use in SnPb or Pb-free soldering processes.

Marking Information

RT5779AGJ8F

Applications

 WLANASIC Power / Storage (SSDand HDD)

28= : Product Code

28=DNN

DNN : Date Code

 Mobile Phones and HandheldDevices

 STB, Cable Modem, and xDSL Platforms

 General Purpose for POL LV Buck Converter

RT5779BGJ8F

27= : Product Code

27=DNN

DNN : Date Code

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RT5779A/B

Functional Pin Description

Pin No.

Pin Name

Pin Function

Open-drain power-good indication output. The power-good function is activated

after soft-start is finished.

1

PGOOD

Power input. The input voltage range is from 2.5V to 6V. Connect a suitable input

bypass capacitor (typically of greater than 10F) between this pin and PGND. The

bypass capacitor should be placed as close to the IC as possible.

2

VIN

Switch node between the internal switch and the synchronous rectifier. Connect

the output LC filter from this pin to the output load.

3

4

5

6

LX

Power ground. This pin, connected to analog ground, must be soldered to a large

PCB copper area for maximum power dissipation.

PGND

VOUT

AGND

Output voltage sense input. This pin is used to monitor and adjust output voltage

to enhance load transient regulation.

Analog ground. It provides a ground return path for the control circuitry and

internal reference.

Feedback voltage input. Connect this pin to the midpoint of the external feedback

resistive divider to set the output voltage of the converter to the desired regulation

level. The device regulates the FB voltage at a feedback reference voltage,

typically 0.6V.

7

8

FB

EN

Enable control input. A logic-high enables the converter; a logic-low forces the

device into shutdown mode.

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RT5779A/B

Functional Block Diagram

EN

VOUT

TON

AGND

VIN

UVLO

OTP

Shutdown

Control

LX

Error Amplifier

Comparator

+

-

+

-

FB

Logic

Control

LX

Driver

Current

Limit

Detector

LX

V

REF

Ramp

Generator

PGOOD

+

-

LX

AZC

FB

PGND

Operation

The RT5779A/B is a low-voltage, high-efficiency,

synchronous step-downDC-DC converter that can deliver

up to 5A output current from a 2.5V to 6V input supply.

The RT5779A/B adoptsACOT^TMcontrol mode, which can

reduce the output capacitance and provide ultrafast

transient responses, and allow minimal component sizes

without any additional external compensation network. The

device includes a built-in ramp voltage generator, which

takes up the virtual inductor current as an input. With the

internal ramp signal, the device can be compensated to

achieve good stability even with low-ESR ceramic

capacitors, since the need for the output capacitor's ESR

to generate an ESR ramp voltage can be eliminated.

is below the current limit I_{LIM_L}. During the on-time, the

high-side switch is turned on and the inductor current

ramps up linearly. After the on-time, the high-side switch

is turned off and the synchronous rectifier is turned on

and the inductor current ramps down linearly. If the output

voltage has not reached its nominal level, another on-time,

however, can only be generated after a short blanking time,

or called minimum off-time, which is triggered by the

minimum-off-time one-shot generator. It is to prevent

another immediate on-time being triggered during the noisy

switching time and allow the feedback voltage and current

sense signals to settle. The minimum off-time t_{OFF_MIN}is

kept short so that the inductor current can be raised up

quickly by rapidly repeated on-times when needed. Such

feature makes reaction speed of the ACOT-based

converters to load transients extremely fast.

Low V_INACOT^TMOne-Shot Operation

For a low V_INACOT converter, a built-in error amplifier is

used to keep track of the feedback voltage, as shown in

the Functional Block Diagram. In steady state, the error

amplifier compares the feedback voltage V_FBand an

internal reference voltage. If the virtual inductor current

ramp voltage is lower than the output of the error amplifier,

a new pre-determined fixed on-time will be triggered by

the on-time one-shot generator, provided that minimum-

off-time one-shot is cleared and the measured inductor

current through the synchronous rectifier (low-side switch)

Enable Control

The RT5779A/B provides an EN pin, as an external chip

enable control, to enable or disable the device. If V_ENis

held below a logic-low threshold voltage (V_ENL) of the

enable input (EN), the converter will enter into shutdown

mode, that is, the converter is disabled and switching is

inhibited even if the VIN voltage is above VIN under-voltage

lockout threshold (V_UVLO). During shutdown mode, the

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DS5779A/B-00 October 2017

RT5779A/B

supply current can be reduced to I_SHDN(1μA or below). If

the EN voltage rises above the logic-high threshold voltage

(V_ENH) while the V_INvoltage is higher than UVLO threshold

(V_UVLO), the device will be turned on, that is, switching

being enabled and soft-start sequence being initiated.

the low-side current limit.

If the output load current exceeds the available inductor

current (clamped by the above-mentioned low-side current

limit), the output capacitor needs to supply the extra

current such that the output voltage will begin to drop. If it

drops below the output under-voltage protection trip

threshold, the IC will stop switching to avoid excessive

heat.

Input Under-Voltage Lockout

In addition to the EN pin, the RT5779A/B also provides

enable control through the VIN pin. It features an under-

voltage lockout (UVLO) function that monitors the internal

linear regulator (VCC). If V_ENrises above V_ENHfirst,

switching will still be inhibited until the VIN voltage rises

above V_UVLO. It is to ensure that the internal regulator is

ready so that operation with not-fully-enhanced internal

MOSFET switches can be prevented. After the device is

powered up, if the input voltage V_INgoes below the UVLO

falling threshold voltage (V_UVLO− ΔV_UVLO), this switching

will be inhibited; if V_INrises above the UVLO-rising

threshold (V_UVLO), the device will resume switching.

Output Under-Voltage Protection

The RT5779A/B includes output under-voltage protection

(UVP) against over-load or short-circuited condition by

constantly monitoring the feedback voltage V_FB. If V_FB

drops below the under-voltage protection trip threshold

(typically 50% of the internal reference voltage), the UV

comparator will go high to turn off both the internal high-

side and low-side MOSFET switches.

Hiccup Mode

If the output under-voltage condition continues for a period

of time, the RT5779A/B will enter output under-voltage

protection with hiccup mode. During hiccup mode, the

device remains shut down. After a period of time, a soft-

start sequence for auto-recovery will be initiated. Upon

completion of the soft-start sequence, if the fault condition

is removed, the converter will resume normal operation;

otherwise, such cycle for auto-recovery will be repeated

until the fault condition is cleared. Hiccup mode allows

the circuit to operate safely with low input current and

power dissipation, and then resume normal operation as

soon as the over-load or short-circuit condition is removed.

Over-Current Protection

The RT5779A/B is protected from over current conditions

by cycle-by-cycle current limiting on both the high-side

MOSFET and the low-side MOSFET. The robust over

current protection mechanism prevents the converter to

be damaged from a catastrophic condition, i.e. the inductor

is shorted or saturated.

High-Side MOSFET Over-Current Protection

The device senses high-side current after a deglitch time

when the high-side MOSFET is turned-on. Each cycle

when the sensed current is higher than the high-side switch

peak current limit threshold, I_{LIM_H}, then the device enters

high side over current protection. At the same time, the

high-side MOSFET is turned off and the low-side MOSFET

is turned on.

Soft-Start (SS)

The soft-start function is used to prevent large inrush

currents while the converter is being powered up. The

RT5779A/B provides an internal soft-start feature for inrush

control.During the start-up sequence, the internal capacitor

is charged by an internal current source I_SSto generate a

soft-start ramp voltage as a reference voltage to an error

amplifier. The device will initiate switching and the output

voltage will smoothly ramp up to its targeted regulation

voltage only after this ramp voltage is greater than the

feedback voltage V_FBto ensure the converters have a

smooth start-up. The typical soft-start time is 1.5ms.

Low-Side MOSFET Over-Current Protection

The RT5779A/B detects low side current after a minimum-

off time while the low-side MOSFET is turned on. If the

detected low-side current is higher than the low-side switch

valley current limit threshold, I_{LIM_L}, the on-time one-shot

will be inhibited until the inductor current ramps down to

the current limit level (I_LIM). That is, another on-time can

only be triggered when the inductor current goes below

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RT5779A/B

Power Good Indication

The RT5779A/B provides a power-good (PGOOD) open-

drain output pin. It is to be connected to an external voltage

source through a pull-up resistor. The power-good function

is activated after soft-start is finished and is controlled by

a comparator connected to the feedback signal V_FB. If

V_FBrises above a power-good threshold (V_{TH_PGLH}

)

(typically 95% of the target value), the PGOOD pin will be

in high impedance and V_PGOODwill be held high after a

certain delay elapsed. When V_FBdrops by a power-good

hysteresis (ΔV_{TH_PGLH}) (typically 5% of the target value)

or exceeds V_{TH_PGHL}(typically 110% of the target value),

the PGOOD pin will be pulled low. For V_FBhigher than

V_{TH_PGHL}, V_PGOODcan be pulled high again if V_FBdrops

back by a power-good hysteresis (ΔV_{TH_PGHL}) (typically

5% of the target value). Once being started-up, if any

internal protection is triggered, PGOOD will be pulled low

toGND.

Over-Temperature Protection (Thermal Shutdown)

The RT5779A/B includes an over-temperature protection

(OTP) circuitry to prevent overheating due to excessive

power dissipation. The OTP will shut down switching

operation when junction temperature exceeds a thermal

shutdown threshold T_SD. Once the junction temperature

cools down by a thermal shutdown hysteresis (ΔT_SD), the

IC will resume normal operation with a complete soft-start.

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RT5779A/B

Absolute Maximum Ratings (Note 1)

 Supply Input Voltage, VIN ----------------------------------------------------------------------------------------- −0.3V to 7V

 LX Pin Switch Voltage ---------------------------------------------------------------------------------------------- −0.3V to (V_IN+ 0.3V)

<10ns ------------------------------------------------------------------------------------------------------------------ −5V to 8.5V

 Other Pins------------------------------------------------------------------------------------------------------------- −0.3V to (V_IN+ 0.3V)

 PowerDissipation, P_D@ T_A= 25°C

TSOT-23-8 (FC) ------------------------------------------------------------------------------------------------------ 1.46W

 Package Thermal Resistance (Note 2)

TSOT-23-8 (FC), θ_JA------------------------------------------------------------------------------------------------- 68.2°C/W

TSOT-23-8 (FC), θ_JC------------------------------------------------------------------------------------------------ 17.1°C/W

 Junction Temperature ----------------------------------------------------------------------------------------------- 150°C

 Lead Temperature (Soldering, 10 sec.)------------------------------------------------------------------------- 260°C

 Storage Temperature Range -------------------------------------------------------------------------------------- −65°C to 150°C

 ESD Susceptibility (Note 3)

HBM (Human Body Model)---------------------------------------------------------------------------------------- 2kV

Recommended Operating Conditions (Note 4)

 Supply Input Voltage, VIN ----------------------------------------------------------------------------------------- 2.5V to 6V

 Junction Temperature Range-------------------------------------------------------------------------------------- −40°C to 125°C

 Ambient Temperature Range-------------------------------------------------------------------------------------- −40°C to 85°C

Electrical Characteristics

(V_IN= 5V, T_A= 25°C, unless otherwise specified)

Parameter

Supply Voltage

Symbol

Test Conditions

Min

Typ

Max Unit

Input Operating Voltage

V_IN

2.5

--

6

V

Under-Voltage Lockout

Threshold

V_UVLO

2.15

2.3

2.45

Under-Voltage Lockout

Threshold Hysteresis

∆V_UVLO

I_SHDN

I_Q

--

260

--

mV

A

Shutdown Current

Quiescent Current

Enable Voltage

V_EN= 0V

RT5779A

RT5779B

--

0

1

50

--

35

600

V_ENH

V_ENL

V_ENrising

1.2

--

0.4

--

Enable Threshold Voltage

V

V_ENfalling

_EN= 2V

V

--

1.5

0

Enable Input Current

I_IH

A

V_EN= 0V

--

Feedback Voltage

Feedback Input Current

Feedback Voltage

I_FB

V_FB= 0.6V

--

10

--

nA

V

V_FB

0.588 0.6 0.612

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RT5779A/B

Parameter

Symbol

Test Conditions

Min

Typ

Max

Unit

Current Limit

High-Side Switch Peak

Current Limit

I_{LIM_H}

--

9.7

7

--

A

Low-Side Switch Valley

Current Limit

I_{LIM_L}

Switching

Switching Frequency

f_SW

V_OUT= 1.2V

1300 1500 1700

kHz

ns

Minimum Off-Time

Internal MOSFET

High-Side On-Resistance

Low-Side On-Resistance

Soft-Start

t_{OFF_MIN}

--

60

--

R_{DS(ON)_H}

R_{DS(ON)_L}

--

20

18

--

m

Fixed Soft-Start Time

V_OUT

t_SS

1

1.5

1

--

ms

Output Discharge Resistor

Power Good

(Note 5)

--

k

Power-Good High Threshold V_{TH_PGLH}

V_FBrising. PGOOD goes high

--

95

5

--

%V_FB

s

Power-Good High Hysteresis V_{TH_PGLH}V_FBfalling. PGOOD goes low

Power-Good Low Threshold V_{TH_PGHL}V_FBrising. PGOOD goes low

110

5

Power-Good Low Hysteresis V_{TH_PGHL}V_FBfalling. PGOOD goes high

Power Good Delay Time

15

Power Good Sink Current

I_PGOODsinks 1mA

Capability

--

0.4

--

V

Power Good Internal Pull Up

Resistance

550

k

Over-Temperature Protection

Thermal Shutdown

T_SD

--

150

30

--

C

Thermal Shutdown

Hysteresis

∆T_SD

Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are

stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in

the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may

affect device reliability.

Note 2. θ_JAis measured in the natural convection at T_A= 25°C on a four-layer Richtek Evaluation Board for TSOT-23-8 (FC).

Note 3. Devices are ESD sensitive. Handling precaution is recommended.

Note 4. The device is not guaranteed to function outside its operating conditions.

Note 5. Guarantee by design.

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RT5779A/B

Typical Application Circuit

L

0.47µH

V

3

7

2

V

IN

OUT

VIN

LX

FB

2.5V to 6V

1.2V/5A

C

22µF

IN

C

44µF

R

OUT

PGOOD

100k

R1

RT5779A/B

20k

1

8

PGOOD

Enable

PGOOD

EN

R2

20k

5

VOUT

PGND

4

AGND

6

Table 1. Suggested Component Values

R1 (k)

13.3

20

R2 (k)

20

L (H)

0.33

0.47

V_OUT(V)

1

C_OUT(F)

44

1.2

20

1.8

40.2

63.4

90.9

20

2.5

20

3.3

20

Note : All the input and output capacitances are the suggested values, which refer to the effective capacitances, and are subject

to any de-rating effect, like a DC bias.

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RT5779A/B

Typical Operating Characteristics

Efficiency vs. Output Current

100

90

80

70

60

50

40

30

20

10

0

90

80

V_OUT= 3.3V

V_OUT= 2.5V

V_OUT= 3.3V

70

60

50

40

30

20

10

0

V

_OUT= 2.5V

_OUT= 1.8V

V

_OUT= 1.8V

_OUT= 1.2V

V_OUT= 1.2V

_OUT= 1V

V_OUT= 1V

V

RT5779A, V_IN= 5V

10

RT5779B, V_IN= 5V

4

0.001

0.01

0.1

1

0

1

2

3

5

6

Output Current (A)

Output Voltage vs. Output Current

1.32

1.30

1.28

1.26

1.24

1.22

1.20

1.18

1.16

1.14

1.12

1.10

1.08

3.40

3.38

3.36

3.34

3.32

3.30

3.28

3.26

3.24

3.22

3.20

V_IN= 6V

_IN= 2.5V

V_IN= 3.6V

V

RT5779A, V_OUT= 1.2V

RT5779A, V_OUT= 3.3V

0

1

2

3

4

5

0

1

2

3

4

Output Current (A)

Output Voltage vs. Input Voltage

1.32

1.30

1.28

1.26

1.24

1.22

1.20

1.18

1.16

1.14

1.12

1.10

1.08

3.40

3.38

3.36

3.34

3.32

3.30

3.28

3.26

3.24

3.22

3.20

I_OUT= 0A

I_OUT= 5A

I_OUT= 0A

I_OUT= 5A

RT5779A, V_OUT= 1.2V

RT5779A, V_OUT= 3.3V

2.5

3

3.5

4

4.5

5

5.5

6

3.5

4

4.5

5

5.5

Input Voltage (V)

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RT5779A/B

UVLO vs. Temperature

EN Voltage Threshold vs. Temperature

2.45

2.40

2.35

2.30

2.25

2.20

2.15

2.10

2.05

2.00

1.95

0.90

0.85

0.80

0.75

0.70

0.65

0.60

0.55

0.50

Rising

Falling

RT5779A/B

100 125

-50

-25

0

25

50

75

-50

-25

0

25

50

75

100

125

Temperature (°C)

Load Transient Response

RT5779A, V_IN= 5V, V_OUT= 3.3V,

I_OUT= 0A to 5A to 0A

V_OUT

(20mV/Div)

V_OUT

(40mV/Div)

RT5779B, V_IN= 5V, V_OUT= 3.3V,

I_OUT= 0A to 5A to 0A

LX

(5V/Div)

LX

(4V/Div)

I_OUT

(3.5A/Div)

I_OUT

(3.5A/Div)

Time (200μs/Div)

Time (400μs/Div)

Load Transient Response

RT5779A, V_IN= 5V,

RT5779B, V_IN= 5V, V_OUT= 1.2V,

I_OUT= 0A to 5A to 0A

V_OUT= 1.2V, I_OUT= 0A to 5A to 0A

V_OUT

(20mV/Div)

V_OUT

(20mV/Div)

LX

(4V/Div)

LX

(4V/Div)

I_OUT

(3.5A/Div)

I_OUT

(3.5A/Div)

Time (400μs/Div)

Time (200μs/Div)

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RT5779A/B

Application Information

The output stage of a synchronous buck converter is

composed of an inductor and capacitor, which stores and

delivers energy to the load, and forms a second-order low-

pass filter to smooth out the switch node voltage to maintain

a regulated output voltage.

and the approximate inductance can be calculated by the

selected input voltage, output voltage, switching frequency

(f_SW), and inductor current ripple (ΔI_L), as below :

V

 V  V



IN OUT



OUT

L =

V f

IN SW

I

L

Once the inductance is chosen, the inductor ripple current

(ΔI_L) and peak inductor current (I_{L_PEAK}) can be calculated,

as below :

100% Duty-Cycle

When the input voltage drops, these Buck converters

gradually increase the duty-cycle and will continuously

switch-on the high side MOSFET when the input voltage

drops below the regulated output voltage. This function is

especially suitable in battery powered applications, and

can extend application operation time when the battery is

almost depleted.

V_OUT V_IN V_OUT





I_L=

V f_SWL

IN

1

2

I_{L_PEAK}= I_{OUT_MAX}



I_L

1

2

I_{L_VALLEY}= I_{OUT_MAX}



I_L

where I_{OUT_MAX}is the maximum rated output current or

the required peak current.

Inductor Selection

When designing the output stage of the synchronous buck

converter, it is recommended to start with the inductor.

However, it may require several iterations because the

exact inductor value is generally flexible and is optimized

for low cost, small form factor, and high overall performance

of the converter. Further, inductors vary with manufacturers

in both material and value, and typically have a tolerance

of 20%.

The inductor must be selected to have a saturation current

and thermal rating which exceed the required peak inductor

current I_{L_PEAK}. For a robust design to maintain control of

inductor current in overload or short-circuit conditions,

some applications may desire inductor saturation current

rating up to the high-side switch current limit of the device.

However, the built-in output under-voltage protection (UVP)

feature makes this unnecessary for most applications.

Three key inductor parameters to be specified for operation

with the device are inductance (L), inductor saturation

current (I_SAT), and DC resistance (DCR), which affects

performance of the output stage. An inductor with lower

DCR is recommended for applications of higher peak

current or load current, and it can improve system

performance. Lower inductor values are beneficial to the

system in physical size, cost, DCR, and transient

response, but they will cause higher inductor peak current

and output voltage ripple to decrease system efficiency.

Conversely, higher inductor values can increase system

efficiency at the expense of larger physical size, slower

transient response due to the longer response time of the

inductor. Agood compromise among size, efficiency, and

transient response can be achieved by setting an inductor

current ripple (ΔI_L) of about 20% to 50% of the desired full

output load current. To meet the inductor current ripple

(ΔI_L) requirements, a minimum inductance must be chosen

I_{L_PEAK}should not exceed the minimum value of the

device's high-side switch current limit because the device

will not be able to supply the desired output current. By

reducing the inductor current ripple (ΔI_L) to increase the

average inductor current (and the output current), I_{L_PEAK}

can be lowered to meet the device current limit

requirement.

For best efficiency, a low-loss inductor having the lowest

possible DCR that still fits in the allotted dimensions will

be chosen. Ferrite cores are often the best choice.

However, a shielded inductor, possibly larger or more

expensive, will probably give fewer EMI and other noise

problems.

The following design example is illustrated to walk through

the steps to apply the equations defined above. The

RT5779A/B's TypicalApplication Circuit for output voltage

of 1.2V at maximum output current of 5A and an input

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12

DS5779A/B-00 October 2017

RT5779A/B

voltage of 5V with inductor current ripple of 1.2A (i.e. 24%,

in the recommended range of 20% to 50%, of the

maximum rated output current) is taken as the design

example. The approximate minimum inductor value can

first be calculated as below :

usually occurs at 50% duty cycle, that is, V_IN= 2 x V_OUT.

The maximum I_RMSas I_RMS(Max), can be approximated

,

as 0.5 x I_{OUT_MAX}, where I_{OUT_MAX}is the maximum rated

output current. Besides, the variation of the capacitance

value with temperature, DC bias voltage, switching

frequency, and allowable peal-to-peak ripple voltage that

reflects back to the input, also need to be taken into

consideration. For example, the capacitance value of a

capacitor decreases as the DC bias across the capacitor

increases; also, higher switching frequency allows the use

of input capacitors of smaller capacitance values.

1.2 5 1.2

51500kHz1.2A





L =

= 0.5μH

where f_SWis 1500kHz. The inductor current ripple will be

set at 1.2A, as long as the calculated inductance of 0.5μH

is used. However, the inductor of the exact inductance

value may not be readily available, and therefore an inductor

of a nearby value will be chosen. In this case, 0.47μH

inductance is available and actually used in the Typical

Application Circuit. The actual inductor current ripple (ΔI_L)

and required peak inductor current (I_{L_PEAK}) can be

calculated as below :

Ceramic capacitors are most commonly used to be placed

right at the input of the converter to reduce ripple voltage

amplitude because only ceramic capacitors have

extremely low ESR which is required to reduce the ripple

voltage. Note that the capacitors need to be placed as

close as to the input pins as possible for highest

effectiveness. Ceramic capacitors are preferred also due

to their low cost, small size, high RMS current ratings,

robust inrush surge current capabilities, and low parasitic

inductance, which helps reduce the high-frequency ringing

on the input supply.

1.2 5 1.2

51500kHz0.47μH





I_L=

= 1.294A

1

2

1.294

I_{L_PEAK}= I_{OUT_MAX}



I_L= 5 +

= 5.647A

2

For the 0.47μH inductance value, the inductor saturation

current and thermal rating should exceed 5.647A.

However, care must be taken when ceramic capacitors

are used at the input, and the input power is supplied by

a wall adapter, connected through a long and thin wire.

When a load step occurs at the output, a sudden inrush

current will surge through the long inductive wire, which

can induce ringing at the device's power input and

potentially cause a very large voltage spike at the VIN pin

to damage the device. For applications where the input

power is located far from the device input, it may be required

that the low-ESR ceramic input capacitors be placed in

parallel with a bulk capacitor of other types, such as

tantalum, electrolytic, or polymer, to dampen the voltage

ringing and overshoot at the input, caused by the long

input power path and input ceramic capacitor.

Input Capacitor Selection

Input capacitors are needed to smooth out the RMS ripple

current (I_RMS) imposed by the switching currents and

drawn from the input power source, by reducing the ripple

voltage amplitude seen at the input of the converters. The

voltage rating of the input filter capacitors must be greater

than the maximum input voltage. It's also important to

consider the ripple current capabilities of capacitors.

The RMS ripple current (I_RMS) of the regulator can be

determined by the input voltage (V_IN), output voltage

(V_OUT), and rated output current (I_OUT) as the following

equation :

V

IN

V

OUT

I

= I



1

RMS

OUT

It is suggested to choose capacitors with higher

temperature ratings than required. Several ceramic

capacitors may be parallel to meet application

requirements, such as the RMS current, size, and height.

The Typical Application Circuit can use one 22μF, or two

10μF and one high-frequency-noise-filtering 0.1μF low-ESR

ceramic capacitors at the input.

IN

From the above, the maximum RMS input ripple current

occurs at maximum output load, which will be used as

the requirements to consider the current capabilities of

the input capacitors. Furthermore, for a single phase buck

converter, the duty cycle is approximately the ratio of

output voltage to input voltage. The maximum ripple voltage

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13

RT5779A/B

Output Capacitor Selection :

capacitors with ESR of about 5mΩ as output capacitors,

the two output ripple components are as below :

Output capacitance affects the output voltage of the

converter, the response time of the output feedback loop,

and the requirements for output voltage sag and soar. The

sag occurs after a sudden load step current applied, and

the soar occurs after a sudden load removal. Increasing

the output capacitance reduces the output voltage ripple

and output sag and soar, while it increases the response

time that the output voltage feedback loop takes to respond

to step loads, Therefore, there is a tradeoff between output

capacitance and output response. It is recommended to

choose a minimum output capacitance to meet the output

voltage requirements of the converter, and have a quick

transient response to step loads.

V_{P-P_ESR}= I_LR_ESR= 1.294A5m = 6.47mV

I_L

1.294A

844μF1500kHz

V_{P-P_C}

=

8C_OUTf_SW

ꢀꢀꢀꢀꢀ= 2.451mV

V_P-P= V_{P-P_ESR} V_{P-P_C}= 8.921mV

^Output Transient Undershoot and Overshoot

In addition to the output voltage ripple at the switching

frequency, the output capacitor and its ESR also affect

output voltage sag, which is undershoot on a positive

load step, and output voltage soar, which is overshoot

on a negative load step. With the built-in ACOT^TM

architecture, the IC can have very fast transient

responses to the load steps and small output transients.

The ESR of the output capacitor affects the damping of

the output filter and the transient response. In general,

low-ESR capacitors are good choices due to their

excellent capability in energy storage and transient

performance. The RT5779A/B, therefore, is specially

optimized for ceramic capacitors. Consider also DC bias

and aging effects while selecting the output capacitor.

However, the combination of a small ceramic output

capacitor (that is, of little capacitance) and a low output

voltage (that is, only little charge stored in the output

capacitor), used in low-duty-cycle applications (which

require high inductance to get reasonable ripple currents

for high input voltages), causes an increase in the size

of voltage variations (i.e. sag/soar) in response to very

quick load changes. Typically, the load changes slowly,

compared with the IC's switching frequency. However,

for present-day applications, more and more digital

blocks may exhibit nearly instantaneous large transient

load changes. Therefore, in the following section, how

to calculate the worst-case voltage swings in response

to very fast load steps will be explained in details.

 Output Voltage Ripple

The output voltage ripple at the switching frequency is

a function of the inductor current ripple going through

the output capacitor's impedance. To derive the output

voltage ripple, the output capacitor with capacitance,

C_OUT, and its equivalent series resistance, R_ESR, must

be taken into consideration. The output peak-to-peak

ripple voltage ΔV_P-P, caused by the inductor current ripple

ΔI_L, is characterized by two components, which are ESR

ripple ΔV_{P-P_ESR}and capacitive ripple ΔV_{P-P_C}, can be

expressed as below :

Both of the output transient undershoot and overshoot

have two components : a voltage step caused by the

output capacitor's ESR, and a voltage sag or soar due

to the finite output capacitance and the inductor current

slew rate. The following formulas can be used to check

if the ESR is low enough (which is usually not a problem

with ceramic capacitors) and if the output capacitance

is large enough to prevent excessive sag or soar on

very fast load steps, with the chosen inductor value.

V_P-P= V_{P-P_ESR} V_{P-P_C}

V_{P-P_ESR}= I_LR_ESR

I_L

V_{P-P_C}

=

8C_OUTf_SW

If ceramic capacitors are used as the output capacitors,

both the components need to be considered due to the

extremely low ESR and relatively small capacitance.

The voltage step (ΔV_{OUT_ESR}) caused by the ESR is a

function of the load step (ΔI_OUT) and the ESR (R_ESR) of

the output capacitor, described as below :

For the RT5779A/B's Typical Application Circuit for

output voltage of 1.2V, and actual inductor current ripple

(ΔI_L) of 1.294A, using two paralleled 22μF ceramic

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14

DS5779A/B-00 October 2017

RT5779A/B

ΔV_{OUT_ESR}= ΔI_OUTx R_ESR

An external MOSFET can be added for the EN pin to be

logic-controlled, as shown in Figure 2. In this case, a

100kΩ pull-up resistor, R_EN, is connected between V_IN

and the ENpin. The MOSFET Q1 will be under logic control

to pull down the EN pin. To prevent the device being

enabled when VIN is smaller than the VOUT target level

or some other desired voltage level, a resistive divider (R_EN1

and R_EN2) can be used to externally set the input under-

voltage lockout threshold, as shown in Figure 3.

The voltage amplitude (ΔV_{OUT_SAG}) of the capacitive sag

is a function of the load step (ΔI_OUT), the output capacitor

value (C_OUT), the inductor value (L), the input-to-output

voltage differential, and the maximum duty cycle (D_MAX).

And, the maximum duty cycle during a fast transient

can be determined by the on-time (t_ON) and the minimum

off-time (t_{OFF_MIN}) since the ACOT^TMcontrol scheme

will ramp the current during on-times, which are spaced

apart by a minimum off-time, that is, as fast as allowed.

The approximate on-time (neglecting parasitics) and

maximum duty cycle for a given input and output voltage

can be calculated according to the following equations :

R

EN

V

EN

RT5779A/B

IN

C

EN

GND

V

OUT

t

=

ON

V f

IN SW

Figure 1. Enable Timing Control

t

ON

D

MAX

=

t

 t

OFF_MIN

ON

R

EN

Note the actual on-time will be slightly larger than the

calculated one as the IC will automatically adapt to

compensate the internal voltage drops, such as the

voltage across high-side switch due to on-resistance.

However, both of these can be neglected since the on-

time increase can compensate for the voltage drops.

The output voltage sag (ΔV_{OUT_SAG}) can then be

100k

V

EN

RT5779A/B

GND

IN

Q1

Enable

Figure 2. Logic Control for the EN Pin

calculated as below :

R

EN1

2

V

IN

EN

L(I

)

OUT

V

=

OUT_SAG

2C

 V D

 V





R

EN2

OUT

IN

MAX OUT

RT5779A/B

GND

The voltage amplitude of the capacitive soar is a function

of the load step (ΔI_OUT), the output capacitor value (C_OUT),

the inductor value (L), and the output voltage (V_OUT).

And the output voltage soar (ΔV_{OUT_SOAR}) can be

Figure 3. ResistorDivider for Under-Voltage

Lockout Threshold Setting

calculated as below :

2

L(I

2C

)

OUT

Output Voltage Setting

V

=

OUT_SOAR

 V

OUT

The output voltage can be programmed by a resistive divider

from the output to ground with the midpoint connected to

the FB pin. The resistive divider allows the FB pin to sense

a fraction of the output voltage as shown in Figure 4. The

EN Pin for Start-Up and Shutdown Operation

For automatic start-up, the EN pin, with high-voltage rating,

can be connected to the input supply V_IN, either directly

or through a 100kΩ resistor. The large built-in hysteresis

band makes the ENpin useful for simple delay and timing

circuits. The EN pin can be externally connected to V_IN

by adding a resistor R_ENand a capacitor C_EN, as shown in

Figure 1, to have an additional delay. The time delay can

be calculated with the EN's internal threshold, at which

switching operation begins.

output voltage is set according to the following equation :

R1

V_OUT V_{TH_FB}(1 +

)

R2

where V_{TH_FB}is around 0.6V (Typ).

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DS5779A/B-00 October 2017

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15

RT5779A/B

V

package, the PCB layout, the rate of surrounding airflow,

and the difference between the junction and ambient

temperatures. The maximum power dissipation can be

calculated using the following formula :

OUT

R1

FB

RT5779A/B

R2

P_D(MAX)= (T_J(MAX)− T_A) / θ_JA

GND

where T_J(MAX)is the maximum junction temperature, T_Ais

the ambient temperature, and θ_JAis the junction-to-ambient

thermal resistance.

Figure 4. Output Voltage Setting

The placement of the resistive divider should be within

5mm of the FB pin. The resistance of R2 is suggested

between 10kΩ and 100kΩ to minimize power consumption

and noise pick-up at the FB pin. Once R2 is chosen, the

resistance of R1 can then be obtained as below :

For continuous operation, the maximum operating junction

temperature indicated under Recommended Operating

Conditions is 125°C. The junction-to-ambient thermal

resistance, θ_JA, is highly package dependent. For a TSOT-

23-8 (FC) package, the thermal resistance, θ_JA, is

68.2°C/W on a four-layer Richtek Evaluation Board. The

maximum power dissipation at T_A= 25°C can be calculated

as below :

R2(V

 V

)

OUT

TH_FB

R1

V

TH_FB

For better output voltage accuracy, the divider resistors

(R1 and R2) with 1% tolerance or better should be used.

P_D(MAX)= (125°C − 25°C) / (68.2°C/W) = 1.46W for a

TSOT-23-8 (FC) package.

Power-Good Output

The PGOOD pin is an open-drain power-good indication

output and is to be connected to an external voltage source

through a pull-up resistor. The power-good function is

activated after soft-start is finished and is controlled by

the feedback signal V_FB. During soft-start, PGOOD is

actively held low and only allowed to transition high after

soft-start is over. If V_FBraises above a power-good

threshold (V_{TH_PGLH}) (typically 95% of the target value),

the PGOODpin will be in high impedance and V_PGOODwill

be held high after a certain delay elapsed. When V_FBdrops

by a power-good hysteresis (ΔV_{TH_PGLH}) (typically 5% of

the target value) or exceeds V_{TH_PGHL}(typically 110% of

the target value), the PGOOD pin will be pulled low. For

V_FBabove V_{TH_PGHL}, V_PGOODwill be pulled high again when

The maximum power dissipation depends on the operating

ambient temperature for the fixed T_J(MAX)and the thermal

resistance, θ_JA. The derating curves in Figure 5 allows

the designer to see the effect of rising ambient temperature

on the maximum power dissipation.

2.0

Four-Layer PCB

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2

0.0

V_FBdrops back by a power-good hysteresis (ΔV_{TH_PGHL}

)

(typically 5% of the target value). Once being started-up,

if any internal protection is triggered, PGOODwill be pulled

low to GND.

0

25

50

75

100

125

Ambient Temperature (°C)

Thermal Considerations

Figure 5. Derating Curve of Maximum PowerDissipation

The junction temperature should never exceed the

absolute maximum junction temperature T_J(MAX), listed

under Absolute Maximum Ratings, to avoid permanent

damage to the device. The maximum allowable power

dissipation depends on the thermal resistance of the IC

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DS5779A/B-00 October 2017

RT5779A/B

Layout Considerations

Layout is very important in high frequency switching

converter design. The PCB can radiate excessive noise

and contribute to converter instability with improper layout.

Certain points must be considered before starting a layout

using the IC.

 Make traces of the high current paths as short and wide

as possible.

 Put the input capacitor as close as possible to the device

pins (VINandGND).

 The LX node encounters high frequency voltage swings

so it should be kept in a small area. Keep sensitive

components away from the LX node to prevent stray as

possible.

 The GND pin should be connected to a strong ground

plane for heat sinking and noise protection.

 Avoid using vias in the power path connections that

have switched currents (from C_INto GND and C_INto

VIN) and the switching node (LX).

An TSOT-23-8 (FC)example of PCB layout guide is shown

in Figure 6 for reference.

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DS5779A/B-00 October 2017

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17

RT5779A/B

GND

C

IN

V

IN

The input capacitor

must be placed as

close to the IC as

possible

EN

FB

PGOOD

R2

R1

VIN

LX

AGND

VOUT

LX should be connected to

inductor by wide and short trace.

Keep sensitive components away

from this trace.

PGND

L

C

OUT

V

OUT

The output capacitor must

be placed near the IC

Figure 6. TSOT-23-8 (FC) PCB LayoutGuide

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18

DS5779A/B-00 October 2017

RT5779A/B

Outline Dimension

Dimensions In Millimeters

Dimensions In Inches

Symbol

Min.

0.700

0.000

1.397

0.220

2.591

2.692

0.585

0.080

0.300

Max.

1.000

0.100

1.803

0.380

3.000

3.099

0.715

0.254

0.610

Min.

0.028

0.000

0.055

0.009

0.102

0.106

0.023

0.003

0.012

Max.

A

A1

B

0.039

0.004

0.071

0.015

0.118

0.122

0.028

0.010

0.024

b

C

D

e

H

L

TSOT-23-8 (FC) Surface Mount Package

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19

RT5779A/B

Footprint Information

Footprint Dimension (mm)

Number of

Package

Tolerance

±0.10

P1

A

B

C

D

M

TSOT-28/TSOT-28(FC)/SOT-28

8

0.65

3.60

1.60

1.00

0.45

2.40

Richtek Technology Corporation

14F, No. 8, Tai Yuen 1^stStreet, Chupei City

Hsinchu, Taiwan, R.O.C.

Tel: (8863)5526789

Richtek products are sold by description only. Customers should obtain the latest relevant information and data sheets before placing orders and should verify

that such information is current and complete. Richtek cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek

product. Information furnished by Richtek is believed to be accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use;

nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent

or patent rights of Richtek or its subsidiaries.

www.richtek.com

20

DS5779A/B-00 October 2017


型号：	RT5779A
厂家：	RICHTEK TECHNOLOGY CORPORATION
描述：	暂无描述
文件：	总20页 (文件大小：293K)
中文：	中文翻译
下载：	下载PDF数据表文档文件

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